CN111987961A - Position-sensorless direct torque control method for permanent magnet synchronous motor - Google Patents

Position-sensorless direct torque control method for permanent magnet synchronous motor Download PDF

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CN111987961A
CN111987961A CN202010908811.7A CN202010908811A CN111987961A CN 111987961 A CN111987961 A CN 111987961A CN 202010908811 A CN202010908811 A CN 202010908811A CN 111987961 A CN111987961 A CN 111987961A
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flux
reference voltage
voltage
torque
axis
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张蔚
王家乐
翟良冠
杨泽贤
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Nantong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H02P21/0007—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control using sliding mode control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/10—Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H02P6/182—Circuit arrangements for detecting position without separate position detecting elements using back-emf in windings
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2203/00—Indexing scheme relating to controlling arrangements characterised by the means for detecting the position of the rotor
    • H02P2203/03—Determination of the rotor position, e.g. initial rotor position, during standstill or low speed operation

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

The invention discloses a position sensorless direct torque control method of a permanent magnet synchronous motorα、iβAnd stator voltage uα、uβInput to a modified Lonberg observer module to estimate back EMF
Figure DDA0002662469620000011
The back emf will then be estimated
Figure DDA0002662469620000012
Calculation of rotor position angle in phase locked loop
Figure DDA0002662469620000013
And estimating the rotational speed
Figure DDA0002662469620000014
The method is respectively applied to a dq/alpha beta coordinate transformation module, an abc/dq coordinate transformation module and motor rotating speed feedback. The observer subtracts the estimated current from the actually measured current to be used as an error value to perform closed-loop correction on the estimated value of the state observer, so that the observer can enter a stable and convergent state again under the condition that the system is unstable, the problem of large torque pulsation of the traditional direct torque control method is obviously improved, and the control system has good control precision and stability and good dynamic response capability.

Description

Position-sensorless direct torque control method for permanent magnet synchronous motor
Technical Field
The invention relates to the field of electromechanical control, in particular to a position-sensorless direct torque control method for a permanent magnet synchronous motor.
Background
With the wide application of permanent magnet synchronous motors, the requirements on the control performance of the permanent magnet synchronous motors are increasingly increased. Permanent magnet motors require position feedback for effective control, but installation, maintenance and repair of position sensors can add cost and, in some special cases, do not allow for the installation of position sensors. . The direct torque control has the advantages of fast dynamic response and strong robustness, and is widely applied in many fields. However, the torque ripple of the conventional direct torque control method is large, and affects the effect of the direct torque control. Therefore, the reduction of the torque ripple and the improvement of the stability have important significance for the position-free direct torque control of the permanent magnet synchronous motor. In a permanent magnet synchronous motor position sensorless control system, the relative electric signals in the windings are used to estimate the rotating speed to replace a mechanical sensor. Current position estimation algorithms can be divided into two categories, signal injection-based and observer-based. The former uses the salient polarity of the motor to estimate the position of the rotor, but the continuous injection of the excitation signal requires complex signal processing, resulting in low utilization rate of the inverter voltage and slow dynamic response. The method has the advantages that the rotation speed is estimated by means of the back electromotive force in the dynamic model, the method is easy to achieve in engineering, the Luenberger observer algorithm is one of the methods, the algorithm is simple in structure and stable, but dynamic response capability is poor due to the fact that the estimated back electromotive force needs to be filtered, estimation errors of the rotor position angle are large, the estimated value lags behind an actual value, and the like. Therefore, the method for controlling the Roberter observer can realize accurate tracking of the position of the rotor and has better dynamic response performance, and has wide development prospect.
Disclosure of Invention
In view of this, the present invention provides a position sensorless direct torque control method for a permanent magnet synchronous motor, which can improve the tracking accuracy of the rotor position of a motor driving system, improve the dynamic response performance, reduce the torque ripple, and improve the stability of the system.
The invention provides a permanent magnet synchronous motor direct torque sensorless control method, which comprises the following steps:
s1, estimating the rotating speed of the motor according to the given rotating speed omega through a PI speed regulator
Figure BDA0002662469600000011
Obtaining a reference torque Te by the difference value;
s2, collecting three-phase currents i of a, b and ca，ib，icObtaining the stator current i of the alpha and beta axes through coordinate transformationαAnd iβ(ii) a By duty cycle signal Sa、Sb、ScCalculating three-phase voltage ua，ub，ucObtaining stator voltage u of alpha, beta axis through coordinate transformationαAnd uβ(ii) a The three-phase current ia，ib，icThree-phase voltage ua，ub，ucAnd rotor position angle
Figure BDA0002662469600000012
The stator current i of d and q axes can be obtained by calculationd，iqAnd stator voltage ud，uq。
S3, enabling the d and q axis stator current id，iqAnd electron voltage ud，uqThe actual flux linkage flux and the actual torque Te can be obtained through calculation;
s4, inputting the difference value between the given reference flux linkage flux and the actual flux linkage flux into a sliding mode flux linkage adjuster, and passing through the sliding mode flux linkage adjusterOutputting d-axis reference voltage ud *(ii) a Inputting the difference value between the reference torque Te and the actual torque Te into a sliding mode torque regulator, and outputting a q-axis reference voltage u through the sliding mode torque regulatorq *；
S5, enabling the d-axis reference voltage ud *Q-axis reference voltage uq *And rotor position angle
Figure BDA0002662469600000013
Obtaining alpha-axis reference voltage u through coordinate transformationα *And a beta-axis reference voltage uβ *The alpha axis reference voltage uα *And a beta-axis reference voltage uβ *Inputting into SVPWM sine pulse width modulation module, outputting duty ratio signal S via SVPWM sine pulse width modulation modulea、Sb、ScThen the duty ratio signal Sa、Sb、ScInputting a three-phase inverter to control the on and off of the three-phase inverter, and realizing the driving of the permanent magnet synchronous motor;
s6, converting the alpha and beta axis stator current iα，iβAnd a, beta axis stator voltage uα，uβSubstituting improved Lorberg observer to calculate estimated back electromotive force of permanent magnet synchronous motor
Figure BDA0002662469600000014
S7, estimating the back electromotive force
Figure BDA0002662469600000015
Calculation of rotor position angle in phase locked loop
Figure BDA0002662469600000016
And estimating the rotational speed
Figure BDA0002662469600000017
And S8, repeating S1-S7 to realize the closed-loop stable operation of the motor.
Further, the d-axis reference voltage u in step S4d *And q-axis reference voltage uq *The calculation formula of (2) is as follows:
Figure BDA0002662469600000018
Figure BDA0002662469600000019
in the formula, sfluxThe flux is a sliding mode surface function of the flux linkage, flux is a reference flux linkage, flux is an actual flux linkage, and u is a reference flux linkaged *Is d-axis reference voltage, sTAs a sliding mode surface function of torque, Te is reference torque, Te is actual torque, uq *Is a q-axis reference voltage, KpIs a proportionality coefficient, KiFor the integral coefficient, sgn () is a sign function, r is a design parameter, and r is 0.5 in simulation and experiment.
Further, the modified lunberger observer of step S6 estimates the back electromotive force
Figure BDA00026624696000000110
The estimation formula of (c) is:
Figure BDA0002662469600000021
in the formula, L is an inductor; i.e. iα、iβIs the actual stator current; r is a stator resistor;
Figure BDA0002662469600000022
to estimate the stator current; k1 and k2 are two gain coefficients of the Luenberger observer; u. ofα，uβIs the actual stator voltage; ,
Figure BDA0002662469600000023
to estimate the back emf.
Further, the estimated rotation speed in step S7
Figure BDA0002662469600000024
And rotor position angle
Figure BDA0002662469600000025
The estimation formula of (c) is:
Figure BDA0002662469600000026
Figure BDA0002662469600000027
in the formula: kpIs a proportionality coefficient, KiIn order to be the integral coefficient of the light,
Figure BDA0002662469600000028
in order to estimate the speed of rotation,
Figure BDA0002662469600000029
for rotor position angle, p is a differential operator, λ ═ Ld-Lq)(ωeid-piq)+ωeψfAnd lambda is a variation value related to the rotation speed,
Figure BDA00026624696000000210
to estimate the back emf.
Compared with the prior art, the invention has the following advantages and effects:
the stator voltage is calculated according to the direct-current side voltage and the duty ratio signal of the three-phase inverter, so that the use of a voltage sensor is reduced, and the influence caused by a dead zone effect is reduced; the Luenberger observer is improved, and the robustness to motor parameters and load disturbance is improved; a designed second-order sliding mode regulator is adopted to replace a PI regulator, so that the rapid convergence of errors is improved; different from the traditional method for estimating the back electromotive force, the filter is not needed to filter the estimated back electromotive force, so that the use of the filter is reduced; the phase-locked loop structure is adopted, so that the precision of angle estimation is improved; the problem of direct torque control jitter is improved.
Drawings
Fig. 1 is a control block diagram of a method for controlling a permanent magnet synchronous motor without a position sensor according to an embodiment of the present invention.
Fig. 2 is a schematic structural diagram of a sliding mode flux linkage adjuster and a sliding mode torque adjuster according to an embodiment of the present invention. Fig. 2(a) is a schematic structural view of a sliding mode flux linkage adjuster, and fig. 2(b) is a schematic structural view of a sliding mode torque adjuster.
Fig. 3 is a schematic block diagram of a phase-locked loop provided by an embodiment of the present invention.
Fig. 4 is a comparison graph of simulation results of the rotation speed of the permanent magnet synchronous motor based on the conventional sliding-mode observer position-sensorless control method and the conventional direct torque control method according to the position-sensorless control algorithm provided by the embodiment of the present invention.
Fig. 5 is a comparison graph of simulation results of a rotor position angle of a permanent magnet synchronous motor based on a conventional sliding-mode observer position sensorless control method and a conventional direct torque control method according to a position sensorless control algorithm provided by an embodiment of the present invention.
Detailed Description
The present invention will be described in further detail below by way of examples with reference to the accompanying drawings, which are illustrative of the present invention and are not to be construed as limiting the present invention.
As shown in fig. 1, a method for controlling a permanent magnet synchronous motor without a position sensor for direct torque is characterized in that: comprises the following steps:
s1, estimating the rotating speed of the motor according to the given rotating speed omega through a PI speed regulator
Figure BDA00026624696000000211
Obtaining a reference torque Te by the difference value;
specifically, the calculation process of the reference torque Te in the present invention is described as follows:
Figure BDA00026624696000000212
in the formula, KpIs a proportionality coefficient, KiAs an integral coefficient, ω is a given rotation speed,
Figure BDA00026624696000000213
to estimate the rotational speed, TeReference torque and damping coefficient.
S2, collecting three-phase currents i of a, b and ca，ib，icObtaining the stator current i of the alpha and beta axes through coordinate transformationαAnd iβ(ii) a By duty cycle signal Sa、Sb、ScCalculating three-phase voltage ua，ub，ucObtaining stator voltage u of alpha, beta axis through coordinate transformationαAnd uβ(ii) a The three-phase current ia，ib，icThree-phase voltage ua，ub，ucAnd rotor position angle
Figure BDA0002662469600000031
The stator current i of d and q axes can be obtained by calculationd，iqAnd stator voltage ud，uq。
Specifically, the stator current i of d and q axes in the inventiond，iqAnd stator voltage ud，uqAnd stator currents i of alpha, beta axesα，iβAnd stator voltage uαAnd uβThe calculation process of (2) is as follows:
Figure BDA0002662469600000032
Figure BDA0002662469600000033
Figure BDA0002662469600000034
Figure BDA0002662469600000035
in the formula iα，iβIs an alpha, beta axis stator current, uαAnd uβIs the stator voltage of the alpha, beta axis id，iqStator current of d, q axis, ud，uqThe voltage of the stator, which is the d, q axis,
Figure BDA0002662469600000036
is the rotor position angle.
S3, enabling the d and q axis stator current id，iqAnd electron voltage ud，uqThe actual flux linkage flux and the actual torque Te can be obtained through calculation;
specifically, the calculation process of the actual flux linkage flux and the actual torque Te in the invention is as follows:
Figure BDA0002662469600000037
in the formula, pnIs a logarithm of poles,. psifIs a permanent magnet flux linkage.
S4, inputting the difference value between the given reference flux linkage flux and the actual flux linkage flux into a sliding mode flux linkage regulator, and outputting a d-axis reference voltage u through the sliding mode flux linkage regulatord *(ii) a Inputting the difference value between the reference torque Te and the actual torque Te into a sliding mode torque regulator, and outputting a q-axis reference voltage u through the sliding mode torque regulatorq *；
Specifically, in the present invention, the d-axis reference voltage ud *And q-axis reference voltage uq *The calculation process of (2) is as follows:
defining a sliding mode surface function of the flux linkage as
sflux＝flux*-flux (7)
The basic principle of second-order sliding mode control based on the super-twisting algorithm is utilized, and the expression of the sliding mode flux linkage regulator is shown as
Figure BDA0002662469600000038
Defining a sliding mode surface function of torque as
Figure BDA0002662469600000039
The basic principle of second-order sliding mode control based on the super-twisting algorithm is utilized, and the expression of the sliding mode torque regulator is shown in the specification
Figure BDA00026624696000000310
Thus, the d-axis reference voltage ud *And q-axis reference voltage uq *The calculation process of (2) is as follows:
Figure BDA0002662469600000041
Figure BDA0002662469600000042
in the formula, sfluxThe flux is a sliding mode surface function of the flux linkage, flux is a reference flux linkage, flux is an actual flux linkage, and u is a reference flux linkaged *Is d-axis reference voltage, sTAs a sliding mode surface function of torque, Te is reference torque, Te is actual torque, uq *Is a q-axis reference voltage, KpIs a proportionality coefficient, KiFor the integral coefficient, sgn () is a sign function, r is a design parameter, and r is 0.5 in simulation and experiment.
S5, enabling the d-axis reference voltage ud *Q-axis reference voltage uq *And rotor position angle
Figure BDA0002662469600000043
Obtaining alpha-axis reference voltage u through coordinate transformationα *And beta axis referenceVoltage uβ *The alpha axis reference voltage uα *And a beta-axis reference voltage uβ *Inputting into SVPWM sine pulse width modulation module, outputting duty ratio signal S via SVPWM sine pulse width modulation modulea、Sb、ScThen the duty ratio signal Sa、Sb、ScInputting a three-phase inverter to control the on and off of the three-phase inverter, and realizing the driving of the permanent magnet synchronous motor;
s6, converting the alpha and beta axis stator current iα，iβAnd a, beta axis stator voltage uα，uβSubstituting the Longberger observer into the PMSM to calculate the estimated back electromotive force
Figure BDA0002662469600000044
Specifically, the improved lunberger observer in the present invention is constructed as follows:
on a two-phase static alpha-beta axis coordinate system, the state equation of a traditional lunberg observer is expressed as follows:
Figure BDA0002662469600000045
wherein each variable is defined as x ═ iα iβ Eα Eβ]TIs a state vector, u ═ uα uβ]TFor the input vector, y ═ iα iβ]TFor the output vector, A, B, C are the state space matrix, input matrix, and output matrix coefficients, respectively, defined as follows:
Figure BDA0002662469600000046
Figure BDA0002662469600000047
Figure BDA0002662469600000048
the Lonberg observer is improved to the form
Figure BDA0002662469600000049
In the formula (I), the compound is shown in the specification,
Figure BDA00026624696000000410
is a state vector, u ═ uα uβ]TIn order to input the vector, the vector is input,
Figure BDA00026624696000000411
a, B, C are state space matrix, input matrix and output matrix coefficients, respectively, all of which remain unchanged for the output vector, L is a feedback gain matrix having the form
Figure BDA00026624696000000412
Wherein k1 and k2 are observation gains, and the values and sizes of the gains influence the stability of the system.
Finally, the expression of the modified lunberger observer is:
Figure BDA0002662469600000051
wherein L is an inductor; i.e. iα、iβIs the actual stator current; r is a stator resistor;
Figure BDA0002662469600000052
to estimate the stator current; k1 and k2 are two gain coefficients of the Luenberger observer; u. ofα，uβIs the actual stator voltage;
Figure BDA0002662469600000053
to estimate the back emf.
As can be seen from equation (19), the modified lunberg observer approaches the error to zero using feedback, so that the obtained back emf estimate is more accurate and does not require a filter for filtering.
S7, estimating the back electromotive force
Figure BDA0002662469600000054
Calculation of rotor position angle in phase locked loop
Figure BDA0002662469600000055
And estimating the rotational speed
Figure BDA0002662469600000056
Specifically, the working principle and the stability proving process of the phase-locked loop are as follows:
the functional block diagram of the phase-locked loop is shown in fig. 3, and the input back electromotive force error for constructing the phase-locked loop is:
Figure BDA0002662469600000057
in the formula: λ ═ Ld-Lq)(ωeid-piq)+ωeψf；θeIs the motor rotor position angle; the approximate sign is taken when the angle error is less than 30 °.
The position angle error transfer function of the phase locked loop is:
Figure BDA0002662469600000058
in the formula:
Figure BDA0002662469600000059
is the bandwidth of the PI controller.
When the motor speed is stable, i.e. the motor estimated speed
Figure BDA00026624696000000510
Is a fixed value, then
Figure BDA00026624696000000511
For a linear equation of one degree, the steady state error of the estimated rotor position is:
Figure BDA00026624696000000512
at this time, the phase-locked loop stability is verified.
Thus, rotor position angle
Figure BDA00026624696000000513
And estimating the rotational speed
Figure BDA00026624696000000514
The estimation formula of (c) is:
Figure BDA00026624696000000515
in the formula: kpIs a proportionality coefficient, KiIn order to be the integral coefficient of the light,
Figure BDA00026624696000000516
in order to estimate the speed of rotation,
Figure BDA00026624696000000517
for rotor position angle, p is a differential operator, λ ═ Ld-Lq)(ωeid-piq)+ωeψfAnd lambda is a variation value related to the rotation speed,
Figure BDA0002662469600000061
to estimate the back emf.
According to the control block diagram shown in fig. 1, a permanent magnet synchronous motor position sensorless control system simulation model is built under MATLAB/SIMULINK environment, and motor parameters are selected as follows: rated power600W, rated rotation speed of 600rpm, rated torque of 5 N.m, pole pair number of 4, permanent magnet flux linkage amplitude of 0.175Wb, armature winding resistance of 1.2 omega, alternating-direct axis inductance of 8.5mH and 8.5mH respectively, and rotary inertia of 0.0008 kg.m2. The idling speed is initially given at 300rpm, the mutation at 0.1s is 450rpm, the mutation at 0.2s is 600rpm, and the loading at 0.s is 3 N.m. Under the above conditions, the simulation data based on the traditional sliding-mode observer control method without a position sensor, the traditional direct torque control method and the method of the patent are compared with the actual values, and the simulation result graphs of the rotating speed, the rotor position angle and the rotor position angle error are shown in fig. 4-5. As can be seen from fig. 4, the method of the present invention can effectively reduce the rotational speed buffeting of the conventional sliding mode observer method, has the advantages of fast response, good steady-state performance, and strong robustness of load disturbance, and meanwhile, the method of the present invention has better static effect and speed tracking performance than the conventional direct torque control method; as can be seen from FIG. 5, the tracking of the rotor position angle by the method of the present invention is more accurate than the position-sensor-free control method based on the conventional sliding-mode observer and the conventional direct torque control method.
The above description of the present invention is intended to be illustrative. Various modifications, additions and substitutions for the specific embodiments described may be made by those skilled in the art without departing from the scope of the invention as defined in the accompanying claims.

Claims (4)

1. A permanent magnet synchronous motor position sensorless control method is characterized by comprising the following steps:
s1, estimating the rotating speed of the motor according to the given rotating speed omega through a PI speed regulator
Figure FDA0002662469590000011
Obtaining a reference torque Te by the difference value;
s2, collecting three-phase currents i of a, b and ca，ib，icObtaining the stator current i of the alpha and beta axes through coordinate transformationαAnd iβ(ii) a By passingDuty ratio signal Sa、Sb、ScCalculating three-phase voltage ua，ub，ucObtaining stator voltage u of alpha, beta axis through coordinate transformationαAnd uβ(ii) a The three-phase current ia，ib，icThree-phase voltage ua，ub，ucAnd rotor position angle
Figure FDA0002662469590000012
The stator current i of d and q axes can be obtained by calculationd，iqAnd stator voltage ud，uq；
S3, enabling the d and q axis stator current id，iqAnd electron voltage ud，uqThe actual flux linkage flux and the actual torque Te can be obtained through calculation;
s4, inputting the difference value between the given reference flux linkage flux and the actual flux linkage flux into a sliding mode flux linkage regulator, and outputting a d-axis reference voltage u through the sliding mode flux linkage regulatord *(ii) a Inputting the difference value between the reference torque Te and the actual torque Te into a sliding mode torque regulator, and outputting a q-axis reference voltage u through the sliding mode torque regulatorq *；
S5, enabling the d-axis reference voltage ud *Q-axis reference voltage uq *And rotor position angle
Figure FDA0002662469590000013
Obtaining alpha-axis reference voltage u through coordinate transformationα *And a beta-axis reference voltage uβ *The alpha axis reference voltage uα *And a beta-axis reference voltage uβ *Inputting into SVPWM sine pulse width modulation module, outputting duty ratio signal S via SVPWM sine pulse width modulation modulea、Sb、ScThen the duty ratio signal Sa、Sb、ScInputting a three-phase inverter to control the on and off of the three-phase inverter, and realizing the driving of the permanent magnet synchronous motor;
s6, electrifying the alpha and beta axis statorStream iα，iβAnd a, beta axis stator voltage uα，uβSubstituting improved Lorberg observer to calculate estimated back electromotive force of permanent magnet synchronous motor
Figure FDA0002662469590000014
S7, estimating the back electromotive force
Figure FDA0002662469590000015
Calculation of rotor position angle in phase locked loop
Figure FDA0002662469590000016
And estimating the rotational speed
Figure FDA0002662469590000017
And S8, repeating S1-S7 to realize the closed-loop stable operation of the motor.
2. The sensorless control method of a permanent magnet synchronous motor according to claim 1, wherein the d-axis reference voltage u of step S4d *And q-axis reference voltage uq *The calculation formula of (2) is as follows:
Figure FDA0002662469590000018
Figure FDA0002662469590000019
in the formula, sfluxThe flux is a sliding mode surface function of the flux linkage, flux is a reference flux linkage, flux is an actual flux linkage, and u is a reference flux linkaged *Is d-axis reference voltage, sTAs a sliding mode surface function of torque, Te is reference torque, Te is actual torque, uq *Is a q-axis reference voltage, KpIs a proportionality coefficient, KiAs an integral coefficient, sgn () is a sign function and r is a design parameter.
3. The sensorless control method of a permanent magnet synchronous motor according to claim 1, wherein the modified lunberger observer estimates the back electromotive force at step S6
Figure FDA00026624695900000110
The estimation formula of (c) is:
Figure FDA00026624695900000111
wherein L is an inductor; i.e. iα、iβIs the actual stator current; r is a stator resistor;
Figure FDA00026624695900000112
to estimate the stator current; k1 and k2 are two gain coefficients of the Luenberger observer; u. ofα，uβFor the purpose of the actual stator voltage,
Figure FDA00026624695900000113
to estimate the back emf.
4. The sensorless control method of a permanent magnet synchronous motor according to claim 1, wherein the estimated rotation speed of step S7
Figure FDA00026624695900000114
And rotor position angle
Figure FDA00026624695900000115
The estimation formula of (c) is:
Figure FDA0002662469590000021
Figure FDA0002662469590000022
in the formula: kpIs a proportionality coefficient, KiIn order to be the integral coefficient of the light,
Figure FDA0002662469590000023
in order to estimate the speed of rotation,
Figure FDA0002662469590000024
for rotor position angle, p is a differential operator, λ ═ Ld-Lq)(ωeid-piq)+ωeψfAnd lambda is a variation value related to the rotation speed,
Figure FDA0002662469590000025
to estimate the back emf.
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