CN111987812B - Wireless charging system dynamic tuning method for string compensation topology - Google Patents
Wireless charging system dynamic tuning method for string compensation topology Download PDFInfo
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- CN111987812B CN111987812B CN202010738560.2A CN202010738560A CN111987812B CN 111987812 B CN111987812 B CN 111987812B CN 202010738560 A CN202010738560 A CN 202010738560A CN 111987812 B CN111987812 B CN 111987812B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
- H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
- H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J3/00—Circuit arrangements for ac mains or ac distribution networks
- H02J3/18—Arrangements for adjusting, eliminating or compensating reactive power in networks
- H02J3/1807—Arrangements for adjusting, eliminating or compensating reactive power in networks using series compensators
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
- Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
- Y02E40/30—Reactive power compensation
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Abstract
The invention discloses a dynamic tuning method of a wireless charging system for a series compensation topology, and belongs to the technical field of dynamic tuning of wireless charging systems. Step one, obtaining a zero-phase frequency stabilization criterion of a series compensation topology based on a hardware circuit which is provided with a soft switch controllable capacitor with a symmetrical structure and can realize dynamic tuning; step two, realizing a zero-phase frequency stabilization criterion of the series compensation topology through a phase judgment method and a zero value search method; and step three, adjusting the equivalent capacitance value of the soft switch controllable capacitor according to a zero-phase frequency stabilization criterion, thereby carrying out dynamic tuning. The invention provides a dynamic tuning method of a wireless charging system for a series compensation topology, which realizes frequency stabilization control when a passive element in the series compensation topology has parameter drift. Meanwhile, compared with the existing tuning method, the method reduces the consumption of passive elements and reduces the control complexity.
Description
Technical Field
The invention relates to a dynamic tuning method of a wireless charging system based on a string compensation topology, and belongs to the technical field of dynamic tuning of wireless charging systems.
Background
The compensation topology is used as a key link of the wireless charging system, plays roles in reactive power compensation, system efficiency and output power improvement and system stable operation, and the existing research focuses on composite structure design and performance parameter optimization, so that constant current and constant voltage output is realized, and the influence of offset on system performance is reduced. However, the prior art still has poor research on compensating for the frequency stability of the topology. On one hand, the resonance state is damaged by the parameter drift of a passive element in the compensation topology under the influence of accumulated temperature rise and device aging; on the other hand, the compensation topology tuning error causes the natural frequency drift under the influence of the processing technology and parameter tolerance. Both of these conditions cause system performance degradation and, in severe cases, cause system operation anomalies.
Disclosure of Invention
The invention aims to provide a dynamic tuning method of a wireless charging system for a series (series-series) compensation topology, which is used for solving the problems in the prior art and preventing the performance of the system from being reduced.
A wireless charging system dynamic tuning method facing a string compensation topology comprises the following steps:
step one, obtaining a zero-phase frequency stabilization criterion of a series compensation topology based on a hardware circuit which is provided with a soft switch controllable capacitor with a symmetrical structure and can realize dynamic tuning;
step two, realizing a zero-phase frequency stabilization criterion of the series compensation topology through a phase judgment method and a zero value search method;
and step three, adjusting the equivalent capacitance value of the soft switch controllable capacitor according to a zero-phase frequency stabilization criterion, thereby carrying out dynamic tuning.
Further, in step one, the hardware circuit capable of realizing dynamic tuning is equivalent to a mutual inductance model of a series compensation topology, where the mutual inductance model includes: the transmitting terminal comprises a direct current power supply, a transmitting terminal coil and a transmitting terminal soft switch controllable capacitor which are sequentially connected to form a closed loop, the receiving terminal comprises an equivalent load, a receiving terminal coil and a receiving terminal soft switch controllable capacitor which are sequentially connected to form a closed loop,
definition of L 1 And L 2 Self-inductance values, C, of the transmitter-side coil and the receiver-side coil, respectively 1 And C 2 Compensation capacitance values of the transmitting end coil and the receiving end coil respectively, M is between the transmitting end coil and the receiving end coilMutual inductance value, i 1 And i 2 Resonant currents, R, in the transmitter-side coil and the receiver-side coil, respectively o And Uo are the resistance and voltage values of the equivalent load, u, respectively s Is the output voltage value of the inverter.
Further, the zero-phase frequency stabilization criterion includes a receiving-end frequency stabilization criterion and a transmitting-end frequency stabilization criterion, wherein,
the receiving end frequency stabilization criterion is as follows: l is 2 And C 2 In the presence of parameter drift, the resonant current i in the receiving coil 2 With resonant current i in the transmitting coil 1 The phasor expression between is:
the formula (1) shows that: when the receiver is operating in resonance, i 2 Lead i 1 And a phase difference of gamma 2 Is 90 degrees; when the receiving end works in the state of vibration loss, gamma shown in formula (2) 2 Not 90 degrees, will i 2 Lag by 90 DEG and i 1 The phase difference therebetween is defined as gamma d2 I.e. gamma when the receiver is operating in resonance d2 ＝0，
The transmitting end frequency stabilization criterion is as follows: assuming that the receiving end works in a resonance state and the transmitting end works in the resonance state, the full-bridge inverter outputs a voltage u s And i 1 The phasor relationship between them is expressed by the formula (3),
as can be seen from the formula (3): when the transmitting end works in a resonance state, u s And i 1 In phase, i.e. gamma 1 Is 0 degree; when the transmitting end works in a detuning state, gamma shown in formula (4) 1 Not 0 deg., when considering ZVS soft switch of switch tube in full bridge inverter,correcting gamma 1 Implementation of i 1 With appropriate hysteresis u s And using the obtained signal as the criterion of resonance state of transmitting terminal,
further, the soft-switched controllable capacitor with the symmetrical structure comprises: switch tube S 1 And a switch tube S 2 Diode D 1 Diode D 2 Capacitor C a Capacitor C 1s And a capacitor C 2s Said capacitance C 1s Switch tube S 1 Capacitor C 2s And a switching tube S 2 Are sequentially connected in series, the capacitor C a And said capacitor C 1s And a switching tube S 1 Formed bridge arm and capacitor C 2s And a switching tube S 2 The formed bridge arms are connected in parallel, and the diode D 1 And a switching tube S 1 Antiparallel, the diode D 2 And a switching tube S 2 Antiparallel soft switch controllable capacitor with symmetrical structure has a main circuit current of i s And terminal voltage is u s 。
Further, S 1 Control signal of and i s Is synchronized with the zero crossing point of S 2 The control signal is delayed by half a cycle, thereby realizing the switch tube S 1 And a switching tube S 2 The ZVS soft switch, meanwhile, the symmetrical controllable capacitor adopts two switching tubes to realize symmetrical conduction in positive and negative half periods, thereby the waveform is symmetrical and has no distortion and no DC bias,
let C be 1s ＝C 2s = C, according to switching tube S 1 And a switching tube S 2 The duty ratio D value intervals of the control signals are different, and the working modes of the soft switch controllable capacitor are as follows: d is more than or equal to 0 and less than or equal to 0.25,0.25 and more than D and less than or equal to 0.5 and more than 0.5,
the first condition is as follows: d is more than or equal to 0 and less than or equal to 0.25, and in one period, the switch tube S is switched on and off 1 And a switching tube S 2 And a diode D 1 And a diode D 2 Each conducting for a time of 2D pi, and 6 operating modes corresponding to t 0 ～t 6 In the 6 time periods, the number of the channels is equal to or less than the total number of the channels,
modal 1[t 0 ～t 1 ]: switch tube S 1 Conducting and switching tube S 2 Turn-off, diode D 1 And a diode D 2 All are not working, capacitor C 1s And a capacitor C a Simultaneous charging, capacitor C 2s Terminal voltage u of 2c Keeping the same;
modal 2[t 1 ～t 2 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 And a diode D 2 All are not working, capacitor C 1s And a capacitor C 2s Not charged and capacitor C 1s Terminal voltage u of 1c And u 2c Remaining unchanged, capacitance C a Charging;
modal 3[t 2 ～t 3 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Non-operating, diode D 2 Operation, capacitance C a And a capacitor C 2s Are charged simultaneously and u 1c Keeping the same;
modal 4[t 3 ～t 4 ]: switch tube S 1 Switch off and switch on tube S 2 Conducting, diode D 1 And a diode D 2 All are not working, capacitor C a Charging and capacitor C 2s Discharge, u 1c Keeping the same;
modal 5[t 4 ～t 5 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 And a diode D 2 All are not working, capacitor C a Discharge, u 1c And u 2c Keeping the same;
modal 6[t 5 ～t 6 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Operation, diode D 2 Off-working, capacitor C 1s And a capacitor C a Discharge, u 2c Diode D in mode 3 and mode 6, which remains unchanged 1 And a diode D 2 When conducting, the switch tube S 1 And a switching tube S 2 End voltage of is zero, thereby realizing the switch tube S 1 And a switching tube S 2 The soft switching of the ZVS of (1),
based on the above analysis, the equivalent capacitance C is equal to or greater than 0 and equal to or less than 0.25 eq The expression of (a) is as follows,
and a second condition: d is more than 0.25 and less than or equal to 0.5, and the capacitance C 1s And a capacitor C 2s Cross, diode D, during charging 1 And a diode D 2 The conduction time is changed, and 6 working modes correspond to t 0 ～t 6 In the 6 time periods, the number of the channels is equal to or less than the total number of the channels,
modal 1[t 0 ～t 1 ]: switch tube S 1 Conducting and switching tube S 2 Off, diode D 1 And a diode D 2 All are not working, capacitor C 1s And a capacitor C a Simultaneous charging, u 2c Keeping the same;
modal 2[t 1 ～t 2 ]: switch tube S 1 Conducting and switching tube S 2 Off, diode D 1 Non-operating, diode D 2 Operation, capacitance C 1s Capacitor C a And a capacitor C 2s Charging at the same time;
modal 3[t 2 ～t 3 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Off-working, diode D 2 Operation, capacitor C a And a capacitor C 2s Simultaneous charging, u 1c Keeping the same;
modal 4[t 3 ～t 4 ]: switch tube S 1 Switch off and switch tube S 2 Conducting, diode D 1 And a diode D 2 All are not working, capacitor C a And a capacitor C 2s Simultaneous discharge of u 1c Keeping the original shape;
modal 5[t 4 ～t 5 ]: switch tube S 1 Switch off and switch tube S 2 Conducting, diode D 1 Operating, diode D 2 Off-working, capacitor C 1s Capacitor C a And a capacitor C 2s Discharging at the same time;
mode 6[t 5 ～t 6 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Operating, diode D 2 Off-working, capacitor C 1s And a capacitor C a Simultaneous discharge, diode D in mode 3 and mode 6 1 And a diode D 2 When conducting, the switch tube S 1 And a switching tube S 2 The end voltage of the switching element is zero, so that ZVS soft switching is realized,
based on the above analysis, the adjustable equivalent capacitance C is more than 0.25 and less than or equal to 0.5 eq Is expressed as
And a third situation: d is more than 0.5, and the capacitance C 1s And a capacitor C 2s Charging and discharging period of (2) and capacitor C a The charging and discharging periods of (A) are completely synchronous, and at the moment, the control D can not adjust C eq That is, the controllable capacitance of the soft switch fails, and as described above, the effective regulation interval of D is 0 to 0.5 and C eq The adjustment range is C a ～(C a +2C)。
Further, in the third step, the control circuit of the soft switch controllable capacitor works according to the following principle: the current sensor collects the resonance current flowing through the soft switch controllable capacitor main circuit, and generates two paths of PWM signals with the phase difference of 180 degrees by taking the resonance current as a reference, thereby realizing the switching tube S in the soft switch controllable capacitor 1 And a switching tube S 2 The output signal of the ZVS soft switch and the zero phase difference search circuit controls the duty ratio D of the two paths of PWM signals, thereby adjusting the equivalent capacitance value.
Further, in step three, the working principle of the dynamic tuning method is as follows: i collected by current sensor 1 And i 2 After passing through the signal conditioning circuit and the high-speed A/D converter, the controller calculates i 1 And i 2 And i 1 And u s And searching for zero phase difference by adopting a variable step size disturbance observation method, thereby outputting a control signal of the soft switch controllable capacitor.
The main advantages of the invention are: the invention provides a dynamic tuning method of a wireless charging system for a series (series-series) compensation topology, which realizes frequency stabilization control (namely frequency stability control) when a passive element in the series compensation topology has parameter drift. Meanwhile, compared with the existing tuning method, the method reduces the consumption of passive elements and reduces the control complexity.
Drawings
FIG. 1 is a diagram of a mutual inductance model of a string compensation topology;
FIG. 2 is a diagram of a soft-switched controllable capacitor based on a symmetrical structure;
FIG. 3 is a working waveform of the soft-switching controllable capacitor, wherein, FIG. 3 (a) is the working waveform of the soft-switching controllable capacitor when D is more than or equal to 0 and less than or equal to 0.25; FIG. 3 (b) is the working waveform of the soft-switching controllable capacitor when D is not less than 0.25 and not more than 0.5;
FIG. 4 is a diagram of the phase relationship between two sinusoidal signals;
FIG. 5 is a functional block diagram of a variable step perturbation observation method;
fig. 6 is a flow chart of a dynamic tuning method of a wireless charging system facing a series compensation topology, and fig. 6 (a) is a soft-switch controllable capacitance control circuit; FIG. 6 (b) is a functional block diagram of a dynamic tuning method;
fig. 7 is a diagram for analyzing the resonance state of the receiving coil, in which fig. 7 (a) is a diagram of an operation waveform; FIG. 7 (b) is a graph of LC resonance frequency versus phase difference;
FIG. 8 is a diagram of a resonance state analysis of the transmitting coil;
FIG. 9 is a graph of simulation results for soft-switched controllable capacitance, where FIG. 9 (a) is a graph of duty cycle versus equivalent capacitance value; FIG. 9 (b) is a graph of duty cycle versus operating frequency;
FIG. 10 is a graph of simulation results of a dynamic tuning method, wherein FIG. 10 (a) is a tuning process operating waveform; fig. 10 (b) shows a disturbance observation method.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be obtained by a person skilled in the art without making any creative effort based on the embodiments in the present invention, belong to the protection scope of the present invention.
The mutual inductance model of the series compensation topology of the invention is shown in figure 1. L is 1 And L 2 Self-inductance values of the transmitter coil and receiver coil, C, respectively 1 And C 2 Compensation capacitance values of the transmitting coil and the receiving coil respectively, M is a mutual inductance value between the transmitting coil and the receiving coil, i 1 And i 2 Resonant currents in the transmitter coil and the receiver coil, R, respectively o And u o Equivalent load resistance and voltage values, u s Is the output voltage value of the inverter.
(1) Zero-phase frequency stabilization criterion of series compensation topology
(a) And (5) receiving end frequency stabilization criterion. L is 2 And C 2 In the presence of parameter drift, combining with the law of electromagnetic induction and the figure 1, the resonant current i in the receiving coil 2 With resonant current i in the transmitting coil 1 The expression of phasor between is
The formula (1) shows that: when the receiver is operating in resonance, i 2 Advance i 1 And a phase difference of gamma 2 Is 90 degrees; when the receiving end works in the state of vibration loss, gamma shown in formula (2) 2 Not 90. For subsequent analysis, i is 2 Lag by 90 DEG and i 1 The phase difference therebetween is defined as gamma d2 I.e. gamma when the receiver is operating in resonance d2 ＝0。
(b) And (5) transmitting end frequency stabilization criterion. Assuming that the receiving end works in a resonance state and the transmitting end works in the resonance state, the full-bridge inverter outputs a voltage u s And i 1 The phasor relationship therebetween is expressed by the formula (3). Wherein "&"means an operation of" and between two formulas。
As can be seen from the formula (3): when the transmitting end works in a resonance state, u s And i 1 In phase, i.e. gamma 1 Is 0 degree; when the transmitting end works in a detuning state, gamma shown in formula (4) 1 Is not 0 deg. Correcting gamma when considering ZVS soft switching of switching tube in full bridge inverter 1 Implementation i 1 With appropriate hysteresis u s And the resonance state is taken as the criterion of the resonance state of the transmitting terminal.
(2) Soft switch controllable capacitor based on symmetrical structure
As shown in fig. 2, the controllable capacitance component unit based on the symmetrical structure includes: switch tube S 1 And S 2 And its anti-parallel diode D 1 And D 2 Capacitor C 1s 、C 2s And C a . The main circuit current flowing into the controllable capacitor is i s And terminal voltage is u s 。S 1 Is synchronized with the zero crossing of is and S 2 Is delayed by half a cycle, thereby realizing S 1 And S 1 ZVS soft switching of (1). Meanwhile, the symmetrical controllable capacitor adopts two switching tubes to realize symmetrical conduction in positive and negative half periods, so that the waveform is symmetrical and has no distortion and no direct current bias.
To ensure circuit symmetry and simplify theoretical analysis, assume C 1s ＝C 2s The symmetric controllable capacitance having ZVS characteristic is simply referred to as soft-switched controllable capacitance. According to S 1 And S 2 The duty ratio D value intervals of the control signals are different, and the working modes of the soft switch controllable capacitor are as follows: d is more than or equal to 0 and less than or equal to 0.25,0.25 and more than D and less than or equal to 0.5 and more than 0.5.
The first condition is as follows: d is more than or equal to 0 and less than or equal to 0.25, and the working waveform is shown in figure 3 (a). Within one period, S 1 And S 2 And D 1 And D 2 Each conducting for a time of 2D pi, and 6 operating modes corresponding to t 0 ～t 6 6 time periods.
[1]Modal 1[t 0 ～t 1 ]：S 1 Is on and S 2 Off, D 1 And D 2 All are not working, C 1s And C a Simultaneous charging, C 2s Terminal voltage u of 2c Keeping the original shape;
[2]modal 2[t 1 ～t 2 ]：S 1 And S 2 Are all turned off, D 1 And D 2 All are not working, C 1s And C 2s Not charged and C 1s Terminal voltage u of 1c And u 2c Remains unchanged, C a Charging;
[3]modal 3[t 2 ～t 3 ]：S 1 And S 2 All are turned off, D 1 Not in operation, D 1 Work, C a And C 2s Are charged simultaneously and u 1c Keeping the same;
[4]modal 4[t 3 ～t 4 ]：S 1 Off and S 2 On, D 1 And D 2 All are not working, C a Charging and C 2s Discharge, u 1c Keeping the same;
[5]modal 5[t 4 ～t 5 ]：S 1 And S 2 Are all turned off, D 1 And D 2 All are not working, C a Discharge, u 1c And u 2c Keeping the same;
modal 6[t 5 ～t 6 ]：S 1 And S 2 Are all turned off, D 1 Work, D 2 Not working, C 1s And C a Discharge, u 2c Remain unchanged. D in modalities 3 and 6 1 And D 2 When conducting, S 1 And S 2 Is zero, thereby realizing S 1 And S 2 ZVS soft switching of (1).
Based on the above analysis, the equivalent capacitance C is equal to or greater than 0 and equal to or less than 0.25 eq Is expressed as
And a second condition: d is more than 0.25 and less than or equal to 0.5, and the working waveform is shown in figure 3 (b). C 1s And C 2s Cross-over during charging, D 1 And D 2 The conduction time is changed, and 6 working modes correspond to t 0 ～t 6 6 time periods.
[1]Modal 1[t 0 ～t 1 ]：S 1 Is on and S 2 Off, D 1 And D 2 All are not working, C 1s And C a Simultaneous charging, u 2c Keeping the same;
[2]modal 2[t 1 ～t 2 ]：S 1 Is on and S 2 Off, D 1 Not in operation, D 2 Work, C 1s 、C a And C 2s Charging at the same time;
[3]modal 3[t 2 ～t 3 ]：S 1 And S 2 All are turned off, D 1 Not in operation, D 2 Work, C a And C 2s Simultaneous charging, u 1c Keeping the same;
[4]modal 4[t 3 ～t 4 ]：S 1 Off and S 2 On, D 1 And D 2 All are not working, C a And C 2s Simultaneous discharge of u 1c Keeping the same;
[5]modal 5[t 4 ～t 5 ]：S 1 Off and S 2 On, D 1 Work, D 2 Not working, C 1s 、C a And C 2s Discharging at the same time;
[6]modal 6[t 5 ～t 6 ]：S 1 And S 2 Are all turned off, D 1 Work, D 2 Not working, C 1s And C a And simultaneously discharges. D in modalities 3 and 6 1 And D 2 When conducting, S 1 And S 2 The terminal voltage of (b) is zero, thereby realizing ZVS soft switching.
Based on the above analysis, the adjustable equivalent capacitance C is more than 0.25 and less than or equal to 0.5 eq Is expressed as
And a third situation: d is more than 0.5. C 1s And C 2s Charge and discharge period of (1) and (C) a The charging and discharging periods of (A) are completely synchronous, and at the moment, the control D can not adjust C eq I.e. the soft-switched controllable capacitance fails. In summary, the effective regulation interval of D is 0-0.5 and C eq The adjustment range is C a ～(C a +2C)。
(3) Dynamic tuning method
(a) And (4) phase judgment. The phase relationship between two signals is obtained by adopting the digitized zero-crossing phase discrimination method shown in fig. 4, and the working principle is as follows: the two paths of same-frequency sinusoidal signals are processed and then input to an A/D converter, and a controller performs data processing and phase difference calculation.
In FIG. 4, signal i 1 And i 1p The periods of (a) and (b) are both T and m and n are sampling value serial numbers, i 1p Zero crossing between the (m-1) th and m samples, i 1 Zero crossing between the (n-1) th and n samples. At this time, i 1 And i 1p The phase relationship determination method is expressed as follows: i.e. i 1p When zero-crossing, i 1 <0 corresponds to i 1p Advance i 1 I.e. gamma d >0；i 1 >0 corresponds to i 1p Lag i 1 I.e. gamma d <0。
Suppose i 1 And i 1p Straight line near zero crossing and sampling period T s ，(i 1p_m -i 1p_m-1 )/T s Is the slope of (m-1) point and m point, (i) 1p_m T s )/(i 1p_m -i 1p_m-1 ) Is i 1p The zero crossings of the instantaneous value are spaced from m, whereby the spacing Δ t between two zero crossings is expressed as equation (7). In the same way, i 1 The analysis of (2) is similar. The phase difference γ is derived from fig. 4 and equation (7) d Represented by formula (8).
(b) A zero value search method. As shown in fig. 5, the zero phase difference is searched by using a variable step size disturbance observation method for maximum power tracking in photovoltaic power generation, and the working principle is as follows: determining a disturbance step length according to the absolute value of the slope of each point on the curve, and increasing the tracking speed by adopting a larger step length when the measured value is far away from the maximum power point; when the measured value is close to the maximum power point, the tracking speed is ensured by adopting a smaller step length, so that the tracking effect with high speed and high precision is realized.
(c) Functional block diagram of dynamic tuning method
As shown in fig. 6, the hardware circuit for dynamic tuning mainly includes a soft-switch controllable capacitor and a zero-phase difference search circuit.
The working principle of the control circuit of the soft-switch controllable capacitor shown in fig. 6 (a) is as follows: the current sensor collects resonance current flowing through the soft-switch controllable capacitor main circuit, and two paths of PWM signals with phase difference of 180 degrees are generated by taking the resonance current as a reference, so that ZVS soft switching of two switching tubes in the soft-switch controllable capacitor is realized. And the output signal of the zero phase difference search circuit controls the duty ratio D of the two paths of PWM signals, so that the equivalent capacitance value is adjusted.
The working principle of the dynamic tuning method shown in fig. 6 (b): i collected by current sensor 1 And i 2 After passing through the signal conditioning circuit and the high-speed A/D converter, the controller calculates i 1 And i 2 And i 1 And u s And searching for zero phase difference by adopting a variable step size disturbance observation method, thereby outputting a control signal of the soft switch controllable capacitor.
Specific embodiments of the invention are set forth below:
(1) Zero phase frequency stabilization criterion
Assuming that there is parameter drift in the receiving end inductance or capacitance, the operating waveform is shown in fig. 7 (a) and γ 2 And gamma 2d Receiving end associated LC resonance frequency f 2 The simulation results of the changes are shown in fig. 7 (b).Wherein the waveform 1 is i 1 The waveform 2 is i 2 Waveform 3 is i 2 Waveform i delayed by 90 DEG 2d Waveform 4 is i 1 After zero-crossing comparison, the square wave A has a waveform 5 of i 2d And after zero-crossing comparison, the square wave B is obtained, and the waveform 6 is the square wave AB after the exclusive OR between the A and the B.
As shown in fig. 7, the working states of the receiving end are divided into three types: i.e. i 1 Lead i 2 ，Z 2 Is of capacitive and gamma d2 ＞0；i 1 And i 2 In phase, Z 2 Is resistive and gamma d2 ＝0；i 1 Lag i 2 ，Z 2 Is inductive and gamma d2 Is less than 0. Obviously, theoretical analysis and simulation results show that: equation (1) can be used as a frequency stability criterion for the receiving coil by adopting series compensation.
When the receiving coil works in a resonance state, a simulated waveform between the output voltage and the output current of the full-bridge inverter is as shown in fig. 8, assuming that parameter drift exists in the self-inductance value or the compensation capacitance value of the transmitting coil. Obviously, the operating states are divided into three types: u. u s Lead i 1 ，γ II ＞0；u s And i 1 In phase, gamma II ＝0；u s Lag i 1 ，γ II Is less than 0. Obviously, both theoretical analysis and simulation results show that: equation (3) can be used as a frequency stability criterion for the transmit coil using series compensation.
(2) Embodiments of Soft-switched controllable capacitor
C 1s And C 2s Are all 220nF, and the proportionality coefficient is gamma = C 2 /C 1s The value range of (1) is 2.1-3.1, the inductance value of the series resonance formed by the soft switch controllable capacitor is 53 mu H, and the normalized equivalent capacitance value C eq /C 2 And the simulation results between the operating frequencies f and D are shown in fig. 9. Therefore, the following steps are carried out: adjusting D achieves monotonic and continuous C eq And f, adjusting, wherein different gamma correspond to C in different value ranges eq And f.
(3) Detailed description of the invention
As can be seen from fig. 10 (a): control of C by variable step disturbance observation method 2_eq PWM signal duty cycle D, regulation C 2_eq In-process gamma 2a =0 corresponds to the receiving coil being fully resonant, and when i is 2 Is the largest. As can be seen from fig. 10 (b): output voltage U when receiving coil is completely resonant o And max. Meanwhile, the dynamic tuning process of the transmitting coil is similar to that of fig. 10.
Claims (5)
1. A wireless charging system dynamic tuning method facing a string compensation topology is characterized by comprising the following steps:
step one, obtaining a zero-phase frequency stabilization criterion of a series compensation topology based on a hardware circuit which is provided with a soft switch controllable capacitor with a symmetrical structure and can realize dynamic tuning;
step two, realizing a zero-phase frequency stabilization criterion of the series compensation topology through a phase judgment method and a zero value search method;
step three, adjusting the equivalent capacitance value of the soft switch controllable capacitor according to a zero-phase frequency stabilization criterion, thereby carrying out dynamic tuning;
in step one, the hardware circuit capable of realizing dynamic tuning is equivalent to a mutual inductance model of a series compensation topology, and the mutual inductance model comprises: the transmitting terminal comprises a direct current power supply, a transmitting terminal coil and a transmitting terminal soft switch controllable capacitor which are sequentially connected to form a closed loop, the receiving terminal comprises an equivalent load, a receiving terminal coil and a receiving terminal soft switch controllable capacitor which are sequentially connected to form a closed loop,
definition of L 1 And L 2 Self-inductance values, C, of the transmitting-side coil and the receiving-side coil, respectively 1 And C 2 Compensation capacitance values of the transmitting end coil and the receiving end coil respectively, M is a mutual inductance value between the transmitting end coil and the receiving end coil, i 1 And i 2 Resonant currents, R, in the transmitting-side coil and the receiving-side coil, respectively o And Uo are the resistance and voltage values of the equivalent load, u, respectively s Is the output voltage value of the inverter;
the zero-phase frequency stabilization criterion includes a receiving end frequency stabilization criterion and a transmitting end frequency stabilization criterion, wherein,
the receiving end frequency stabilization criterion is as follows: l is 2 And C 2 In the presence of parameter drift, the resonant current i in the receiving coil 2 With resonant current i in the transmitter coil 1 The phasor expression between is:
according to the formula (1): when the receiver is operating in resonance, i 2 Lead i 1 And a phase difference of gamma 2 Is 90 degrees; when the receiving end works in a non-oscillation state, gamma is shown in formula (2) 2 Not 90 degrees, will i 2 Lags by 90 DEG and i 1 The phase difference therebetween is defined as gamma d2 I.e. gamma when the receiver is operating in resonance d2 ＝0，
The transmitting end frequency stabilization criterion is as follows: assuming that the receiving end works in a resonance state and the transmitting end works in the resonance state, the full-bridge inverter outputs a voltage u s And i 1 The phasor relationship between them is expressed by the formula (3),
as can be seen from the formula (3): when the transmitting end works in a resonance state, u s And i 1 In phase, i.e. gamma 1 Is 0 degree; when the transmitting end works in a detuning state, gamma shown in formula (4) 1 Not at 0 deg., correcting gamma when considering ZVS soft switch of switch tube in full bridge inverter 1 Implementation of i 1 Lag u of s And using the obtained signal as the criterion of resonance state of transmitting terminal,
2. the dynamic tuning method for the wireless charging system oriented to the string compensation topology according to claim 1, wherein the soft-switched controllable capacitor with the symmetric structure comprises: switch tube S 1 Switch tube S 2 Diode D 1 Diode D 2 Capacitor C a Capacitor C 1s And a capacitor C 2s Said capacitor C 1s And a switching tube S 1 In series, said capacitor C 2s And a switching tube S 2 In series, said capacitor C a And said capacitor C 1s And a switching tube S 1 Formed bridge arm and capacitor C 2s And a switching tube S 2 The formed bridge arms are connected in parallel, and the diode D 1 And a switching tube S 1 Antiparallel, the diode D 2 And a switching tube S 2 The main circuit current of the soft switch controllable capacitor with a symmetrical structure is i s And terminal voltage is u s 。
3. The dynamic tuning method of the wireless charging system for the string compensation topology as recited in claim 2, wherein S is 1 Control signal of and i s Is synchronized with the zero crossing point of S 2 The control signal is delayed by half a cycle, thereby realizing the switch tube S 1 And a switching tube S 2 The ZVS soft switch adopts two switch tubes to realize symmetrical conduction in positive and negative half periods, so that the waveform is symmetrical and has no distortion and no DC bias,
suppose C 1s ＝C 2s = C, according to the switching tube S 1 And a switching tube S 2 The duty ratio D value intervals of the control signals are different, and the working modes of the soft switch controllable capacitor are as follows: d is more than or equal to 0 and less than or equal to 0.25,0.25 and more than D and less than or equal to 0.5 and more than 0.5,
the first condition is as follows: d is more than or equal to 0 and less than or equal to 0.25, and in one period, the switch tube S is switched on and off 1 And a switching tube S 2 And a diode D 1 And a diode D 2 Each conducting2D pi time, and 6 operating modes correspond to t 0 ～t 6 In the middle of the 6 time periods,
modal 1[t 0 ～t 1 ]: switch tube S 1 Conducting and switching tube S 2 Turn-off, diode D 1 And a diode D 2 All are not working, capacitor C 1s And a capacitor C a Simultaneous charging, capacitor C 2s Terminal voltage u of 2c Keeping the same;
modal 2[t 1 ～t 2 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 And a diode D 2 All are not working, capacitor C 1s And a capacitor C 2s Not charged and capacitor C 1s Terminal voltage u of 1c And C 2s Terminal voltage u of 2c Remaining unchanged, capacitance C a Charging;
modal 3[t 2 ～t 3 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Non-operating, diode D 2 Operation, capacitance C a And a capacitor C 2s Are charged simultaneously and u 1c Keeping the same;
modal 4[t 3 ～t 4 ]: switch tube S 1 Switch off and switch tube S 2 Conducting, diode D 1 And a diode D 2 All are not working, capacitor C a Charging and capacitor C 2s Discharge, u 1c Keeping the same;
modal 5[t 4 ～t 5 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 And a diode D 2 All are not working, capacitor C a Discharge, u 1c And u 2c Keeping the same;
modal 6[t 5 ～t 6 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Operating, diode D 2 Off-working, capacitor C 1s And a capacitor C a Discharge, u 2c Diode D in mode 3 and mode 6, which remains unchanged 1 And a diode D 2 When conducting, the switch tube S 1 And a switching tube S 2 End voltage of is zero, thereby realizing the switch tube S 1 And a switching tube S 2 The soft switching of the ZVS of (1),
based on the above analysis, the equivalent capacitance C is equal to or greater than 0 and equal to or less than 0.25 eq The expression of (a) is as follows,
and a second condition: d is more than 0.25 and less than or equal to 0.5, and the capacitance C 1s And a capacitor C 2s Cross, diode D, during charging 1 And a diode D 2 The conduction time is changed, and 6 working modes correspond to t 0 ～t 6 In the middle of the 6 time periods,
modal 1[t 0 ～t 1 ]: switch tube S 1 Conducting and switching tube S 2 Turn-off, diode D 1 And a diode D 2 All are not working, capacitor C 1s And a capacitor C a Simultaneous charging, u 2c Keeping the same;
modal 2[t 1 ～t 2 ]: switch tube S 1 Conducting and switching tube S 2 Turn-off, diode D 1 Non-operating, diode D 2 Operation, capacitance C 1s Capacitor C a And a capacitor C 2s Charging at the same time;
modal 3[t 2 ～t 3 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Non-operating, diode D 2 Operation, capacitance C a And a capacitor C 2s Simultaneous charging, u 1c Keeping the original shape;
modal 4[t 3 ～t 4 ]: switch tube S 1 Switch off and switch tube S 2 Conducting, diode D 1 And a diode D 2 All are not working, capacitor C a And a capacitor C 2s Simultaneous discharge of u 1c Keeping the same;
modal 5[t 4 ～t 5 ]: switch tube S 1 Switch off and switch tube S 2 Conducting, diode D 1 Operating, diode D 2 Off-working, capacitor C 1s Capacitor C a And a capacitor C 2s Discharging at the same time;
modal 6[t 5 ～t 6 ]: switch tube S 1 And a switching tube S 2 Are all turned off, diode D 1 Operating, diode D 2 Off-working, capacitor C 1s And a capacitor C a Simultaneous discharge, diode D in mode 3 and mode 6 1 And a diode D 2 When conducting, the switch tube S 1 And a switching tube S 2 The end voltage of the switching element is zero, so that ZVS soft switching is realized,
based on the above analysis, the adjustable equivalent capacitance C is more than 0.25 and less than or equal to 0.5 eq Is expressed as
Case three: d is more than 0.5, and the capacitance C 1s And a capacitor C 2s Charge-discharge period and capacitor C a The charging and discharging periods of (A) are completely synchronous, and at the moment, the control D can not adjust C eq That is, the controllable capacitance of the soft switch fails, and as described above, the effective regulation interval of D is 0 to 0.5 and C eq The adjustment range is C a ～(C a +2C)。
4. The dynamic tuning method of the wireless charging system oriented to the series compensation topology as claimed in claim 3, wherein in step three, the operation principle of the control circuit of the soft switch controllable capacitor is as follows: the current sensor collects the resonance current flowing through the soft switch controllable capacitor main circuit, and generates two paths of PWM signals with the phase difference of 180 degrees by taking the resonance current as a reference, thereby realizing the switching tube S in the soft switch controllable capacitor 1 And a switching tube S 2 The output signal of the ZVS soft switch and the zero phase difference search circuit controls the duty ratio D of the two paths of PWM signals, thereby adjusting the equivalent capacitance value.
5. The method of claim 3, wherein in step three, the wireless charging system is dynamically tunedThe working principle of the harmonic method is as follows: i collected by current sensor 1 And i 2 After passing through the signal conditioning circuit and the high-speed A/D converter, the controller calculates i 1 And i 2 And i 1 And u s And searching for zero phase difference by adopting a variable step size disturbance observation method, thereby outputting a control signal of the soft switch controllable capacitor.
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