CN111953360A

CN111953360A - Signal transmission device and correction method

Info

Publication number: CN111953360A
Application number: CN201910408882.8A
Authority: CN
Inventors: 王文山; 张元硕
Original assignee: Realtek Semiconductor Corp
Current assignee: Realtek Semiconductor Corp
Priority date: 2019-05-16
Filing date: 2019-05-16
Publication date: 2020-11-17

Abstract

The disclosure relates to a signal transmission apparatus and a correction method. The signal transmission apparatus includes a transmitter including first oscillator circuitry, signal processing circuitry and correction circuitry, and second oscillator circuitry. The first oscillator circuitry outputs a first oscillator signal. The signal processing circuit system mixes a plurality of correction signals according to the first oscillating signal to transmit a first output signal. Correction circuitry detects the power of the first output signal to generate a plurality of coefficients and generates the correction signal based on the coefficients, an in-phase data signal and a quadrature data signal. The second oscillator circuit system is disposed adjacent to the transmitter and outputs a second oscillating signal. Wherein the calibration signal is configured to reduce a pull of both the first output signal and the second oscillator signal to the first oscillator circuitry.

Description

Signal transmission device and correction method

Technical Field

The present disclosure relates to a signal transmission apparatus, and more particularly, to a signal transmission apparatus having a plurality of transmitters and a calibration method thereof for eliminating a pulling effect.

Background

When the electronic device is equipped with multiple transmitters, multiple rf signals generated by the multiple transmitters may be coupled back to the oscillator in each transmitter, so that the oscillator signal generated by the oscillator generates a phase error. This phenomenon is generally referred to as a pulling effect.

In some techniques, the pulling effect of the own oscillator is corrected for only a single transmitter, and the correction mechanism is usually placed after the mixer. Thus, the calibration mechanism requires a higher bandwidth, which increases the cost and design complexity of the transmitter. In other techniques, a correction circuit to eliminate the pulling effect is provided in the phase locked loop. Thus, unwanted phase noise may be introduced, reducing the overall performance of the transmitter.

Disclosure of Invention

One embodiment of the present disclosure provides a signal transmission apparatus, which includes a first transmitter and a second oscillator circuitry. The first transmitter includes first oscillator circuitry, signal processing circuitry, and correction circuitry. The first oscillator circuitry outputs a first oscillator signal. The signal processing circuit system mixes a plurality of correction signals according to the first oscillating signal to transmit a first output signal. The correction circuitry detects power of the first output signal to generate a plurality of coefficients and generates the correction signal based on the coefficients, an in-phase data signal and a quadrature data signal. The second oscillator circuit system is disposed adjacent to the first transmitter and is configured to output a second oscillating signal. Wherein the calibration signal is configured to reduce a pull of both the first output signal and the second oscillator signal to the first oscillator circuitry.

One embodiment of the present disclosure provides a calibration method, which includes the following operations: mixing, by a first transmitter, a plurality of calibration signals according to a first oscillation signal to transmit a first output signal, wherein the first oscillation signal is provided by a first oscillator circuitry of the first transmitter; detecting the power of the first output signal to generate a plurality of coefficients; and generating the calibration signal based on the coefficient, an in-phase data signal and a quadrature data signal, wherein the calibration signal is configured to reduce a pulling of both the first output signal and a second oscillator signal to the first oscillator circuit, and the second oscillator signal is provided by a second oscillator circuitry disposed adjacent to the first transmitter.

In summary, the signal transmission apparatus and the calibration method provided by the embodiments of the present disclosure can generate a plurality of calibration signals to simultaneously eliminate errors generated by the transmitter itself and an external circuit adjacent to the transmitter due to the pulling effect. Thus, the performance of the plurality of transmitters in the signal transmission device applied to the multi-channel transmission application can be improved.

Drawings

Fig. 1 is a schematic illustration of a signal transmission device according to some embodiments of the present disclosure;

FIG. 2A is a schematic diagram of the emitter of FIG. 1, constructed in accordance with some embodiments of the present disclosure;

FIG. 2B is a diagram of a mathematical equivalent model in the time domain when a transmitter generates a pulling effect;

FIG. 2C is a schematic diagram of a mathematical equivalent model of a correction matrix for suppressing the effects of traction, according to some embodiments of the present disclosure;

FIG. 3 is a schematic diagram of the transmitter of FIG. 1 shown in accordance with some embodiments of the present disclosure;

FIG. 4 is a flow chart of a method of adjusting coefficients shown in accordance with some embodiments of the present disclosure;

FIG. 5 is a schematic diagram of the phase correction circuit of FIG. 3, constructed in accordance with some embodiments of the present disclosure; and

fig. 6 is a schematic diagram of the phase correction circuit of fig. 3, drawn in accordance with some embodiments of the present disclosure.

Description of the symbols

100: signal transmission means 120, 130: emitter

124: oscillator circuitry 134: oscillator circuit system

122: signal processing circuitry 124: signal processing circuit system

201. 211: digital-to-analog converters I (t), I [ n ]: in-phase data signal

202. 212, and (3): low-pass filter

206: voltage-controlled oscillators Q (t), Q [ n ]: orthogonal data signal

207: local oscillator signal generators C1, C2: coefficient of performance

203. 213: mixers I '(t), Q' (t): correcting signal

215: adders I '[ n ], Q' [ n ]: correcting signal

204: the power amplifier 320: correction circuit system

205: antenna S_DC1: digital code

S_DBB: the baseband signal 322: feedback control circuit

S_ABB: analog signals 324: calculation circuit

S_VCO1、S_VCO2: oscillating signal 324A: phase correction circuit

S_LO、S_ILO: local oscillation signal 324B: correction signal generating circuit

S_QLO: local oscillation signal 322A: attenuator

S_VM: modulated signal 322B: self-mixer

S_VM1～S_VM3: modulation signal 322C: amplifier with a high-frequency amplifier

S_VO1: output signal 322D: analog-to-digital converter

S_VO1’、S_VO2: output signal 322E: correction circuit

200A: the correction matrix 323: signal power detector

θ (t): phase error 325: adjusting circuit

ω_LO(t): angular frequency S_VD、S_VD’: detecting the signal

: preliminary phase correction signal S_VA: adjusting signals

I²[n]、Q²[n]: the calculated value is 400: method of producing a composite material

I [ n ] Q [ n ]: calculation values S401 to S409: operation of

I²[n]-Q²[n]: the calculation values are 501-505: multiplier and method for generating a digital signal

C1*(I²[n]-Q²[n]): the calculated value 506: subtracter

2C2*(I²[n]-Q²[n]): calculated value 507: adder

601. 602: finite impulse filters θ 1(t), θ 2 (t): phase error

Detailed Description

The following embodiments are described in detail with reference to the accompanying drawings, but the embodiments are not provided to limit the scope of the disclosure, and the description of the structural operation is not intended to limit the execution sequence thereof, and any structure resulting from the rearrangement of elements to produce an apparatus with equivalent technical effect is within the scope of the disclosure. In addition, the drawings are for illustrative purposes only and are not drawn to scale. For ease of understanding, the same or similar elements will be described with the same reference numerals in the following description.

Further, as used herein, the term "couple" or "connect" refers to two or more elements being in direct physical or electrical contact with each other, or in indirect physical or electrical contact with each other, or to two or more elements operating or acting together.

As used herein, the term "circuit system" may generally refer to a single system comprising one or more circuits (circuits). The term "circuit" may generally refer to an object connected in some manner by one or more transistors and/or one or more active and passive components to process a signal.

As used herein, "signal A (t)" refers to a continuous signal in analog form, "signal A [ n ]" refers to a discrete signal in digital form, and corresponds to signal A (t). For example, the signal A [ n ] can be converted to the corresponding signal A (t) by a digital-to-analog converter. Similarly, in other embodiments, the signal a (t) may be converted to the corresponding signal a [ n ] by an analog-to-digital converter.

For ease of understanding, like elements in the various figures will be designated with the same reference numerals.

Fig. 1 is a schematic diagram of a signal transmission device 100 according to some embodiments of the present disclosure. The signal transmission device 100 includes the

transmitters

120 and 130, and thus can be applied to related applications of multi-channel transmission data. For ease of illustration, fig. 1 shows only two sets of emitters, but is not so limited. In other embodiments, the number of transmitters of the signal transmission device 100 may be greater than 2.

The transmitter 120 includes signal processing circuitry 122 and oscillator circuitry 124. The oscillator circuitry 124 generates a signal having a frequency f_VCO1Is oscillating signal S_VCO1To signal processing circuitry 122. The signal processing circuit 122 is based on the oscillation signal S_VCO1Processing a baseband signal S_DBBAnd emits an output signal S_VO1. The transmitter 130 includes signal processing circuitry 132 and oscillator circuitry 134. The oscillator circuitry 134 generates a signal having a frequency f_VCO2Is oscillating signal S_VCO2To signal processing circuitry 132. The signal processing circuit 132 is based on the oscillation signal S_VCO2Processing a baseband signal S_DBBAnd emits an output signal S_VO2。

The baseband signals S processed by the

transmitters

120 and 130 may depend on different applications_DBBMay be the same or different, and the frequency f_VCO1And frequency f_VCO2May be the same or different. In some embodiments, the oscillating signal S is generated without the signal processing circuitry 132_VCO2May be communicated to signal processing circuitry 122. In this case, the signal processing circuit 122 can selectively generate the oscillation signal S_VCO1Or oscillating signal S_VCO2Transmitting an output signal S_VO1. Thus, when the frequency f is_VCO1And frequency f_VCO2In the meantime, the transmitter 120 may be adapted for dual mode applications.

In some embodiments, the

emitters

120 and 130 are disposed adjacent to each other. For example, the transmitter 120 and the transmitter 130 (and/or the oscillator circuitry 134) are integrally disposed within a single die. Alternatively, the transmitter 120 and the transmitter 130 (and/or the oscillator circuitry 134) are disposed on the first die and the second die, respectively, but both the first die and the second die are packaged in a single package (i.e., the signal transmission device 100 may be implemented by a plurality of dies and packaged as a single chip or integrated circuit). In practical applications, the transmitter 120 and the transmitter 130 may interfere with each otherTo let the output signal S_VO1And/or output signal S_VO2An error occurs. For example, the output signal S_VO1Coupled to the oscillator circuitry 124 (i.e., pulling effect occurs inside the transmitter 120) and oscillates the signal S_VCO2And/or output signal S_VCO1Coupled to the oscillator circuitry 124 (i.e., the transmitter 130 pulls the transmitter 120), resulting in the output signal S of the transmitter 120_VO1Causing errors. In some embodiments, as described below, the transmitter 120 (and/or the transmitter 130) may also be provided with calibration circuitry 320 to ameliorate the effects of the pulling effect from multiple sources.

Fig. 2A is a schematic diagram of the emitter 120 of fig. 1, drawn in accordance with some embodiments of the present disclosure. The signal processing circuitry 122 includes a digital-to-analog converter 201, a low pass filter 202, a mixer 203, a power amplifier 204, and an antenna 205. The oscillator circuitry 124 includes a voltage controlled oscillator 206 and a local oscillator signal generator 207.

The digital-to-analog converter 201 is based on the base frequency signal S_DBBGenerating an analog signal S_ABB. The low-pass filter 202 removes the analog signal S_ABBThe image noise on the image. The voltage controlled oscillator 206 generates a signal having a frequency f_VCO1Is oscillating signal S_VCO1. The local oscillator signal generator 207 may be coupled to the oscillator signal S_VCO1Frequency division is performed to generate a signal having a local frequency f_LOLocal oscillation signal S_LO. The mixer 203 can be based on the local oscillation signal S_LO1For the filtered analog signal S_ABBPerforming frequency raising to output a modulated signal S_VM. The power amplifier 204 amplifies the modulated signal S_VMTo generate an output signal S_VO1. The antenna 205 emits an output signal S_VO1. Output signal S_VO1Can be expressed in time domain as the following formula (1):

S_VO1＝GA_BB(t)cos(ω_LOt+θ_BB(t)+σ)…(1)。

in equation (1), G is the overall gain of the transmitter 120, S_ABB(t) is an analog signal S_ABBAmplitude of (a), ω_LOTo correspond to a local frequency f_LOAngular frequency of (theta)_BB(t) is an analog signal S_ABBAnd sigma is the base frequency signal S_DBBThe additional phase introduced when passing through the transmitter 120.

When the pulling effect described in fig. 1 occurs, a signal S is output_VO1The correction can be given by the following formula (2):

S_VO1＝GA_BB(t)cos(ω_LOt+θ_BB(t)+σ+θ₁(t)+θ₂(t))

＝GA_BB(t)cos(ω_LOt+θ_BB(t)+σ+θ(t))…(2)。

wherein, theta₁(t) is the output signal S_VO1Phase error, theta, introduced by the resulting drag effect₂(t) is another transmitter 130 (e.g., from an oscillating signal SV)_CO2And/or output signal S_VO2Coupling) and/or phase error introduced by drag effects, so that the total phase error θ (t) is θ₁(t) and θ₂(t) sum of (d). If the extra phase σ in equation (2) is 0 and the gain G of the transmitter 120 is 1, the output signal S can be obtained_VO1Further simplified to the following formula (3):

S_VO1＝A_BB(t)cos(ω_LOt+θ_BB(t)+θ(t))…(3)。

expansion (3) results in the following equation (4):

S_VO1＝[A_BB(t)cos(θ_BB(t))cos(θ(t))cos(ω_LOt)]

+[A_BB(t)sin(θ_BB(t))cos(θ(t))(-sin(ω_LOt)]

+[A_BB(t)cos(θ_BB(t))sin(θ(t))(-sin(ω_LOt)]

-[A_BB(t)sin(θ_BB(t))sin(θ(t))(cos(ω_LOt)]

＝[I(t)cos(θ(t))cos(ω_LOt)+Q(t)cos(θ(t))(-sin(ω_LOt))]

+[I(t)sin(θ(t))(-sin(ω_LOt)-Q(t)sin(θ(t))(cos(ω_LOt))]…(4)。

wherein, I (t) ═ S_ABB(t)cos(θ_BB(t)), and I (t)) To correspond to the base frequency signal S_DBBIn-phase (in-phase) data signal. Q (t) ═ S_ABB(t)sin(θ_BB(t)), and Q (t) corresponds to the baseband signal S_DBBQuadrature (quadrature) data signals.

Fig. 2B is a diagram of a mathematical equivalent model in the time domain when the transmitter 120 generates a pulling effect. Fig. 2C is a diagram illustrating a mathematical equivalent model of a correction matrix for suppressing the effects of traction according to some embodiments of the present disclosure.

In some embodiments, the mixing of the analog signal S_ABBFront, analog signal S_ABBMay be corrected by the correction matrix 200A of fig. 2C to eliminate the total phase error θ (t). According to fig. 2B and 2C, the in-phase data signal i (t) and the quadrature data signal q (t) satisfy the following formula (5):

analog signal S by correction matrix 200A according to equation (5)_ABBBy doing this in advance, the total phase error θ (t) can be eliminated.

Rewriting formula (5) as a complex function to formula (6):

(t)+jQ'(t)＝[I(t)+0(t)]e^[-jθ(t)]＝[I(t)+Q(t)][α(t)+jβ(t)]…(6)。

here, I '(t) + jQ' (t) is a correction signal calculated by the correction matrix 200A, the phase correction signal α (t) is cos (θ (t)), and the phase correction signal β (t) is-sin (θ (t)). Equivalently, the analog signal S is corrected by the correction matrix 200A_ABBPerforming calculation in advance to generate a pre-phase correction signal

And is

Thus, when the correction signal I '(t) + jQ' (t) is mixed by the mixer 203, the phase correction signal is preliminarily corrected

May cancel out the phase error θ (t).

Referring to the contents of the document (Pulling simulation in Wireless Transmitter IEEE JSSC vol.49, No.9, Sep.2014.) and FIG. 3, the phase error θ (t) and the base frequency signal S_DBBIn connection with, wherein the base frequency signal S_DBBCorresponding analog signal S_ABBCan be formed by the superposition of an in-phase data signal I (t) and a quadrature data signal Q (t), i.e. S_ABBI (t) + jq (t). According to FIG. 3 and equation (6) of the above document, the phase correction signal is preliminarily corrected

After coordinate transformation, it can be expressed as the following formula (7):

therefore, the coefficients C1 and C2 in the above formula (7) can be used to generate the pre-phase correction signal

Due to the fact that

Preliminary phase correction signal

After generation, the correction matrix 200A may generate the correction signal I '(t) + jQ' (t) to the transmitter 120 accordingly to eliminate the effect of the pulling effect.

The following embodiments will be presented in terms of either the time domain or the frequency domain. Fig. 3 is a schematic diagram of the transmitter 120 of fig. 1 shown in accordance with some embodiments of the present disclosure.

Transmitter 120 includes correction circuitry 320 and signal processing circuitry 122 also includes low pass filter 212, mixer 213, summer 215, power amplifier 204, and antenna 205.

The digital-to-analog converter 201 is based on the correction signal I' n]A correction signal I' (t) is generated. Low pass filter 202 removes correctionsThe image on the signal I' (t) caused by the digital-to-analog conversion. The mixer 203 is based on the local oscillation signal S_ILOUp-converting the filtered correction signal I' (t) to output a modulated signal S_VM1。

The digital-to-analog converter 211 generates a correction signal Q' n]A correction signal Q' (t) is generated. The low-pass filter 212 removes the image on the correction signal Q' (t). The mixer 213 is based on the local oscillation signal S_QLOUp-converting the filtered correction signal Q' (t) to output a modulated signal S_VM2. Adder 215 adds modulated signal S_VM1And a modulated signal S_VM2To generate a modulated signal S_VM3. The power amplifier 204 amplifies the modulated signal S_VM3To generate an output signal S_VO1And transmits the output signal S via the antenna 205_VO1。

In some embodiments, the calibration circuitry 320 includes a feedback control circuit 322 and a calculation circuit 324. Feedback control circuit 322 analyzes output signal S_VO1To generate a digital code S_DC1And according to the digital code S_DC1Yielding coefficients C1-C2. The calculation circuit 324 calculates the in-phase data signal I [ n ] according to the coefficients C1-C2]And quadrature data signal Q [ n ]]Generating a correction signal I' [ n ]]And a correction signal Q' [ n ]]To signal processing circuitry 122.

The feedback control circuit 322 includes an attenuator 322A, a self-mixer 322B, an amplifier 322C, an analog-to-digital converter 322D, and a correction circuit 322E.

Attenuator 322A reduces output signal S_VO1To generate an output signal S_VO1To self mixer 322B. In some embodiments, the attenuator 322A may be implemented by at least one coupling capacitor. From mixer 322B based on output signal S_VO1’Modulating the output signal S_VO1’To generate a detection signal S_VD. In some embodiments, self-mixer 322B may be implemented with cross-coupled transistor pairs.

In some embodiments, if the gain of the power amplifier 204 is low, the output signal S is outputted_VO1May be input directly to the self-mixer 322B. In this example, the output signal S is output from the mixer 322B_VO1Self-mixing (e.g. ofFor the output signal S_VO1Performs a squaring operation) to generate a detection signal S_VD。

The amplifier 322C amplifies the detection signal S_VDTo generate a detection signal S_VD’. Amplifier 322C may be an amplifier circuit with fixed gain or adjustable gain. The analog-to-digital converter 322D is based on the detection signal S_VD’Generating a digital code S_DC1. The correction circuit 322E is based on the digital code S_DC1Yielding coefficients C1-C2.

Referring to FIG. 2A and FIG. 8 and its contents of the related documents, the output signal S of the transmitter 120_VO1Has a frequency of f_LO+f_MWherein f is_MFor an analogue signal S_ABBFor example, the frequency of the quadrature data signal q (t) or the in-phase data signal i (t). When affected by the pulling effect, two dominant noises appear at the output of the transmitter 120, each with a frequency f_LO+3f_MAnd f_LO-f_M. In other words, the signal S is output_VO1Mainly comprising a frequency of f_LO+f_M、f_LO+3f_MAnd f_LO-f_MA plurality of signals. After frequency mixing (equivalent to squaring), the signal S is detected_VDAt least having a frequency of 2f_MAnd 4f_MA plurality of signal components. Detecting signal S_VDIs approximately twice or four times the frequency of the quadrature data signal q (t) or the in-phase data signal i (t). Accordingly, the digital code S_DC1At least having a frequency of 2f_MAnd 4f_MA plurality of signal components. Thus, having a frequency 2f_MAnd 4f_MMay reflect the effects of the pulling effect.

In some embodiments, the calibration circuit 322E includes a signal power detector 323 and an adjustment circuit 325. The signal power detector 323 detects the digital code S_DC1Has a frequency of 2f_MOr 4f_MTo generate the adjustment signal S_VA. The adjusting circuit 325 is based on the adjusting signal S_VAAdjusting the coefficient C1-C2. In other embodiments, with respect to having frequency 2f_MHas a signal component of frequency 4f_MThe signal components of (a) are higher in frequency and tend to undergo greater attenuation when transmitted. Therefore, the signal power detector 323 can detect only the digital code S_DC1Has a frequency of 2f_MTo generate the adjustment signal S_VA。

By the above feedback control method, the coefficients C1-C2 can be adjusted to decrease the output signal S_VO1Has a frequency f_LO+3f_MOr f_LO-f_MThe power of the plurality of noise signal components. In this manner, the transmitter 120 will be less affected by the pulling effect.

The following example to detect a signal having a frequency 2f_MThe power of the signal component(s) in (c) is illustrated as an example, but the disclosure is not limited thereto. In other embodiments, the related circuit configuration may be analogized, modified or replaced according to the configuration of each embodiment to detect the frequency 4f_MThe power of the signal component of (a).

FIG. 4 is a flow chart illustrating a method 400 of adjusting coefficients C1-C2 according to some embodiments of the present disclosure. In some embodiments, the adjusting circuit 325 may be implemented by a digital signal processing circuit to perform the method 400 of FIG. 4 to generate the coefficients C1-C2. The digital signal processing circuit can adjust the signal S_VAIs obtained with a frequency 2f_MOr 4f_MThe power of the signal component of (a).

In some embodiments, the frequency 2f is detected by comparing two consecutive previous detections_MOr 4f_MThe power of the signal components of (2) can be adjusted by the coefficients C1-C2 in turn. In FIG. 4, E (n) is a signal having a frequency of 2f_MOr 4f_MN is the number of adjustments. In operation S401, the adjustment directions of the coefficients C1-C2 are all increased, i.e., SIGN _ C1 and SIGN _ C2 are set to 1, wherein SIGN _ C1 and SIGN _ C2 respectively represent the adjustment directions of the coefficients C1 and C2. In operation S402, it is confirmed that the first three times of measurements have a frequency of 2f_MOr 4f_MIs lower than the power of the signal component of (i.e. E (n-3)) measured in the first two times with a frequency of 2f_MOr 4f_MThe power of the signal component of (i.e., E (n-2)). If so,operation S403 is performed. Otherwise, operation S404 is performed.

In operation S403, the adjustment direction of the coefficient C1 is reset to decrease, i.e., SIGN _ C1 is set to-SIGN _ C1. As mentioned above, the coefficients C1-C2 are adjusted to reduce the output signal S_VO1Has a frequency f_LO+3f_MOr f_LO-f_MThe power of the plurality of signal components. When the power E (n-3) is lower than the power E (n-2), it indicates that the adjustment direction is wrong. Under this condition, one of the coefficients C1-C2 can be adjusted to correct the adjustment direction of the coefficients C1-C2. Alternatively, when the power E (n-3) is higher than the power E (n-2), it indicates that the adjustment direction is correct.

In operation S404, a coefficient C1(n) is generated, wherein

C1(n) ═ C1(n-2) + SIGN _ C1 STEP _ C1. In the above equation, C1(n-2) is the value of the coefficient C1 at the previous 2 times, and STEP _ C1 is the predetermined adjustment value of the coefficient C1. For example, when the adjustment directions of the coefficients C1-C2 are incorrect, the coefficient C1 may be changed to decrease the predetermined adjustment value STEP _ C1 to generate a new coefficient C1. Alternatively, when the adjustment direction of the coefficients C1-C2 is correct, the coefficient C1 may be increased by the predetermined adjustment value STEP _ C1 to generate a new coefficient C1.

In operation S405, a new coefficient C1(n) is output and the coefficient C2 is maintained, and the adjustment number n is increased, i.e., n is n + 1.

In operation S406, it is confirmed that the first three measurements have a frequency of 2f_MOr 4f_MIs lower than the power of the signal component of (i.e. E (n-3)) measured in the first two times with a frequency of 2f_MOr 4f_MThe power of the signal component of (i.e., E (n-2)). If so, operation S407 is performed. Otherwise, operation S408 is performed.

In operation S407, the adjustment direction of the coefficient C2 is reset to decrease, i.e., SIGN _ C2 is set to-SIGN _ C2.

In operation S408, a coefficient C2(n) is generated, wherein

C2(n) ═ C2(n-2) + SIGN _ C2 STEP _ C2. C2(n-2) is the value of the coefficient C2 at the first two adjustments, and STEP _ C2 is the predetermined adjustment value of the coefficient C2.

After the adjustment of the coefficient C1(n), it is possible to confirm whether an error occurs in the adjustment direction of the coefficient C2 by the same procedure, and output the coefficient C2(n) after confirming the adjustment direction of the coefficient C2. Operations S406-S408 are similar to operations S402-S404, and therefore are not described herein.

In operation S409, it is determined whether the adjustment number n exceeds a threshold. If yes, the adjustment is ended, and the coefficients C1-C2 are output. If not, operation S402 is repeated to further adjust the coefficients C1-C2 to better values. By setting operation S409, the operating efficiency of the transmitter 120 can be maintained.

The above-described modes of adjusting the coefficients C1 to C2 are merely examples. The arrangement of the various adjustable coefficients C1-C2 should be considered to be within the scope of the present disclosure.

With continued reference to fig. 3, the calculating circuit 324 includes a phase correcting circuit 324A and a correcting signal generating circuit 324B. The phase correction circuit 324A generates the in-phase data signal I [ n ] according to the coefficients C1-C2]And quadrature data signal Q [ n ]]Generating a pre-phase correction signal

The correction signal generation circuit 324B corrects the signal according to the pre-phase

In-phase data signal I [ n ]]And quadrature data signal Q [ n ]]The generated correction signal I' n]And Q' [ n ]]To digital-to-analog converters 201 and 112. In some embodiments, the correction signal generation circuit 324B is implemented using digital circuitry that can implement the correction matrix 200A shown in fig. 2C. The correction signal generation circuit 324B can correct the signal according to the pre-phase

The phase error θ (t) is generated and the operation of equation (5) is performed to generate the correction signals I '(t) and Q' (t).

Fig. 5 is a schematic diagram of the phase correction circuit 324A of fig. 3 according to some embodiments of the present disclosure. In this example, the phase correction circuit 324A includes multipliers 501 to 505, a subtractor 506 and an adder 507. The phase correction circuit 324A in this example is suitable for narrow frequency applications.

Multiplier 501 square-multiplies inphase data signal I n]To generate an operation value I²[n]. Multiplier 502 square-multiplies quadrature data signal Q n]To generate the calculated value Q²[n]. Multiplier 503 multiplies the in-phase data signal I n]And quadrature data signal Q [ n ]]To generate a calculation value I [ n ]]Q[n]. Subtractor 506 self-operation value I²[n]Minus the calculated value Q²[n]To generate an operation value I²[n]-Q²[n]. The multiplier 504 multiplies the coefficient C1 by the operation value I²[n]-Q²[n]To generate a calculated value C1 (I)²[n]-Q²[n])。

Multiplier 505 multiplies the doubled coefficient C2 by the operation value I [ n ]]Q[n]To generate the calculated value 2C2 (I [ n ]]Q[n]). The adder 507 adds the calculated value C1 (I)²[n]-Q²[n]) And the calculated value 2C2 (I n)]Q[n]) To generate a pre-phase correction signal

According to equation (7), the phase correction circuit 500 generates the pre-phase correction signal

To eliminate the effect of the pulling effect.

Fig. 6 is a schematic diagram of the phase correction circuit 324A of fig. 3 according to some embodiments of the present disclosure. The phase correction circuit 324A in this example is suitable for wideband applications.

Compared to FIG. 5, the phase calibration circuit 324A further includes finite impulse filters 601-602, which replace the mixer 504 and the mixer 505, respectively.

In some embodiments, the finite impulse filters 601-602 can be designed to generate the desired operation values by designing the coefficients of each stage (TAP). For example, in the bandwidth to be corrected of the transmitter 120, N frequencies f can be inputted sequentially_iWhere i is 1,2, … and N is a positive integer, to transmitter 120. The signal power detector 323 can accordingly detect the signal having the frequency 2f_iOr 4f_iThe power of the signal component of (a). Meanwhile, the method 400 adjusts the coefficients C1-C2 to obtainHaving a frequency 2f_iOr 4f_iThe power of the signal component of (a) is reduced. When having a frequency 2f_iOr 4f_iWhen the power of the signal component(s) is minimized, the stored coefficients C1-C2 are filter coefficients C1, i and C2, i. After obtaining N sets of coefficients C1, i and C2, i, the inverse fourier transform of C1, i through C1, N and their respective conjugates may be performed. In this way, each coefficient of the N-th order of the finite impulse filter 601 can be obtained from the real part of the result after the operation. Similarly, 2C2, i 2C2, N and their respective conjugates can be inverse Fourier transformed. In this way, each coefficient of the nth order of the finite impulse filter 602 can be obtained from the real part of the result after the operation. When calculating the value I²[n]-Q²[n]And the calculated value I [ n ]]Q[n]When passing through the finite impulse filters 601-602, the finite impulse filters 601-602 can output the corresponding operation values to the adder 507 to generate the pre-phase correction signal

Although the present disclosure has been described with reference to the above embodiments, it will be understood by those skilled in the art that various changes and modifications may be made without departing from the spirit and scope of the disclosure, and therefore, the scope of the disclosure is to be determined by the appended claims.

Claims

1. A signal transmission device, comprising:

a first transmitter, comprising:

a first oscillator circuit system for outputting a first oscillation signal;

a signal processing circuit system for mixing a plurality of calibration signals according to the first oscillating signal to transmit a first output signal; and

a correction circuitry for detecting the power of the first output signal to generate a plurality of coefficients and generating the correction signal according to the coefficients, an in-phase data signal and a quadrature data signal; and

a second oscillator circuit system disposed adjacent to the first transmitter for outputting a second oscillating signal,

wherein the calibration signal is configured to reduce a pull of both the first output signal and the second oscillator signal to the first oscillator circuitry.

2. The signal transmitting apparatus of claim 1, wherein the second oscillator circuit outputs the second oscillating signal to a second transmitter for transmitting a second output signal, and the calibration signal is further configured to reduce the pulling of the first output signal, the second oscillating signal and the second output signal to the first oscillator circuit.

3. The signal transmitting device of claim 1, wherein the first transmitter and the second oscillator circuitry are disposed in a single die, or disposed in a first die and a second die, respectively, and the first die and the second die are packaged in a single package.

4. The signal transmitting apparatus according to claim 1, wherein the frequency of the first oscillating signal is the same as the frequency of the second oscillating signal.

5. The signal transmitting apparatus according to claim 1, wherein a frequency of the first oscillating signal is different from a frequency of the second oscillating signal.

6. The signal transmission apparatus of claim 1, wherein the calibration circuitry comprises a feedback control circuit, and the feedback control circuit comprises:

a self-mixer for modulating the first output signal according to the first output signal to generate a detection signal;

an amplifier for amplifying the detection signal;

an analog-to-digital converter for generating a digital code according to the amplified detection signal; and

a correction circuit for generating the coefficients according to the digital code.

7. The signal transmission device of claim 6, wherein the calibration circuit comprises:

a signal power detector for detecting the power of a signal component according to the digital code to generate an adjustment signal, wherein the frequency of the signal component is two times or four times the frequency of the in-phase data signal or the quadrature data signal; and

and the adjusting circuit adjusts the coefficient according to the adjusting signal so as to reduce the power of the signal component.

8. The signal transmission apparatus according to claim 7, wherein the adjustment circuit obtains the power of the signal component according to the adjustment signal, and sequentially adjusts a first coefficient and a second coefficient of the coefficients.

9. A calibration method, comprising:

mixing, by a first transmitter, a plurality of calibration signals according to a first oscillation signal to transmit a first output signal, wherein the first oscillation signal is provided by a first oscillator circuitry of the first transmitter;

detecting the power of the first output signal to generate a plurality of coefficients; and

generating the correction signal based on the coefficients, an in-phase data signal and a quadrature data signal,

wherein the calibration signal is configured to reduce a pull-in of both the first output signal and a second oscillator signal to the first oscillator circuitry, and the second oscillator signal is provided by a second oscillator circuitry disposed adjacent to the first transmitter.

10. The calibration method of claim 9, wherein the first transmitter and the second oscillator circuitry are disposed in a single die, or disposed in a first die and a second die, respectively, and the first die and the second die are packaged in a single package.