CN111934577B - Current source inverter variable switching frequency modulation method and system - Google Patents

Current source inverter variable switching frequency modulation method and system Download PDF

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CN111934577B
CN111934577B CN202010696679.8A CN202010696679A CN111934577B CN 111934577 B CN111934577 B CN 111934577B CN 202010696679 A CN202010696679 A CN 202010696679A CN 111934577 B CN111934577 B CN 111934577B
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current
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CN111934577A (en
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蒋栋
刘康
王若栋
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Huazhong University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H02M7/53876—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output based on synthesising a desired voltage vector via the selection of appropriate fundamental voltage vectors, and corresponding dwelling times
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Power Engineering (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • Inverter Devices (AREA)

Abstract

The invention discloses a variable switching frequency modulation method and system based on a current source inverter, and belongs to the field of electrical systems. The scheme adopts variable switching frequency modulation, takes the current ripple of the direct current side inductor as a control target, and dynamically reduces the average switching frequency by adjusting the switching frequency in real time to ensure that the inductance current ripple is at the maximum tolerance value as much as possible. The scheme can not increase the design volume of the inductor, effectively reduce the overall switching loss of the inverter and improve the electromagnetic interference characteristic of a system. The invention is based on classical space vector modulation, has larger direct current utilization ratio and fast dynamic response; and a symmetrical five-section wave-sending scheme is adopted in the time sequence, so that the low-order harmonic component of the alternating-current side voltage is reduced while the prediction accuracy of the inductive current ripple is ensured.

Description

Current source inverter variable switching frequency modulation method and system
Technical Field
The invention belongs to the field of electrical systems, and particularly relates to a variable switching frequency modulation method and system for a current source inverter.
Background
With the rapid development of power electronics technology, Voltage Source Inverters (VSI) have become popular in various fields, while Current Source Inverters (CSI) have relatively fewer application fields, which are mainly limited by the volume of energy storage elements and energy characteristics. However, in some occasions, the CSI has a longer service life and a boosting capability, and has a very high application potential in various fields such as photovoltaic grid-connected power generation, high-voltage direct-current power transmission, wind power generation and the like.
As with most power electronic converters, because there is switching loss during the on/off process of the switching tube, the CSI also generates corresponding power loss during the operation process. At present, two ideas are mainly used for solving the problem, one is to add an additional energy storage circuit to realize soft switching, the method increases the cost and has complex parameter design, and generally has better benefits under specific working conditions; the other is to reduce the requirements on some indexes, and reduce the actions of the switch as much as possible, thereby reducing the loss.
At present, a variable switching frequency PWM technology for a voltage source inverter has been developed to some extent, but no corresponding strategy exists for a current source inverter. Chinese patent document CN110932583A discloses a ZVS implementation method for a current source type dual three-phase permanent magnet synchronous motor driving system, which belongs to the field of implementing ZVS by controlling and suppressing the charging and discharging of a resonant capacitor, can reduce dv/dt of a wide bandgap semiconductor, suppress electromagnetic interference of a high frequency converter, and simultaneously, the addition of a soft switch improves the efficiency of the system, which is beneficial to improving the power density of the system and reducing the heat dissipation cost of the converter. However, the above patents have increased system complexity and are not very versatile. Therefore, it is one of the key issues to be solved urgently how to reduce the switching loss of the current source inverter by reducing the switching operation as much as possible without increasing the system complexity.
Disclosure of Invention
In view of the drawbacks of the related art, the present invention provides a method and a system for modulating a variable switching frequency of a current source inverter, which aim to reduce the switching loss of the current source inverter without increasing the complexity of the system.
To achieve the above object, one aspect of the present invention provides a method for modulating a variable switching frequency of a current source inverter, comprising the steps of:
s1, utilizing the switching frequency f of the last control periods_NCombining the output current command given by the preceding stage controller according to the power transmission requirement
Figure BDA0002590599410000021
For the last control period pairPeak value of current ripple of inductoririp_NCarrying out prediction;
s2, setting the target peak value delta i of the direct-current side inductor current ripple according to the magnetic flux saturation condition of the direct-current side inductorrip_reqObtaining the target peak value delta i of the inductor current ripplerip_reqCorresponding actual required switching frequency fs_req；
S3, according to the output current instruction
Figure BDA0002590599410000022
And said actual desired switching frequency fs_reqAnd a five-section wave-sending time sequence is adopted for PWM control.
Further, the step S1 includes:
s11, the front-stage controller gives an output current instruction according to the requirement of power transmission
Figure BDA0002590599410000023
Deriving the time parameter T by space vector decomposition1/TsAnd T2/TsWherein T issFor a switching period, T1、T2Respectively, basic vectors in space vector modulation
Figure BDA0002590599410000027
And
Figure BDA0002590599410000028
the respective duration of time;
s22, utilizing the switching frequency f of the last control periods_NObtaining the time of action of the basis vector
Figure BDA0002590599410000024
And
Figure BDA0002590599410000025
according to the formula
Figure BDA0002590599410000026
Predicting the peak value delta i of the inductive current ripple corresponding to the previous control periodrip_N(ii) a Wherein, VdcIs the bus voltage, LdcIs a direct current inductance value, va，vb，vcIs the three-phase voltage on the alternating current side.
Further, the step S2 includes:
setting a target peak value delta i of a direct-current side inductor current ripple according to the magnetic flux saturation condition of a direct-current side inductorrip_reqAccording to the formula
fs_req＝fs_N×Δirip_N/Δirip_req
Obtaining a target peak value delta i of the inductive current ripplerip_reqCorresponding actual required switching frequency fs_req。
Further, the step S2 further includes:
obtaining the actually required switching frequency fs_reqThen, checking whether the current voltage is within the normal working range of the switching device, if so, continuing to execute S3; otherwise, Δ i needs to be readjustedrip_reqUp to said actual desired switching frequency fs_reqWithin the normal operating range of the switching device.
Further, the step S3 includes:
according to the actually required switching frequency fs_reqAnd said time parameter T1/Ts、T2/TsTo obtain new action time of adjacent vector
Figure BDA0002590599410000031
Sum zero vector action time
Figure BDA0002590599410000032
And then PWM control is carried out by adopting a five-segment wave-sending time sequence.
In another aspect of the present invention, a system for modulating the switching frequency of a current source inverter is provided, which comprises
Inductor current ripple prediction unit using the switch of the previous control cycleFrequency fs_NCombining the output current command given by the preceding stage controller according to the power transmission requirement
Figure BDA0002590599410000033
The peak value delta i of the inductive current ripple corresponding to the previous control periodrip_NCarrying out prediction;
a switching frequency updating unit for setting a target peak value Delta i of the DC side inductor current ripple according to the magnetic flux saturation condition of the DC side inductorrip_reqObtaining the target peak value delta i of the inductor current ripplerip_reqCorresponding actual required switching frequency fs_req；
A control drive unit for controlling the drive unit according to the output current command
Figure BDA0002590599410000047
And said actual desired switching frequency fs_reqAnd a five-section wave-sending time sequence is adopted for PWM control.
Further, the inductor current ripple prediction unit comprises
An active time acquisition unit, wherein the pre-stage controller gives an output current command according to the power transmission requirement
Figure BDA0002590599410000041
Deriving the time parameter T by space vector decomposition1/TsAnd T2/TsWherein T issFor a switching period, T1、T2Respectively, basic vectors in space vector modulation
Figure BDA0002590599410000042
And
Figure BDA0002590599410000043
the respective duration of time;
an inductive current ripple peak value obtaining unit for obtaining the switching frequency f of the previous control periods_NObtaining the time of action of the basis vector
Figure BDA0002590599410000044
And
Figure BDA0002590599410000045
according to the formula
Figure BDA0002590599410000046
Predicting the peak value delta i of the inductive current ripple corresponding to the previous control periodrip_N(ii) a Wherein, VdcIs the bus voltage, LdcIs a direct current inductance value, va，vb，vcIs the three-phase voltage on the alternating current side.
Further, the switching frequency updating unit updates the switching frequency according to a formula
fs_req＝fs_N×Δirip_N/Δirip_req
Obtaining a target peak value delta i of the inductive current ripplerip_reqCorresponding actual required switching frequency fs_req。
Further, the switching frequency updating unit further comprises
A checking unit for obtaining the actually required switching frequency fs_reqThen, checking whether the switching element is positioned in the normal working range of the switching element, if so, continuing to execute; otherwise, Δ i needs to be readjustedrip_reqUp to said actual desired switching frequency fs_reqWithin the normal operating range of the switching device.
Further, the control drive unit controls the switching frequency f according to the actual required switching frequencys_reqAnd said time parameter T1/Ts、T2/TsTo obtain new action time of adjacent vector
Figure BDA0002590599410000051
Figure BDA0002590599410000052
Sum zero vector action time
Figure BDA0002590599410000053
And then PWM control is carried out by adopting a five-segment wave-sending time sequence.
Through the technical scheme, compared with the prior art, the invention has the following beneficial effects:
(1) according to the variable switching frequency modulation method of the current source inverter based on the inductor current ripple prediction, the peak value of the inductor current ripple is controlled to be the maximum tolerance value of the peak value by fully utilizing the saturation critical point of the inductor magnetic flux on the direct current side, and the switching frequency can be properly reduced. The modulation mode of changing the switching frequency can not increase the design volume of the inductor, effectively reduces the turn-on and turn-off times of the inverter switching tube, and further reduces the loss of the whole operation.
(2) According to the variable switching frequency modulation method of the current source inverter based on the inductor current ripple prediction, the switching frequency can be changed in real time based on the predicted value of the inductor current ripple on the direct current side. By setting the switching frequency to change within a certain range, the peak value of EMI interference can be effectively reduced.
Drawings
FIG. 1 is a general topology of a current source inverter provided by an embodiment of the present invention;
fig. 2 is a schematic diagram of space vector modulation of a current source inverter according to an embodiment of the present invention;
FIG. 3 is a wave-making timing sequence of the five-segment SVM provided by the embodiment of the present invention in sector 1;
fig. 4 shows a dc side inductor current ripple of the five-segment SVM provided in the embodiment of the present invention in sector 1;
fig. 5 is a control flow chart of a variable switching frequency SVM modulation technique based on inductor current ripple prediction according to an embodiment of the present invention;
fig. 6(a) is a simulation diagram of a dc-side inductor current ripple of a conventional constant switching frequency current source inverter according to an embodiment of the present invention;
fig. 6(b) is a simulation diagram of a dc-side inductor current ripple of a current source inverter adopting a steady modulation method according to an embodiment of the present invention;
fig. 7 is a simulation diagram of the switching frequency reduction effect according to the embodiment of the present invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.
The embodiment of the invention provides a variable switching frequency modulation method taking direct-current side inductive current as an optimization target for a classic topology (as shown in figure 1) of a universal current source inverter for controlling a three-phase load, and the method comprises the following steps:
s1, utilizing the switching frequency f of the last control periods_NCombining the output current command given by the preceding stage controller according to the power transmission requirement
Figure BDA0002590599410000061
The peak value delta i of the inductive current ripple corresponding to the previous control periodrip_NCarrying out prediction;
s2, setting the target peak value delta i of the direct-current side inductor current ripple according to the magnetic flux saturation condition of the direct-current side inductorrip_reqObtaining the target peak value delta i of the inductor current ripplerip_reqCorresponding actual required switching frequency fs_req；
S3, according to the output current instruction
Figure BDA0002590599410000062
And said actual desired switching frequency fs_reqAnd a five-section wave-sending time sequence is adopted for PWM control.
The invention adopts variable switching frequency modulation, takes the current ripple of the direct current side inductor as a control target, and leads the current ripple of the inductor to be at the maximum tolerance value as much as possible through the real-time adjustment of the switching frequency, thereby dynamically changing the reduction of the average switching frequency. Under the premise of knowing the direct-current voltage and the three-phase voltage of the alternating-current side of the bus, the peak value of the inductive current ripple of the next control period can be directly predicted according to the control command. If the three-phase voltage at the alternating current side is not measured, the voltage can be further calculated according to known feedback signals such as filter parameters and current and the voltage of the power grid; if the load is a resistive load or a resistive-inductive load, the load parameter may be indirectly estimated by a feedback signal such as a control command or a current when the load parameter is known.
According to the scheme combining the five-segment space vector modulation and the inductive current ripple prediction, the switching action is always symmetrical in each switching period, the inductive current ripple is guaranteed to be symmetrical up and down, and prediction is facilitated. Compared with a three-stage wave-generating mode, the five-stage wave-generating mode increases the switching times once in each switching period, but also greatly improves the output voltage ripple.
The basic principle of the present invention is described below. Since most energy sources exhibit the characteristics of a voltage source or a large capacitor is connected in parallel to the power source side for voltage stabilization, the voltage on the capacitor can be assumed to be basically constant in the analysis process. The three-phase voltage on the load side is often the target of current source inverter regulation, so direct measurement or indirect calculation is generally needed.
The invention takes the peak value of the direct current side inductance ripple as the primary regulation and control target, and the inductance ripple is always kept at the same value by changing the switching frequency in real time and works near the inductance magnetic flux saturation point so as to fully utilize the optimal working point of the magnetic core and simultaneously prevent the inductance from being saturated. Specifically, the maximum ripple value of the inductor is predicted in each control period, and the switching frequency capable of maintaining the maximum ripple value is designed in the next control period and is cycled all the time.
For the PWM modulation itself, the present invention adopts a five-segment Space Vector Modulation (SVM) method, as shown in fig. 2 and 3, which has 9 sets of standard current vectors, as shown in table 1, where 3 sets of zero vectors respectively correspond to the three-phase bridge arm through condition. The direct current modulated by the SVM has high utilization rate, quick dynamic response and less generated harmonic waves, and is widely applied at present. The five-section time distribution method can make the inductance ripple waves symmetrical in time and is beneficial to improving the content of the output voltage low-order harmonic waves.
TABLE 1 space vector modulation and Voltage State Table
Figure BDA0002590599410000071
Figure BDA0002590599410000081
To direct current inductance LdcComprises the following steps:
Figure BDA0002590599410000082
bus voltage V in generaldcAre constantly known or can be measured directly during actual operation. Therefore, only the voltage V at the two ends of the bridge arm is knownpnThen the change rate of the inductive current at that moment can be calculated. When the current source inverter switching tube works, the voltage V at the two ends of the bridge armpnIs a three-phase voltage v on the AC sidea，vb，vcLinear combination of the two output voltages. Under the modulation of SVM, the 9 standard current vectors all have their respective VpnExpressions, as shown in table 1. When any standard current vector acts, the fundamental frequency is far lower than the switching frequency, the three-phase voltage change in one switching period is small, the output voltage can be considered to be approximately constant in the period, the change rate of the inductive current can be considered as a constant value at the moment, and the inductive current ripple approximately linearly rises or falls, as shown in fig. 4.
Taking the output current vector in the first sector as an example, according to the space vector synthesis calculation principle, the output reference current vector is required
Figure BDA0002590599410000083
Can be expressed as:
Figure BDA0002590599410000084
wherein T issFor a switching period, T1Is a base vector
Figure BDA0002590599410000085
Duration of time, T2Is a base vector
Figure BDA0002590599410000086
Duration of time, remaining time T0＝Ts-T1-T2Is the action time of the zero vector. With the five-segment output sequence shown in FIG. 3, when T is more than 0 and less than T1At/2, the action vector is
Figure BDA0002590599410000087
The corresponding inductance current change rate k can be known from the formula (1)1＝(Vdc-va+vb)/Ldc. The state is maintained by T1The ripple amplitude of the inductive current caused by the time of/2 is h1：
Figure BDA0002590599410000091
When T is1/2＜t＜(T1+T2) At/2, the action vector is
Figure BDA0002590599410000092
Corresponding inductance current change rate k2＝(Vdc-va+vc)/LdcThe state is maintained as T2The ripple amplitude of the inductive current caused by the time of/2 is h2：
Figure BDA0002590599410000093
Finally, the peak value of the ripple of the inductor current corresponding to the switching period is as follows:
Δirip＝max(|h1|，|h2|) (5)
in a real system, VdcAnd a DC inductance value LdcIn known amounts. When the variable switching frequency modulation method is adopted, the switching period TsWill change, T1，T2And therefore the peak inductor current ripple can be adjusted by changing the switching period, i.e. changing the switching frequency. Under the condition of complex loads such as grid connection at the alternating current side or motor connection, the output three-phase voltage can be obtained by measurement; when the load is a resistive or resistive-inductive load, the output voltage can be directly calculated according to the circuit parameters and the three-phase duty ratio. Therefore, before the switching pulse is sent out, the ripple value of the inductive current corresponding to the specific switching frequency at the moment can be predicted and obtained through mathematical calculation. Therefore, a ripple prediction model of the inductive current is established.
Next, a variable switching frequency SVM modulation technique aimed at maintaining the peak value of the inductor current ripple constant is implemented based on the ripple prediction model described above. As can be seen from equations (3) and (4), the magnitude of the inductor current ripple in any switching period is proportional to the total time for maintaining the state, and thus inversely proportional to the switching frequency. Let fs_NFor the switching frequency of the last switching cycle, Δ irip_NIs fs_NCorresponding value of ripple, Δ irip_reqFor a set target value of the inductor current ripple, fs_reqIs to satisfy Δ irip＝Δirip_reqAnd the actually required switching frequency, the iterative process satisfies the following relation:
fs_req＝fs_N×Δirip_N/Δirip_req (6)
thus, the variable switching frequency PWM model shown in fig. 5 is established: the preceding controller gives a reference current vector
Figure BDA0002590599410000101
Can then be based on the vector of FIG. 2Decomposing to obtain a time parameter T1/TsAnd T2/Ts(ii) a Time parameter T1/TsAnd T2/TsSending the ripple prediction model to obtain the standard switching frequency f from the formulas (3), (4) and (5)s_NCorresponding ripple value of inductor current Δ irip_NThen, the actually required switching frequency f is obtained according to the formula (6)s_reqAnd finally, updating the switching frequency through the SVM module and sending out switching pulses.
The invention mainly focuses on the direct-current side current of the CSI, and the alternating-current side load influences the change of the actual switching frequency through the three-phase voltage, but does not influence the ripple prediction model and the SVM modulation technology of changing the switching frequency. The SVM modulation technique of varying switching frequency is applicable to various loads. When loads such as a parallel power grid or a motor are output, the output three-phase voltage needs to be obtained through measurement and used for calculation of a prediction model, but when the output is a resistive load or an RL load with known parameters, the three-phase voltage can be directly calculated through circuit parameters and a duty ratio.
The contents of the above embodiments will be described with reference to a preferred embodiment.
According to the connection mode shown in FIG. 1, the voltage source is connected in series with the DC side inductor L from front to backdcThe three-phase inverter bridge and the LC filter circuit are finally connected to a load. In order to ensure the accuracy of control, the control circuit needs to apply a direct current voltage VdcMonitoring while measuring the AC side voltage va，vb，vcAnd also monitored by a sensor or calculated in real time by measuring the inductive current. The modulation method of the invention comprises the following steps:
step 1: the pre-stage controller gives an output current instruction according to the requirement of power transmission
Figure BDA0002590599410000102
Then, T can be obtained according to space vector decomposition1/TsAnd T2/TsThereby making it possible to
Figure BDA0002590599410000103
Means capable of acting on T by two adjacent basic vectors1And T2The time is obtained, and the specific calculation process is obtained according to the sine theorem, which is not described herein again.
Step 2: switching frequency f calculated using the last control periods_NObtaining the time of action
Figure BDA0002590599410000104
And
Figure BDA0002590599410000105
bring it into formula
Figure BDA0002590599410000106
Obtaining the peak value of the inductive current ripple wave corresponding to the switching period as delta irip_N。
And step 3: setting a ripple target peak value delta i of the direct-current side inductive current according to the magnetic flux saturation condition of the direct-current side inductive currentrip_reqThen according to formula
fs_req＝fs_N×Δirip_N/Δirip_reqTo obtain the actually required switching frequency fs_req. To ensure that the switching device can operate properly, the switching frequency f should be checked at this times_reqWhether it is within the normal operating range of the switching device, and if it is out of the normal operating range, the Δ i is readjustedrip_req。
And 4, step 4: by output current command
Figure BDA0002590599410000111
And a new switching frequency fs_reqUsing the calculated time parameter T1/TsAnd T2/TsTo obtain the new action time of the adjacent vector in FIG. 2
Figure BDA0002590599410000112
And
Figure BDA0002590599410000113
and zero vector action time
Figure BDA0002590599410000114
And then the five-segment wave-sending time sequence shown in fig. 3 is adopted to be output to the PWM module of the control chip, and switching pulses are sent to drive each switching tube.
In order to further illustrate the technical objects and effects of the present invention, simulation experiments were performed on the application under RL load by software using the above modulation method.
Fig. 6(a) shows the dc side inductor current ripple when the conventional current source inverter is driven with a constant switching frequency, and it can be seen that it reaches a maximum value only in a partial region, and this maximum value directly determines the design of the inductor saturation flux. In order to fully utilize the designed maximum flux margin, the modulation method provided by the invention is adopted to obtain the direct-current side inductor current ripple shown in fig. 6(b), and the ripple is always kept near the same peak value. The switching frequency change in this state is shown in fig. 7, and compared with the conventional fixed switching frequency modulation, the method provided by the invention reduces the average switching frequency of the device, and can effectively reduce the turn-on and turn-off loss of the device.
In another aspect, a system for modulating a switching frequency of a current source inverter is provided, which includes
An inductor current ripple prediction unit using the switching frequency f of the previous control periods_NCombining the output current command given by the preceding stage controller according to the power transmission requirement
Figure BDA0002590599410000121
The peak value delta i of the inductive current ripple corresponding to the previous control periodrip_NCarrying out prediction;
a switching frequency updating unit for setting a target peak value Delta i of the DC side inductor current ripple according to the magnetic flux saturation condition of the DC side inductorrip_reqObtaining the target peak value delta i of the inductor current ripplerip_reqCorresponding actual required switching frequency fs_req；
A control drive unit for controlling the drive unit according to the output current command
Figure BDA0002590599410000122
And said actual desired switching frequency fs_reqAnd a five-section wave-sending time sequence is adopted for PWM control.
The functions of each unit can be referred to the description of the foregoing method embodiments, and are not described herein again.
In order to reduce the overall loss of the inverter, the invention selects and fully utilizes the ripple characteristic of the direct-current side inductive current to ensure that the inductive current ripple is at the maximum tolerance value as much as possible, and dynamically switches to reduce the switching frequency, namely, adopts a variable switching frequency pulse width modulation method. The modulation mode of changing the switching frequency does not increase the design volume of the inductor, but can effectively reduce the switching loss of the system and the peak of electromagnetic interference (EMI) in certain frequency bands, so that the peak is more even.
It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.

Claims (8)

1. A current source inverter variable switching frequency modulation method is characterized by comprising the following steps:
s1, utilizing the switching frequency f of the last control periods_NCombining the output current command given by the preceding stage controller according to the power transmission requirement
Figure FDA0003217558970000011
The peak value delta i of the inductive current ripple corresponding to the previous control periodrip_NCarrying out prediction;
s2, setting the target peak value delta i of the direct-current side inductor current ripple according to the magnetic flux saturation condition of the direct-current side inductorrip_reqObtaining the target peak value delta i of the inductor current ripplerip_reqCorresponding actual required switching frequency fs_req；
S3, according to the output current instruction
Figure FDA0003217558970000012
And said actual desired switching frequency fs_reqPerforming PWM control by adopting a five-section wave-sending time sequence;
the step S1 includes:
s11, the front-stage controller gives an output current instruction according to the requirement of power transmission
Figure FDA0003217558970000013
Deriving the time parameter T by space vector decomposition1/TsAnd T2/TsWherein T issFor a switching period, T1、T2Respectively, basic vectors in space vector modulation
Figure FDA0003217558970000014
And
Figure FDA0003217558970000015
the respective duration of time;
s22, utilizing the switching frequency f of the last control periods_NObtaining the time of action of the basis vector
Figure FDA0003217558970000016
And
Figure FDA0003217558970000017
according to the formula
Figure FDA0003217558970000018
Predicting the peak value delta i of the inductive current ripple corresponding to the previous control periodrip_N(ii) a Wherein, VdcIs the bus voltage, LdcIs a direct current inductance value, va,vb,vcIs the AC sideThree-phase voltage.
2. The current source inverter variable switching frequency modulation method according to claim 1, wherein the step S2 includes:
setting a target peak value delta i of a direct-current side inductor current ripple according to the magnetic flux saturation condition of a direct-current side inductorrip_reqAccording to the formula
fs_req＝fs_N×Δirip_N/Δirip_req
Obtaining a target peak value delta i of the inductive current ripplerip_reqCorresponding actual required switching frequency fs_req。
3. The current source inverter variable switching frequency modulation method according to claim 2, wherein the step S2 further comprises:
obtaining the actually required switching frequency fs_reqThen, checking whether the current voltage is within the normal working range of the switching device, if so, continuing to execute S3; otherwise, Δ i needs to be readjustedrip_reqUp to said actual desired switching frequency fs_reqWithin the normal operating range of the switching device.
4. The current source inverter variable switching frequency modulation method according to claim 1, wherein the step S3 includes:
according to the actually required switching frequency fs_reqAnd said time parameter T1/Ts、T2/TsTo obtain new action time of adjacent vector
Figure FDA0003217558970000021
Sum zero vector action time
Figure FDA0003217558970000022
And then PWM control is carried out by adopting a five-segment wave-sending time sequence.
5. A current source inverter variable switching frequency modulation system is characterized by comprising
An inductor current ripple prediction unit using the switching frequency f of the previous control periods_NCombining the output current command given by the preceding stage controller according to the power transmission requirement
Figure FDA0003217558970000031
The peak value delta i of the inductive current ripple corresponding to the previous control periodrip_NCarrying out prediction;
a switching frequency updating unit for setting a target peak value Delta i of the DC side inductor current ripple according to the magnetic flux saturation condition of the DC side inductorrip_reqObtaining the target peak value delta i of the inductor current ripplerip_reqCorresponding actual required switching frequency fs_req；
A control drive unit for controlling the drive unit according to the output current command
Figure FDA0003217558970000032
And said actual desired switching frequency fs_reqPerforming PWM control by adopting a five-section wave-sending time sequence;
the inductor current ripple prediction unit includes:
an active time acquisition unit, wherein the pre-stage controller gives an output current command according to the power transmission requirement
Figure FDA0003217558970000033
Deriving the time parameter T by space vector decomposition1/TsAnd T2/TsWherein T issFor a switching period, T1、T2Respectively, basic vectors in space vector modulation
Figure FDA0003217558970000034
And
Figure FDA0003217558970000035
the respective duration of time;
inductive current rippleA peak value obtaining unit for obtaining the switching frequency f of the previous control periods_NObtaining the time of action of the basis vector
Figure FDA0003217558970000036
And
Figure FDA0003217558970000037
according to the formula
Figure FDA0003217558970000038
Predicting the peak value delta i of the inductive current ripple corresponding to the previous control periodrip_N(ii) a Wherein, VdcIs the bus voltage, LdcIs a direct current inductance value, va,vb,vcIs the three-phase voltage on the alternating current side.
6. The current source inverter variable switching frequency modulation system of claim 5, wherein the switching frequency update unit is according to a formula
fs_req＝fs_N×Δirip_N/Δirip_req
Obtaining a target peak value delta i of the inductive current ripplerip_reqCorresponding actual required switching frequency fs_req。
7. The current source inverter variable switching frequency modulation system according to claim 6, wherein the switching frequency updating unit further comprises:
a checking unit for obtaining the actually required switching frequency fs_reqThen, checking whether the switching element is positioned in the normal working range of the switching element, if so, continuing to execute; otherwise, Δ i needs to be readjustedrip_reqUp to said actual desired switching frequency fs_reqWithin the normal operating range of the switching device.
8. Such asThe current source inverter variable switching frequency modulation system of claim 5, wherein the control drive unit is configured to drive the inverter according to the actual desired switching frequency fs_reqAnd said time parameter T1/Ts、T2/TsTo obtain new action time of adjacent vector
Figure FDA0003217558970000041
Figure FDA0003217558970000042
Sum zero vector action time
Figure FDA0003217558970000043
And then PWM control is carried out by adopting a five-segment wave-sending time sequence.
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