CN111865865A - Frequency offset and phase offset estimation method suitable for high-sensitivity satellite-borne ADS-B receiver - Google Patents
Frequency offset and phase offset estimation method suitable for high-sensitivity satellite-borne ADS-B receiver Download PDFInfo
- Publication number
- CN111865865A CN111865865A CN202010772113.9A CN202010772113A CN111865865A CN 111865865 A CN111865865 A CN 111865865A CN 202010772113 A CN202010772113 A CN 202010772113A CN 111865865 A CN111865865 A CN 111865865A
- Authority
- CN
- China
- Prior art keywords
- value
- phase
- signal
- frequency
- phase offset
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2668—Details of algorithms
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2657—Carrier synchronisation
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2662—Symbol synchronisation
Abstract
A frequency offset and phase offset estimation method suitable for a high-sensitivity satellite-borne ADS-B receiver belongs to the technical field of aviation monitoring and comprises the following steps: s1: performing coarse frequency offset and phase offset estimation and compensation on the ADS-B digital baseband signal subjected to the preprocessing; s2: low-pass filtering and signal synchronization are carried out; s3: performing non-coherent demodulation; s4: carrying out fine frequency offset and phase offset estimation and compensation; s5: carrying out coherent demodulation; s6 outputs the ADS-B message. The invention makes the range and resolution of the frequency/phase offset estimation meet the coherent demodulation requirement of the satellite-borne ADS-B receiver, greatly increases the precision of the frequency/phase offset estimation, and has simple and convenient hardware realization. The method is carried out in a pipeline mode, matrix inversion is avoided, operation delay is small, and precision is high.
Description
Technical Field
The invention belongs to the technical field of aviation monitoring, and relates to a frequency offset and phase offset estimation method suitable for a high-sensitivity satellite-borne ADS-B receiver.
Background
Because the satellite is far away from the aircraft, the satellite-based ADS-B receiver has weak signals and low signal-to-noise ratio, the minimum power required to be processed by calculation is-102 dBm signals, the decoding rate needs to be more than 90%, a high-sensitivity satellite-based ADS-B receiver is required, the ground-based ADS-B receiver can only process the minimum-90 dBm signals, and the signal processing method is not suitable. At the present stage, coherent demodulation is not considered in the signal processing flow of the land-based ADS-B receiver and the satellite-based ADS-B receiver to improve the sensitivity, coherent demodulation is required to improve the decoding rate when the effective receiving of the ADS-B signals transmitted by the aircraft in the coverage area of the receiving antenna is completed, the coherent demodulation can improve the 3dB signal-to-noise ratio of the baseband signals, and because the carrier frequency transmitted by the transmitter is 1090 +/-1 MHz in DO260-B, frequency/phase deviation exists, in addition, the Doppler effect caused by a channel and the frequency/phase deviation caused by the receiving processing process can also be caused, the frequency/phase deviation estimation and compensation are required to be carried out when the sensitivity is improved by the coherent demodulation.
The frequency/phase offset estimation and compensation algorithm is generally divided into two steps, including coarse frequency/phase offset estimation and compensation and fine frequency/phase offset estimation and compensation.
In the signal processing process, in order to filter out the out-of-band noise as much as possible and improve the signal-to-noise ratio, a low-pass filter with a bandwidth approximate to the bandwidth of the signal is adopted, under the frequency offset, the low-pass filter with the bandwidth cannot filter out the complete signal, so that signal distortion is caused, the signal-to-noise ratio is deteriorated by using a filter with a larger bandwidth, and the decoding rate is reduced, so that the signal needs to be subjected to frequency/phase offset rough estimation and compensation, and after the compensation, the signal-to-noise ratio is improved through the low-pass filter with the bandwidth approximate to the bandwidth of the.
After the estimation and compensation of the coarse frequency/phase offset are performed, the frequency/phase offset is controlled within a certain range, but the estimation resolution is insufficient, and the received signal still has the frequency/phase offset within a certain range, which causes adverse effect on the coherent demodulation rate, so that a step of fine estimation of the frequency/phase offset is required to basically eliminate the signal frequency/phase offset. The traditional frequency/phase offset precise estimation algorithm usually performs zero crossing point calculation or phase difference calculation on a fixed sequence in a message to calculate a precise frequency/phase offset estimation value, and because the ADS-B message fixed sequence is short and the estimation error is large, the requirement of frequency/phase offset estimation error cannot be met.
Disclosure of Invention
In view of the above, the present invention provides a frequency offset and phase offset estimation method suitable for a high-sensitivity satellite-borne ADS-B receiver. Firstly, noncoherent demodulation is utilized, firstly, once decoding is carried out, so that signals of the whole message are points which can be used for estimation operation, the available time length is expanded from 8 mu s to 120 mu s, the least square fitting phase is adopted to estimate the fine frequency/phase offset, in addition, the points with low reliability in the whole message are deleted, the phase of the points is different from the trend of most points, and great influence is generated on the precision of curve least square fitting, therefore, before the fine frequency/phase offset estimation, phase mutation points are removed, and the precision of the frequency/phase offset estimation can be improved.
In order to achieve the purpose, the invention provides the following technical scheme:
a frequency offset and phase offset estimation method suitable for a high-sensitivity satellite-borne ADS-B receiver comprises the following steps:
s1: performing coarse frequency/phase offset estimation and compensation on the ADS-B digital baseband signal subjected to the preprocessing;
s2: low-pass filtering and signal synchronization are carried out;
s3: performing non-coherent demodulation;
s4: performing fine frequency/phase offset estimation and compensation;
s5: carrying out coherent demodulation;
s6 outputs the ADS-B message.
Further, the preprocessing in step S1 includes initial filtering, initial synchronization, identifying the range of the signal, and performing segmentation processing on the signal, so that each signal segment only contains one ADS-B message.
Further, the performing of coarse frequency/phase offset estimation in step S1 includes:
performing fast Fourier transform calculation on the signals;
comparing the calculation result with the signal frequency spectrum without frequency deviation;
calculating the frequency spectrum shifting amount to obtain a numerical value estimated by the coarse frequency offset;
and calculating the phase of the FFT spectrum peak value to obtain a numerical value estimated by the coarse phase offset.
Further, the performing of coarse frequency/phase offset compensation in step S1 includes:
a coordinate rotation digital computer (CORDIC) module generating sine values and cosine values is used as a digital controlled oscillator (NCO), an angle generated by a phase accumulator is used as phase input of the CORDIC module, and sine waves and cosine waves with initial phases of phase deviation and frequency deviation are generated and multiplied by original signals to compensate.
Further, in step S2, the signal is passed through a raised cosine filter with a bandwidth as the signal bandwidth to further improve the signal-to-noise ratio, and then the accurate frame header position in the signal segment is determined through signal synchronization.
Further, the non-coherent demodulation in step S3 includes bit decision and confidence extraction, error detection and error correction; the bit decision and the confidence coefficient are used for extracting and obtaining a bit decoding result and a confidence coefficient matrix of each possible message; the error detection and correction are used for improving the correct decoding rate of the non-coherent demodulation and checking whether the decoding result is correct.
Further, the fine frequency/phase offset estimation and compensation in step S4 includes:
eliminating low confidence bits;
carrying out non-uniform signal interpolation;
and calculating the frequency/phase offset value of the fine and coarse frequency/phase offset estimation by adopting a least square method to fit a curve.
Further, the culling low confidence bits comprises:
sorting the confidence matrixes by adopting a parallel full-comparison sorting algorithm;
selecting the 5-20 bits with the lowest confidence degree according to the signal-to-noise ratio, and setting the value in the corresponding position data register to be 0;
the arc value set of high confidence high level points in the range of-pi, pi is calculated using the arctangent calculation mode of CORDIC.
Further, the performing non-uniform signal interpolation includes:
interpolating the 0 values present inside a matrix Y of length 240, such that the time matrix X0Fixed, matrix R is also fixed to a matrix with dimension 2 x 240;
the state machine of the linear interpolation algorithm of the non-uniformly sampled signal comprises:
IDLE: indicating an idle state, entering a WAIT _ NO _ ZERO state if the enable signal is high, and storing a first data value in the shift register;
WAIT _ NO _ ZERO: the data reading is continued when the next non-0 value arrives, and each effective data enters the shift register; if the value is not 0, entering CAL _ OUTPUT _ CLEAR state, and if the value is still 0, still being in the state; if the count value is larger than or equal to 241, the 240 th bit of the data is not a non-zero value, and a COMPLETE _ END state is entered;
CAL _ OUTPUT _ END: performing linear interpolation calculation according to the numerical values of the first and last non-zero registers stored in the shift register, outputting the interpolated value of the last-dividing register and clearing the value of the last-dividing register, if the count value is not less than 242, indicating that the 240 th bit of the data is a non-zero value, and entering an IDLE state after the interpolation is finished; if the count value is <242, entering the WAIT _ NO _ ZERO state continues to WAIT for a non-ZERO value to arrive;
COMPLETE _ END status: calculating interpolation results of several continuous 0 values behind 240-bit data according to the average increment of the previous interpolation calculation phase, outputting the interpolation results, and entering an IDLE state after the output is finished;
multiplying the value of the matrix R stored in the RAM by the Y matrix output after interpolation according to F-RY and accumulating to calculate a frequency/phase offset value; and according to the generated frequency offset and phase offset value, a CORDIC module is used as NCO, a phase accumulator is used for generating an angle, and sine waves and cosine waves with initial phases of phase offsets and frequency offsets are generated and multiplied by the original signal for compensation.
Further, the frequency/phase offset value of the fine and coarse frequency/phase offset estimation is calculated by fitting a curve by a least square method, and the formula is as follows:
wherein A is1-A240Complex values, t, representing points of the entire ADS-B message at the 2MHz sample rate of synchronization1-t240Representing the time corresponding to each point, wherein omega represents the frequency deviation, and phi represents the phase deviation;
Y＝X0f is a linear equation set required by fitting a curve, Y is an arc value matrix of the signal amplitude value calculated by the inverse tangent of the CORDIC module, and X0F is the calculated fine frequency/phase offset value, which is a time matrix, and is obtained as follows:
F＝(X0 TX0)-1X0 TY
making a time matrix X0FixingThe memory matrix R is then used to omit the inversion step of the least squares fit
R＝(X0 TX0)-1X0 T。
Further, the step of coherent demodulation of step S5 is the same as the step of non-coherent demodulation of step S3.
The invention has the beneficial effects that:
the invention can make the sensitivity of the satellite-borne ADS-B receiver more than or equal to-102 dBm (the decoding rate is more than or equal to 90 percent), and can finish the effective reception of the ADS-B signal transmitted by the aircraft within the coverage area of the receiving antenna. The coarse estimation is carried out for one time before the fine estimation, so that the problem of the fine estimation is solved, and the range and the resolution of the frequency/phase offset estimation meet the coherent demodulation requirements of the satellite-borne ADS-B receiver. The elimination of available points and phase catastrophe points is increased by non-coherent demodulation before fine estimation, and the most effective point is adopted for operation, so that the precision of frequency/phase offset estimation is greatly increased. Aiming at the design of hardware resources on the satellite, the hardware is simple and convenient to realize. The least square algorithm of the signals designed for the satellite-based ADS-B receiver firstly performs linear interpolation, outputs intermediate results in time and is performed in a pipeline mode, matrix inversion is avoided, operation delay is small, and accuracy is high.
Additional advantages, objects, and features of the invention will be set forth in part in the description which follows and in part will become apparent to those having ordinary skill in the art upon examination of the following or may be learned from practice of the invention. The objectives and other advantages of the invention may be realized and attained by the means of the instrumentalities and combinations particularly pointed out hereinafter.
Drawings
For the purposes of promoting a better understanding of the objects, aspects and advantages of the invention, reference will now be made to the following detailed description taken in conjunction with the accompanying drawings in which:
FIG. 1 is a schematic flow chart of a frequency offset and phase offset estimation method suitable for a high-sensitivity satellite-borne ADS-B receiver;
FIG. 2 is a state machine diagram of a linear interpolation algorithm for non-uniformly sampled signals;
fig. 3 is a comparison graph of several decoding rate curves.
Detailed Description
The embodiments of the present invention are described below with reference to specific embodiments, and other advantages and effects of the present invention will be easily understood by those skilled in the art from the disclosure of the present specification. The invention is capable of other and different embodiments and of being practiced or of being carried out in various ways, and its several details are capable of modification in various respects, all without departing from the spirit and scope of the present invention. It should be noted that the drawings provided in the following embodiments are only for illustrating the basic idea of the present invention in a schematic way, and the features in the following embodiments and examples may be combined with each other without conflict.
Wherein the showings are for the purpose of illustrating the invention only and not for the purpose of limiting the same, and in which there is shown by way of illustration only and not in the drawings in which there is no intention to limit the invention thereto; to better illustrate the embodiments of the present invention, some parts of the drawings may be omitted, enlarged or reduced, and do not represent the size of an actual product; it will be understood by those skilled in the art that certain well-known structures in the drawings and descriptions thereof may be omitted.
The same or similar reference numerals in the drawings of the embodiments of the present invention correspond to the same or similar components; in the description of the present invention, it should be understood that if there is an orientation or positional relationship indicated by terms such as "upper", "lower", "left", "right", "front", "rear", etc., based on the orientation or positional relationship shown in the drawings, it is only for convenience of description and simplification of description, but it is not an indication or suggestion that the referred device or element must have a specific orientation, be constructed in a specific orientation, and be operated, and therefore, the terms describing the positional relationship in the drawings are only used for illustrative purposes, and are not to be construed as limiting the present invention, and the specific meaning of the terms may be understood by those skilled in the art according to specific situations.
The baseband signal processing flow designed by the frequency/phase offset estimation algorithm suitable for the high-sensitivity satellite-borne ADS-B receiver is shown in figure 1.
The ADS-B digital baseband signal which passes through initial filtering and initial synchronization has the range of the signal identified, the signal is processed in a segmented mode, each signal segment only contains one ADS-B message, the processing method of each segment is the same, firstly, coarse frequency/phase deviation estimation and compensation are carried out, fast Fourier transform is calculated on the signal in the signal segment, the calculation result is compared with the signal frequency spectrum without frequency deviation, frequency spectrum shifting quantity is calculated, the phase of the FFT frequency spectrum peak value is calculated for the value estimated by the coarse frequency deviation, the value estimated by the coarse phase deviation is calculated, and then compensation is carried out. During compensation, a coordinate rotation digital computer (CORDIC) module generating a sine value and a cosine value is used as a digital control oscillator (NCO), an angle generated by a phase accumulator is used as phase input of the CORDIC module, the error of NCO generated by a ROM is reduced, the resolution is improved, and sine waves and cosine waves which generate initial phases and frequencies are frequency offsets are multiplied by original signals to compensate.
After the coarse frequency/phase offset estimation and compensation are carried out, the signal-to-noise ratio is further improved through a raised cosine filter with the bandwidth being the signal bandwidth, and then the accurate frame head position in the signal section is determined through signal synchronization.
And then carrying out incoherent demodulation which comprises two steps including bit judgment and confidence extraction, error detection and error correction. And extracting the bit decision and the confidence coefficient to obtain a bit decoding result and a confidence coefficient matrix of each possible message, and improving the correct decoding rate of the non-coherent demodulation by error detection and correction and checking whether the decoding result is correct or not.
And (3) performing fine frequency/phase offset estimation and compensation, firstly removing low confidence coefficient bits which represent noise and have large influence on the data bits and high bit decision error probability, sequencing the confidence coefficient matrix by adopting a parallel full-comparison sequencing algorithm, selecting 5-20 bits with the lowest confidence coefficient according to the signal-to-noise ratio to set the numerical value in a corresponding position data register to be 0, and calculating an arc value set of a high-confidence-degree high-level point with the range of [ -pi, pi ] by using an arc tangent calculation mode of CORDIC. At the moment, the frequency/phase offset value of the fine and coarse frequency/phase offset estimation is calculated by using a least square method fitting curve, and the calculation formula is as follows
Wherein A is1-A240Complex values, t, representing points of the entire ADS-B message at the 2MHz sample rate of synchronization1-t240Representing the time corresponding to each point, wherein omega represents the frequency deviation, and phi represents the phase deviation;
Y＝X0f is a linear equation set required by fitting a curve, Y is an arc value matrix of the signal amplitude value calculated by the inverse tangent of the CORDIC module, and X0Is a time matrix, F is the precision frequency/phase offset value, and F is easily obtained as (X)0 TX0)-1X0 TY, because of the elimination of the low confidence point, the length and the content of the time matrix are changed when each message is decoded, the inversion operation in the FPGA consumes more resources and has long time delay, if the time matrix X is eliminated0Fixed, then R-matrix, R ═ X (X) can be stored0 TX0)-1X0 TThe memory matrix R may omit the inversion step of the least squares fit.
If the length of matrix Y is 0 except for the high-level high-confidence point, the value of 0 existing inside matrix Y needs to be interpolated to make time matrix X0Fixed, the matrix R is also fixed to a matrix of dimension 2 x 240. A state machine for designing a linear interpolation algorithm for non-uniformly sampled signals is shown in fig. 2, and each state is explained as follows:
IDLE, representing IDLE state, if the start signal is high, enter WAIT _ NO _ ZERO state and store the first data value into the shift register.
WAIT _ NO _ ZERO: indicating that the data continues to be read waiting for the next non-0 value to arrive, with each valid data entry into the shift register. If the value is not 0, the state of CAL _ OUTPUT _ CLEAR is entered, and if the value is still 0, the state is still existed. If the count value is greater than or equal to 241, indicating that the 240 th bit of the data is not a non-zero value, the COMPLETE _ END state is entered.
CAL _ OUTPUT _ END: the linear interpolation calculation is carried out according to the numerical values of the first and last non-zero registers stored in the shift register, the interpolated value of the last register is output, and the value of the last register is cleared, so that the data output is timely, the subsequent compensation waveform generation and multiplication compensation processes can be carried out in a pipeline mode, if the count value is more than or equal to 242, the 240 th bit of the data is a non-zero value, the interpolation is finished, and the IDLE state is entered. If the count value is <242, the WAIT _ NO _ ZERO state is entered to continue waiting for the arrival of a non-ZERO value.
COMPLETE _ END status: and calculating interpolation results of several continuous 0 values behind 240-bit data according to the average increment of the interpolation calculation phase, outputting the interpolation results, and entering an IDLE state after the output is finished.
The interpolation method has low time delay, the output data and the signal data have the same sequence, according to the F & ltRY & gt, the matrix R is also fixed to the matrix with the dimension of 2 & ltRY & gt 240 & ltY & gt 1 & ltY & gt, the value of the matrix R stored in the RAM and the interpolated output Y matrix are multiplied according to the F & ltRY & gt and accumulated to calculate the frequency/phase offset value. And according to the generated frequency offset and phase offset value, a CORDIC module is used as NCO, a phase accumulator is used for generating an angle, and sine waves and cosine waves with initial phases of phase offsets and frequency offsets are generated and multiplied by the original signal for compensation. At this point, the full frequency/phase offset estimation and compensation is completed.
The effect of coherent demodulation on decoding is tested, the SNR (2MHz) is set to be 2-12 dB, the interval is 0.5dB, 2000 pieces of data are respectively sent, the frequency offset is set to be 136kHz, the phase offset is 30 °, the number of low confidence points is removed to be 15, and the decoding rates of non-coherent demodulation non-violent decoding, non-coherent demodulation violent decoding, coherent demodulation non-violent decoding and coherent demodulation violent decoding are compared as shown in fig. 3.
When SNR (2MHz) is 5.061dB, the decoding rates of noncoherent demodulation violent decoding and coherent demodulation violent decoding are respectively 58.8% and 90.7%, the decoding rate is greatly improved, the decoding rate exceeds 90%, and the performance of the coherent demodulation violent decoding method is obviously improved according to curve trend.
Finally, the above embodiments are only intended to illustrate the technical solutions of the present invention and not to limit the present invention, and although the present invention has been described in detail with reference to the preferred embodiments, it will be understood by those skilled in the art that modifications or equivalent substitutions may be made on the technical solutions of the present invention without departing from the spirit and scope of the technical solutions, and all of them should be covered by the claims of the present invention.
Claims (10)
1. A frequency offset and phase offset estimation method suitable for a high-sensitivity satellite-borne ADS-B receiver is characterized by comprising the following steps: the method comprises the following steps:
s1: performing coarse frequency offset and phase offset estimation and compensation on the ADS-B digital baseband signal subjected to the preprocessing;
s2: low-pass filtering and signal synchronization are carried out;
s3: performing non-coherent demodulation;
s4: carrying out fine frequency offset and phase offset estimation and compensation;
s5: carrying out coherent demodulation;
s6 outputs the ADS-B message.
2. The method for estimating frequency offset and phase offset of the high-sensitivity satellite-borne ADS-B receiver according to claim 1, wherein: the preprocessing in step S1 includes initial filtering, initial synchronization, identifying the range of the signal, and performing segmentation processing on the signal, so that each signal segment only contains one ADS-B message.
3. The method for estimating frequency offset and phase offset of the high-sensitivity satellite-borne ADS-B receiver according to claim 2, wherein: in step S1, the performing coarse frequency offset and phase offset estimation includes:
performing fast Fourier transform calculation on the signals;
comparing the calculation result with the signal frequency spectrum without frequency deviation;
calculating the frequency spectrum shifting amount to obtain a numerical value estimated by the coarse frequency offset;
and calculating the phase of the FFT spectrum peak value to obtain a numerical value estimated by the coarse phase offset.
4. The method for estimating frequency offset and phase offset of the high-sensitivity satellite-borne ADS-B receiver according to claim 3, wherein: in step S1, the performing coarse frequency offset and phase offset compensation includes:
a coordinate rotation digital computer CORDIC module generating sine values and cosine values is used as an NCO of a digital control oscillator, an angle generated by a phase accumulator is used as phase input of the CORDIC module, and sine waves and cosine waves with initial phases of phase deviation and frequency deviation are generated and multiplied by original signals to compensate.
5. The method for estimating frequency offset and phase offset of the high-sensitivity satellite-borne ADS-B receiver according to claim 1, wherein: in step S2, the signal is passed through a raised cosine filter with a bandwidth as the signal bandwidth to further increase the signal-to-noise ratio, and then the accurate frame header position in the signal segment is determined by signal synchronization.
6. The method for estimating frequency offset and phase offset of the high-sensitivity satellite-borne ADS-B receiver according to claim 1, wherein: the non-coherent demodulation in step S3 includes bit decision and confidence extraction, error detection and error correction; the bit decision and the confidence coefficient are used for extracting and obtaining a bit decoding result and a confidence coefficient matrix of each possible message; the error detection and correction are used for improving the correct decoding rate of the non-coherent demodulation and checking whether the decoding result is correct.
7. The method for estimating frequency offset and phase offset of the high-sensitivity satellite-borne ADS-B receiver according to claim 1, wherein: the fine frequency offset and phase offset estimation and compensation in step S4 includes:
eliminating low confidence bits;
carrying out non-uniform signal interpolation;
and calculating the frequency deviation and the phase deviation value of the fine and coarse frequency deviation and the phase deviation estimation by adopting a least square method fitting curve.
8. The method for estimating frequency offset and phase offset of an ADS-B receiver on a high sensitivity satellite according to claim 7, wherein: the culling low confidence bits comprises:
sorting the confidence matrixes by adopting a parallel full-comparison sorting algorithm;
selecting the 5-20 bits with the lowest confidence degree according to the signal-to-noise ratio, and setting the value in the corresponding position data register to be 0;
the arc value set of high confidence high level points in the range of-pi, pi is calculated using the arctangent calculation mode of CORDIC.
9. The method for estimating frequency offset and phase offset of an ADS-B receiver according to claim 8, wherein: the performing non-uniform signal interpolation includes:
interpolating the 0 values present inside a matrix Y of length 240, such that the time matrix X0Fixed, matrix R is also fixed to a matrix with dimension 2 x 240;
the state machine of the linear interpolation algorithm of the non-uniformly sampled signal comprises:
IDLE: indicating an idle state, entering a WAIT _ NO _ ZERO state if the enable signal is high, and storing a first data value in the shift register;
WAIT _ NO _ ZERO: the data reading is continued when the next non-0 value arrives, and each effective data enters the shift register; if the value is not 0, entering CAL _ OUTPUT _ CLEAR state, and if the value is still 0, still being in the state; if the count value is larger than or equal to 241, the 240 th bit of the data is not a non-zero value, and a COMPLETE _ END state is entered;
CAL _ OUTPUT _ END: performing linear interpolation calculation according to the numerical values of the first and last non-zero registers stored in the shift register, outputting the interpolated value of the last-dividing register and clearing the value of the last-dividing register, if the count value is not less than 242, indicating that the 240 th bit of the data is a non-zero value, and entering an IDLE state after the interpolation is finished; if the count value is <242, entering the WAIT _ NO _ ZERO state continues to WAIT for a non-ZERO value to arrive;
COMPLETE _ END status: calculating interpolation results of several continuous 0 values behind 240-bit data according to the average increment of the previous interpolation calculation phase, outputting the interpolation results, and entering an IDLE state after the output is finished;
multiplying the value of the matrix R stored in the RAM by the Y matrix output after interpolation according to the result that F is RY and accumulating to calculate the frequency offset and the phase offset value; and according to the generated frequency offset and phase offset value, a CORDIC module is used as NCO, a phase accumulator is used for generating an angle, and sine waves and cosine waves with initial phases of phase offsets and frequency offsets are generated and multiplied by the original signal for compensation.
10. The method for estimating frequency offset and phase offset of an ADS-B receiver on a high sensitivity satellite according to claim 9, wherein: the method adopts a least square method to fit a curve to calculate the frequency deviation and the phase deviation value of the fine and coarse frequency deviation and the phase deviation estimation, and the formula is as follows:
wherein A is1-A240Complex values, t, representing points of the entire ADS-B message at the 2MHz sample rate of synchronization1-t240Representing the time corresponding to each point, wherein omega represents the frequency deviation, and phi represents the phase deviation;
Y＝X0f is a linear equation set required by fitting a curve, Y is an arc value matrix of the signal amplitude value calculated by the inverse tangent of the CORDIC module, and X0F is the precision frequency offset and phase offset value, which are obtained as the time matrix:
F＝(X0 TX0)-1X0 TY
making a time matrix X0Fixed, then the memory matrix R is used to omit the inversion step of the least squares fit
R＝(X0 TX0)-1X0 T。
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202010772113.9A CN111865865B (en) | 2020-08-04 | 2020-08-04 | Frequency offset and phase offset estimation method suitable for high-sensitivity satellite-borne ADS-B receiver |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202010772113.9A CN111865865B (en) | 2020-08-04 | 2020-08-04 | Frequency offset and phase offset estimation method suitable for high-sensitivity satellite-borne ADS-B receiver |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN111865865A true CN111865865A (en) | 2020-10-30 |
| CN111865865B CN111865865B (en) | 2021-06-15 |
Family
ID=72953372
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202010772113.9A Active CN111865865B (en) | 2020-08-04 | 2020-08-04 | Frequency offset and phase offset estimation method suitable for high-sensitivity satellite-borne ADS-B receiver |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN111865865B (en) |
Cited By (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN112565129A (en) * | 2020-12-08 | 2021-03-26 | 上海擎昆信息科技有限公司 | Frequency offset compensation method and system |
| CN112565131A (en) * | 2020-12-08 | 2021-03-26 | 上海擎昆信息科技有限公司 | Frequency offset compensation method and system |
| CN115442883A (en) * | 2022-11-07 | 2022-12-06 | 长沙驰芯半导体科技有限公司 | Fusion frequency offset correction method and device for ultra-wideband baseband |
Citations (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN101808068A (en) * | 2009-10-29 | 2010-08-18 | 清华大学 | Method and system for MSK iterative demodulation by combining LDPC code |
| CN102255865A (en) * | 2011-09-02 | 2011-11-23 | 东南大学 | Frame head sequence based channel estimating method for orthogonal frequency division multiplexing ultra-wideband system |
| CN102546500A (en) * | 2012-03-20 | 2012-07-04 | 西安电子科技大学 | SOQPSK (shaping offset quadrature phase shift keying) carrier synchronization method based on pilot frequency and soft information combined assistance |
| US8923457B2 (en) * | 2012-04-02 | 2014-12-30 | Nec Laboratories America, Inc. | Method and system for pilot-based time domain phase noise mitigation for coherent receiver |
| CN105763498A (en) * | 2016-04-08 | 2016-07-13 | 南京理工大学 | All-digital receiving method for satellite-borne AIS |
| CN107835035A (en) * | 2017-10-30 | 2018-03-23 | 北京理工大学 | The short frame burst communication open loop demodulation method of low signal-to-noise ratio and device |
| CN109714065A (en) * | 2017-10-25 | 2019-05-03 | 南京理工大学 | A kind of spaceborne AIS and ADS-B integrated receiver based on micro-nano satellite |
-
2020
- 2020-08-04 CN CN202010772113.9A patent/CN111865865B/en active Active
Patent Citations (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN101808068A (en) * | 2009-10-29 | 2010-08-18 | 清华大学 | Method and system for MSK iterative demodulation by combining LDPC code |
| CN102255865A (en) * | 2011-09-02 | 2011-11-23 | 东南大学 | Frame head sequence based channel estimating method for orthogonal frequency division multiplexing ultra-wideband system |
| CN102546500A (en) * | 2012-03-20 | 2012-07-04 | 西安电子科技大学 | SOQPSK (shaping offset quadrature phase shift keying) carrier synchronization method based on pilot frequency and soft information combined assistance |
| US8923457B2 (en) * | 2012-04-02 | 2014-12-30 | Nec Laboratories America, Inc. | Method and system for pilot-based time domain phase noise mitigation for coherent receiver |
| CN105763498A (en) * | 2016-04-08 | 2016-07-13 | 南京理工大学 | All-digital receiving method for satellite-borne AIS |
| CN109714065A (en) * | 2017-10-25 | 2019-05-03 | 南京理工大学 | A kind of spaceborne AIS and ADS-B integrated receiver based on micro-nano satellite |
| CN107835035A (en) * | 2017-10-30 | 2018-03-23 | 北京理工大学 | The short frame burst communication open loop demodulation method of low signal-to-noise ratio and device |
Non-Patent Citations (2)
| Title |
|---|
| 余孙全，陈利虎，李松亭，李兰民: "高灵敏度星载ADS-B信号解调算法", 《太赫兹科学与电子信息学报》 * |
| 刘贤锋: "基于微小卫星的海空目标监测与数据采集系统关键技术研究", 《中国优秀硕士学位论文全文数据库 工程科技Ⅱ辑》 * |
Cited By (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN112565129A (en) * | 2020-12-08 | 2021-03-26 | 上海擎昆信息科技有限公司 | Frequency offset compensation method and system |
| CN112565131A (en) * | 2020-12-08 | 2021-03-26 | 上海擎昆信息科技有限公司 | Frequency offset compensation method and system |
| CN115442883A (en) * | 2022-11-07 | 2022-12-06 | 长沙驰芯半导体科技有限公司 | Fusion frequency offset correction method and device for ultra-wideband baseband |
Also Published As
| Publication number | Publication date |
|---|---|
| CN111865865B (en) | 2021-06-15 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN111865865B (en) | Frequency offset and phase offset estimation method suitable for high-sensitivity satellite-borne ADS-B receiver | |
| CN106855628B (en) | Rapid acquisition and tracking system and method for high-dynamic satellite navigation signals | |
| CN109495410B (en) | High dynamic PCM/FM signal carrier frequency accurate estimation method | |
| CN109387813B (en) | High-precision frequency measurement method based on broadband radar signal reception | |
| US8619908B2 (en) | Wireless ranging system and related methods | |
| CN109490862B (en) | Carrier frequency estimation method based on phase difference statistical spectrum | |
| CN104022981A (en) | Blind carrier frequency deviation estimation method of quadrature amplitude modulation signal | |
| EP2615770B1 (en) | Fast acquisition of frame timing and frequency | |
| CN110161477B (en) | Maneuvering target detection method based on multi-variable resampling correlation function | |
| CN101630000B (en) | System for estimating influence of interference signal on GPS performance | |
| CN109682492B (en) | Frequency estimation method based on frequency domain Gaussian fitting | |
| CN108900445B (en) | Method and device for estimating signal symbol rate | |
| CN104901907B (en) | Data-aided steady signal-noise ratio estimation method is based under a kind of dynamic environment | |
| CN110708267A (en) | Method for determining frequency offset information estimation value | |
| CN114545342A (en) | Radar pulse signal parameter measuring method using multi-channel reconnaissance receiver | |
| CN113589253A (en) | Method for detecting weak echo signal based on wavelet transform algorithm of pseudo time domain | |
| CN113253313A (en) | Method for improving Doppler frequency estimation precision of PMF-FFT system | |
| CN111624632A (en) | Satellite navigation signal capturing method and device | |
| CN106777505A (en) | The frequency estimating methods and device of the robust of the undersampled signal based on frequency deviation identification | |
| CN115296698A (en) | High dynamic satellite communication system signal capturing method | |
| CN112630737B (en) | Preprocessing method for radar intermediate frequency echo signals | |
| CN113406672B (en) | Piecewise linear frequency modulation interference elimination method based on self-adaptive windowing | |
| CN112883787B (en) | Short sample low-frequency sinusoidal signal parameter estimation method based on spectrum matching | |
| CN105681240B (en) | High-precision frequency offset estimation method suitable for low signal-to-noise ratio environment | |
| CN108132460B (en) | Pulse compression compensation algorithm based on frequency domain channel equalization |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |