CN111711492A - Underwater acoustic communication symbol timing estimation and compensation method and device for deep submersible vehicle - Google Patents
Underwater acoustic communication symbol timing estimation and compensation method and device for deep submersible vehicle Download PDFInfo
- Publication number
- CN111711492A CN111711492A CN202010386629.XA CN202010386629A CN111711492A CN 111711492 A CN111711492 A CN 111711492A CN 202010386629 A CN202010386629 A CN 202010386629A CN 111711492 A CN111711492 A CN 111711492A
- Authority
- CN
- China
- Prior art keywords
- timing
- estimation
- symbol
- pilot
- waveform
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B13/00—Transmission systems characterised by the medium used for transmission, not provided for in groups H04B3/00 - H04B11/00
- H04B13/02—Transmission systems in which the medium consists of the earth or a large mass of water thereon, e.g. earth telegraphy
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/0202—Channel estimation
- H04L25/0222—Estimation of channel variability, e.g. coherence bandwidth, coherence time, fading frequency
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/0202—Channel estimation
- H04L25/0224—Channel estimation using sounding signals
Landscapes
- Engineering & Computer Science (AREA)
- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Power Engineering (AREA)
- Measurement Of Velocity Or Position Using Acoustic Or Ultrasonic Waves (AREA)
- Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
Abstract
The invention belongs to the field of underwater acoustic communication, and estimates the timing position of each sampling point by the operations of motion compensation of a fuzzy function, third-order spline interpolation, Farrow filtering sampling and the like. The invention takes the acceleration as the continuous change process as the assumed condition, realizes the separation estimation of timing and channel frequency, has short channel occupying time of the guide signal, excellent performance and low complexity, and lightens the burden of the equalizer. After the data frame is subjected to timing estimation and compensation, no matter the data frame adopts a single carrier transmission mode or an orthogonal frequency division multiplexing transmission mode, a receiving end needs to extract information of a transmission data symbol, only time invariant compensation is needed to be carried out on channel frequency response of the data frame, and channel tracking processing is not needed, so that the overall stability of the system is provided, the overhead of training symbols for channel estimation is reduced, the transmission efficiency is improved, and a basis is provided for application of a new communication system under an invariant channel model such as constellation probability forming transmission, energy expansion transformation and the like.
Description
Technical Field
The invention belongs to the field of underwater acoustic communication, and particularly relates to an underwater acoustic communication symbol timing estimation and compensation method, an underwater acoustic communication symbol timing estimation and compensation device, an underwater acoustic communication method, a receiving device and an underwater acoustic communication system based on the method, which can be used in scenes such as underwater acoustic communication between a deep-sea submersible and a mother ship.
Background
The underwater acoustic communication technology is the only means for the deep-sea submersible to carry out underwater remote wireless communication with the mother ship. Under the influence of sea surface surge fluctuation, the displacement of the mother ship is in quasi-periodic oscillation, so that high-speed underwater acoustic communication signals received by the mother ship have serious stretching and compression distortion in a time domain, and symbol timing time continuously jitters and changes, and serious Doppler frequency shift is caused.
Conventionally, the estimation and compensation method is to perform the resampling compensation at equal intervals after estimating the stretch and compression coefficients (hereinafter referred to as stretch coefficients) which are constant in a short time. The measuring method of the stretching coefficient is mainly divided into two methods of time variation of the guide before and after measurement and waveform stretching amount of the guide before measurement. For the method for measuring the time variation of the front and rear guidance, the relative speed of the front and rear guidance signals is considered to be unchanged, the front and rear guidance signals are matched with the local waveform, the time difference and the relative variation of the expected time difference are measured, the measurement performance of the time difference is insensitive to the movement speed, but the performance is degraded under the condition of acceleration; for the waveform stretching amount of the pilot before measurement, multiple copies of the expected waveform under different stretching coefficients need to be generated, the maximum value of the correlation between the expected waveform and the received waveform is searched, the stretching coefficient of the expected waveform at this moment is regarded as the stretching coefficient of the channel, and traversing the stretching coefficients causes huge calculation amount.
Meanwhile, the existing methods have the common disadvantages that the stretch coefficient is considered to be constant for a period of time, the time length is equal to the interval of the pilot signal, such as 0.5s, but the fluctuation period of the sea surface surge is generally about 7s, the constant of the stretch coefficient is assumed to have great deviation from the actual situation, and accurate timing information is not obtained, so that the current underwater acoustic communication needs to perform accurate time-varying channel compensation after doppler rough estimation: such as adaptive equalization and phase-locked loops in single-carrier communications, symbol-by-symbol doppler estimation or subcarrier interference cancellation based on pilot carriers or null carriers in orthogonal frequency division multiplexing systems (OFDM). This results in a decrease in channel utilization and an increase in signal to noise ratio requirements.
Disclosure of Invention
In view of the above, the present invention aims to reduce the complexity of the underwater acoustic communication timing estimation, reduce the pilot overhead, and improve the accuracy of the underwater acoustic communication timing estimation.
To solve the above technical problem, the present invention provides a symbol timing estimation and compensation method for underwater acoustic communication, wherein,
at a transmitting end, transmitting a superframe signal, wherein the superframe signal comprises a pilot signal and data symbols, the pilot signal is arranged at a plurality of time positions and is used for estimating the timing offset at the corresponding time positions, the data symbols are arranged between the pilot signals, blank intervals are left between the pilot signal and the data symbols, and the number of the pilot signals is N;
at the receiving end, the superframe signal is received, and the following processing is carried out:
step 1: quadrature mixing and decimation
Carrying out orthogonal frequency mixing on the sampled superframe signal waveform, extracting the waveform subjected to orthogonal frequency mixing to obtain a complex baseband waveform, wherein the extracted signal frequency is a baseband sampling rate, and separating the complex baseband waveform into a pilot signal waveform and a data symbol waveform according to the position relation of a pilot signal and a data symbol in a superframe;
step 2: matched filtering and peak position estimation of pilot signal waveforms
The starting position of the k-th pilot signal without timing offset is denoted as pkK is more than or equal to 1 and less than or equal to N; matching and filtering the pilot signal waveform and the local pilot signal, and obtaining the maximum matching peak position corresponding to each pilot signal according to the output result of the matching and filtering
Carrying out third-order spline interpolation of up-sampling rate on the output result of the matched filtering before and after each maximum matching peak position, improving the estimation precision of the maximum matching peak position, and obtaining N interpolated maximum matching peak positions
Using baseband sampling rate to obtain maximum matching peak position after interpolation
Converting into a first timing position estimated value d1(k) of the pilot signal, wherein the N first timing position estimated values sequentially form a first timing position estimated value sequence;
and step 3: doppler estimation and fuzzy function based timing correction
Performing third-order spline fitting on the first timing position estimation value sequence, then obtaining a derivative to obtain a Doppler stretching value v (k) corresponding to each guide signal, and obtaining a matching peak position correction d2(k) corresponding to each Doppler stretching value v (k) according to the fuzzy function property of the guide signal;
and 4, step 4: pilot signal waveform resampling and secondary detection estimation of peak position
Starting position pkThe timing estimation is that the sum of a first timing position estimation value d1(k) and a matching peak position correction value d2(k) is used as a constraint condition, the time offset of the resampling of the pilot signal waveform is obtained by utilizing third-order spline interpolation, then a Farrow filter resampling of time offset correction is carried out on the pilot signal waveform by using a Farrow filter, and the matching peak position estimation is carried out, so that the timing estimation offset is d3 (k);
and 5: timing combining of pilot waveforms and generation of timing sequences from sample point to sample point
Starting position pkThe timing estimation of (2) is the sum of the first timing position estimation value d1(k), the matching peak position correction quantity d2(k) and the timing estimation offset quantity d3(k) as a constraint condition, and the time offset of data symbol resampling is obtained by utilizing third-order spline interpolation.
The invention also provides a symbol timing estimation and compensation device for underwater acoustic communication, which is based on the method and is characterized by comprising the following steps:
the digital down-conversion extraction module is used for carrying out orthogonal frequency mixing on the superframe waveform after filtering and sampling, then extracting the waveform after frequency mixing to obtain a complex baseband waveform, wherein the signal frequency after extraction is the baseband sampling rate, and the complex baseband waveform is separated into two parts, namely a pilot signal waveform and a data symbol waveform, according to the position relation of a pilot signal and a data symbol in the superframe;
matched filterA sweep peak position estimation module for performing matched filtering on the pilot signal waveform and the local pilot signal and obtaining the maximum matched peak position corresponding to each pilot signal according to the output result of the matched filtering
Carrying out third-order spline interpolation of up-sampling rate on the output result of the matched filtering before and after each maximum matching peak position, improving the estimation precision of the maximum matching peak position, and obtaining N interpolated maximum matching peak positions
Using baseband sampling rate to obtain maximum matching peak position after interpolation
Converting into a first timing position estimated value d1(k) of the pilot signal, wherein the N first timing position estimated values sequentially form a first timing position estimated value sequence;
the Doppler estimation and fuzzy correction module is used for performing third-order spline fitting on the first timing position estimation value sequence, then obtaining a derivative to obtain a Doppler stretching value v (k) corresponding to each guide signal, and obtaining a matching peak position correction d2(k) corresponding to each Doppler stretching value v (k) according to the fuzzy function property of the guide signals;
a resampling and peak position estimating module for estimating the starting position pkThe timing estimation is that the sum of a first timing position estimation value d1(k) and a matching peak position correction value d2(k) is used as a constraint condition, the time offset of the resampling of the pilot signal waveform is obtained by utilizing third-order spline interpolation, then a Farrow filter resampling of time offset correction is carried out on the pilot signal waveform by using a Farrow filter, and the matching peak position estimation is carried out, so that the timing estimation offset is d3 (k);
a timing error combining module for combining the starting position pkThe timing estimation is the sum of a first timing position estimation value d1(k), a matching peak position correction d2(k) and a timing estimation offset d3(k) as a constraint condition, and a third-order spline is utilized to interpolateThe value obtains the time offset at which the data symbol is resampled.
The invention also provides an underwater acoustic communication method, which adopts the symbol timing estimation and compensation method to obtain the time offset of data symbol resampling; then also comprises the following steps:
step 6: data symbol resampling
Resampling the data symbol waveform of the complex baseband waveform by adopting a Farrow filter to obtain a data symbol resampling sequence; first N of data symbolsPilotEach symbol is a fixed pilot symbol sequence, the transmitting end and the receiving end are completely known, except the first NPilotThe symbols except the symbol are information symbol sequences;
and 7: channel estimation based on pilot symbol sequences
Using a pilot frequency symbol sequence part in the data symbol resampling sequence, and obtaining a channel estimation sequence by using a least square method;
and 8: time invariant equalization of information symbol sequences
And 7, obtaining a channel estimation sequence by using the step 7, and performing time-invariant equalization of linear minimum mean square error on the information symbol sequence part in the data symbol resampling sequence to obtain an information symbol sequence to finish the estimation of the symbol sequence.
A method for symbol timing estimation and compensation for underwater acoustic communications as claimed in claim 3 wherein the information symbols are modulated using quadrature phase shift keying.
The invention also provides an underwater acoustic communication receiving device, which is based on the underwater acoustic communication method, and is characterized by comprising a symbol timing estimation and compensation device for the underwater acoustic communication, and the device further comprises:
the resampling module is used for resampling the data symbol waveform of the complex baseband waveform by adopting a Farrow filter to obtain a data symbol resampling sequence; first N of data symbolsPilotEach symbol is a fixed pilot symbol sequence, the transmitting end and the receiving end are completely known, except the first NPilotThe symbols except the symbol are information symbol sequences;
a channel estimation module, which is used for obtaining a channel estimation sequence by using a pilot frequency symbol sequence part in a data symbol resampling sequence and utilizing a least square method;
and the equalization module is used for obtaining a channel estimation sequence by using the channel estimation module based on the pilot frequency symbol sequence, and performing time-invariant equalization of linear minimum mean square error on the information symbol sequence part in the data symbol resampling sequence to obtain an information symbol sequence and finish the estimation of the symbol sequence.
Further, the information symbols are modulated by using quadrature phase shift keying.
The invention also provides an underwater acoustic communication system, which is characterized by comprising a transmitting device and a receiving device, wherein,
transmitting means for transmitting a superframe signal, the superframe signal including pilot signals and data symbols, the pilot signals being disposed at a plurality of time positions and used for estimating timing offsets at the respective time positions, the data symbols being disposed between the pilot signals, blank intervals being left between the pilot signals and the data symbols, and the number of the pilot signals being N in total;
and the receiving device adopts the underwater acoustic communication receiving device.
Advantageous effects
The method estimates the timing position of each sampling point by the operations of motion compensation of a fuzzy function, third-order spline interpolation, Farrow filtering sampling and the like. The invention has the following advantages:
1. the transmitted pilot signal can use the common linear frequency modulation signal (LFM) and hyperbolic frequency modulation signal (HFM), and does not need to insert the synchronous waveform in the data symbol, and the pilot waveform occupies short channel time.
2. The short-time stretch coefficient is not assumed to be constant and is consistent with the real channel characteristics, and the timing estimation performance is excellent.
3. The receiving end only needs to carry out conventional matching on linear frequency modulation signals (LFM), does not need the generation of copies with different stretch coefficients and related calculation, and has low calculation complexity.
4. The timing and channel frequency are separately estimated, the burden of a subsequent equalizer is reduced, the structural complexity of the equalizer is simplified, and the overall performance of the system is improved. By using the time-varying separation technology in the invention, the transmitting end can utilize the time-invariant characteristic of the channel after timing compensation, and a foundation is provided for the application of a new communication system under the time-invariant channel model such as constellation probability forming transmission, energy expansion transformation and the like, and the technologies can not be directly used for a time-varying underwater acoustic channel well before the technology is provided by the invention.
Drawings
FIG. 1 is a superframe composition structure used in the embodiment of the present invention;
FIG. 2 is a block diagram of receive timing and equalization of the present invention;
FIG. 3 is a graph of timing estimates for direct Linear modulation (LFM) versus sample-by-sample timing estimates obtained by the method of the present invention;
fig. 4 is a comparison graph of the variation process of the strongest path tap coefficient obtained by using symbols for channel estimation;
fig. 5 is a graph of the phase change of a test waveform after conventional timing and the timing of the present invention, respectively, by time invariant keying.
FIG. 6 is a comparison of constellation diagrams after time invariant equalization of experimental waveforms after conventional timing compensation or timing compensation by the method of the present invention, respectively.
Detailed Description
The following describes in detail embodiments of the present invention with reference to the drawings.
The names, expressions and typical values of the parameters used in the embodiments described below are listed below:
step 1: quadrature mixing and decimation in a conventional manner
First, the received sampling waveform is subjected to spectrum shifting to reduce the sampling rate of the waveform. The method comprises performing quadrature mixing and extraction on sampled superframe waveform y (n), wherein n is more than or equal to 0 and less thanFpassTsuperframeThe quadrature-mixed waveform y1(n) is:
y1(n)＝Hilb{y(n)}exp{-j2πnFcarrier/Fpass}
wherein Hilb {. X } represents Hilbert transform, and components of the waveform on a negative frequency axis are removed, so that the purpose of inhibiting the negative axis frequency spectrum is achieved in the process of frequency spectrum moving. And then extracting the mixed waveform to a baseband sampling rate, wherein the extraction process comprises the following steps:
r(n)＝y1(nFpass/Fbase)
thus, r (n) (0. ltoreq. n < F)baseTsuperframe) Is a complex baseband waveform.
As shown in fig. 1, a superframe includes a plurality of pilot waveforms (in the present embodiment, LFM is used as a pilot waveform, or a waveform with time resolution such as a hyperbolic fm HFM) for completing timing estimation; and simultaneously, the system comprises a plurality of data frames carrying data information to be transmitted. Since the transmitting end follows this superframe structure, the complex baseband waveform is separated at the receiving end according to the position of the pilot and data waveforms.
Step 2: matched filtering and peak position estimation of pilot waveform LFM
A preliminary estimate of the time offset at each pilot location is obtained by means of a matched filtered peak location estimate from the received pilot waveform. For N in superframeLFMThe LFM signals are respectively matched and filtered with local LFM, and the kth (k is more than or equal to 1 and less than or equal to N)LFM) The starting position of the LFM waveform without timing deviation is pk＝TFRAME(k-1). c (l) is the complex baseband value of the ideal LFM waveform at time l, c*(l) Is its conjugate value. The result of the matching output is:
the position of the maximum matching peak is
At this time, the peak valueThe resolution of the position estimate is limited to the baseband sampling rate. In order to improve the estimation accuracy, the up-sampling rate processing of the waveform near the peak is required, and here, a third-order spline interpolation mode is adopted. In that
The third-order spline interpolation of 100 times of the up-sampling rate is carried out at 10 points on the left and the right of the time matching waveform, and the estimation precision of the peak position is improved. The position of the matching peak after interpolation is
Thus the preliminary estimate of the timing position of the kth LFM is
And step 3: doppler estimation and fuzzy function based timing correction
A third order spline fit is performed on the sequence d1(k), and the derivative is taken to obtain the doppler stretch value v (k) at LFM. According to the fuzzy function property of LFM, when the obtained stretching value is v (k), the correction quantity of the matching peak position is
And 4, step 4: pilot waveform resampling and secondary detection estimation of peak position
At pkThe timing of the time is estimated as d1(k) + d2(k), and according to the constraint relation, the time offset of the resampling of the pilot symbols is obtained by utilizing third-order spline interpolation.
After the precise timing offset is obtained, the received baseband waveform is resampled to compensate for the timing offset effect caused by the relative motion of the carrier. And (5) completing resampling by adopting a Farrow filtering mode. Compared with the resampling method of linear interpolation under high sampling rate, the Farrow filter has no higher requirement on the sampling rate and has good in-band flatness. Farrow filtering is designed as follows: pass band of 0.4FbaseLow pass filter design to obtain length KFarrowPolynomial order of LFarrowThe Farrow filtering of (2) performs Farrow filtering resampling for time offset correction on the baseband waveform r (n), and performs matching peak position estimation in step (2) to obtain an estimated timing offset d3 (k).
And 5: LFM timing combining and sample-by-sample timing sequence generation
At pkThe timing of the time is estimated as d1(k) + d2(k) + d3(k), and according to the constraint relation, the time offset of data symbol resampling is obtained by utilizing third-order spline interpolation.
Step 6: data symbol resampling
The data portion of the baseband waveform r (n) may be resampled using Farrow filtering to obtain a resampled sequence r2(n) using Farrow filtering, which may be the same Farrow filtering as in step 4.
And 7: channel estimation based on pilot symbol sequences
The first 200 symbols in the superframe data symbols are fixed pilot sequences, the transmitting end and the receiving end are completely known, and channel estimation h2(n) is obtained by using a least square method.
And 8: time invariant equalization
The symbol sequence in the super frame except the first 200 symbols is a quaternary phase shift keying modulated information symbol. And according to the channel estimation h2(n), performing time-invariant equalization of linear minimum mean square error on r2(n) to obtain an information symbol sequence s2(n), and finishing the estimation of the symbol sequence.
Based on the above method, as shown in fig. 1 and fig. 2, the underwater acoustic communication system of the embodiment of the present invention includes a transmitting device and a receiving device, wherein the transmitting device is configured to transmit a superframe signal, the superframe signal includes LFM pilot signals and data symbols, the LFM pilot signals are disposed at a plurality of time positions and are used for estimating timing offsets at the corresponding time positions, the data symbols are disposed between the pilot signals, a blank interval is left between the LFM pilot signals and the data symbols, and the pilot signals have N total pilot signalsLFMA plurality of;
the receiving device comprises a symbol timing estimation and compensation device, a resampling module, a channel estimation module and an equalization module, wherein the symbol timing estimation and compensation device for underwater acoustic communication comprises a digital down-conversion extraction module, a matched filtering and peak position estimation module, a Doppler estimation and fuzzy correction module, a resampling and peak position estimation module and a timing error combination module.
The method provided by the invention is used for processing a test waveform with the water depth of 5000 meters in 7-month flood dragon of 2011.
Fig. 3 is a sequence of timing estimates of the trial data output at step 5, and is compared to a timing estimate that is linearly interpolated after conventional LFM matching. Fig. 3 shows that the timing position at the LFM time is compensated by the method of the present invention in combination with the preliminary estimation of the doppler stretch coefficient, and the timing estimation curve is smoother by the method of the present invention using third-order spline interpolation, which is consistent with the acceleration continuity of the carrier motion in practice, while the discontinuity of the timing first-order derivative (representing the motion speed of the carrier) obtained by the conventional method of square estimation already occurs.
Fig. 4 is a diagram illustrating the fluctuation of the strongest path obtained by performing the least mean square adaptive estimation on the channel impulse response after the compensation by the conventional method and the method of the present invention. It can be seen that the channel tap coefficient is already stable after the processing by the method of the present invention, and the adaptive tracking or the phase-locked loop tracking processing is not needed.
Fig. 5 is a graph of the phase change of a test waveform after conventional timing and the timing process of the present invention, respectively, by time invariant equalized quadrature phase shift keying. Fig. 6 is a constellation comparison of a test waveform after time invariant equalization after conventional timing compensation or timing compensation by the method of the present invention, respectively. It can be seen that after the processing by the method of the present invention, the phase of the symbol does not drift after time invariant equalization, the four-phase shift keying feature of the constellation diagram is very obvious, and error-free transmission can be realized by simple error correction processing of 1/2 code rate; the traditional method has high symbol error rate after constant equalization and cannot realize correct transmission.
The timing estimation and compensation method in the prior art cannot completely compensate the timing offset caused by motion under the assumption that the velocity or acceleration is constant in a segmented manner, so that further tracking and compensation of the timing are required in subsequent data frames. The invention takes the acceleration as the continuous change process as the assumed condition, and Doppler compensation is carried out when the matching position of the guide signal is estimated, and the timing estimation of the invention is more accurate. After the data frame is subjected to the timing estimation and compensation, the channel becomes stable, no matter the data frame adopts a single carrier transmission mode or an orthogonal frequency division multiplexing transmission mode, a receiving end needs to further extract information of a transmission data symbol, only the channel frequency response of the data frame needs to be subjected to compensation in a time invariant mode, and channel tracking processing is not needed, so that the overall stability of the system is provided, the training symbol overhead for channel estimation is reduced, and the transmission efficiency is improved.
If the data frame adopts a single carrier transmission mode, channel estimation is carried out for one time according to a training symbol sequence, such as least square estimation, and then a time-invariant linear equalizer is designed according to a minimum mean square error criterion to complete the equalization and detection of subsequent information symbols.
If the data frame adopts an orthogonal frequency division multiplexing transmission mode, only the channel estimation pilot frequency is reserved in the first orthogonal frequency division multiplexing symbol in the data frame, the channel estimation result is obtained according to the pilot frequency information in the first symbol, the estimation mode can be least square channel estimation or minimum mean square error linear channel estimation, and then the maximum likelihood symbol detection is carried out on all data symbols in the super frame according to the channel estimation.
The method of the invention solves the problem of separation estimation of timing error and channel frequency selectivity for the first time, achieves the expected effect, and is applied to high-speed underwater acoustic communication of future full-sea deep submersible.
The present invention is not limited to the above preferred embodiments, and any modifications, equivalents, improvements, etc. made within the principle of the present invention are included in the scope of the present invention.
Claims (7)
1. A method for symbol timing estimation and compensation in underwater acoustic communications,
at a transmitting end, transmitting a superframe signal, wherein the superframe signal comprises a pilot signal and data symbols, the pilot signal is arranged at a plurality of time positions and is used for estimating the timing offset at the corresponding time positions, the data symbols are arranged between the pilot signals, blank intervals are left between the pilot signal and the data symbols, and the number of the pilot signals is N;
at the receiving end, the superframe signal is received, and the following processing is carried out:
step 1: quadrature mixing and decimation
Carrying out orthogonal frequency mixing on the sampled superframe signal waveform, extracting the waveform subjected to orthogonal frequency mixing to obtain a complex baseband waveform, wherein the extracted signal frequency is a baseband sampling rate, and separating the complex baseband waveform into a pilot signal waveform and a data symbol waveform according to the position relation of a pilot signal and a data symbol in a superframe;
step 2: matched filtering and peak position estimation of pilot signal waveforms
The starting position of the k-th pilot signal without timing offset is denoted as pkK is more than or equal to 1 and less than or equal to N; matching and filtering the pilot signal waveform and the local pilot signal, and obtaining the maximum matching peak position corresponding to each pilot signal according to the output result of the matching and filtering
Carrying out third-order spline interpolation of up-sampling rate on the output result of the matched filtering before and after each maximum matching peak position, improving the estimation precision of the maximum matching peak position, and obtaining N interpolated maximum matching peak positions
Using baseband sampling rate to obtain maximum matching peak position after interpolation
Converting into a first timing position estimated value d1(k) of the pilot signal, wherein the N first timing position estimated values sequentially form a first timing position estimated value sequence;
and step 3: doppler estimation and fuzzy function based timing correction
Performing third-order spline fitting on the first timing position estimation value sequence, then obtaining a derivative to obtain a Doppler stretching value v (k) corresponding to each guide signal, and obtaining a matching peak position correction d2(k) corresponding to each Doppler stretching value v (k) according to the fuzzy function property of the guide signal;
and 4, step 4: pilot signal waveform resampling and secondary detection estimation of peak position
Starting position pkThe timing estimation is that the sum of a first timing position estimation value d1(k) and a matching peak position correction value d2(k) is used as a constraint condition, the time offset of the resampling of the pilot signal waveform is obtained by utilizing third-order spline interpolation, then a Farrow filter resampling of time offset correction is carried out on the pilot signal waveform by using a Farrow filter, and the matching peak position estimation is carried out, so that the timing estimation offset is d3 (k);
and 5: timing combining of pilot waveforms and generation of timing sequences from sample point to sample point
Starting position pkThe timing estimation of (2) is the sum of the first timing position estimation value d1(k), the matching peak position correction quantity d2(k) and the timing estimation offset quantity d3(k) as a constraint condition, and the time offset of data symbol resampling is obtained by utilizing third-order spline interpolation.
2. A symbol timing estimation and compensation apparatus for underwater acoustic communication based on the method of claim 1, comprising:
the digital down-conversion extraction module is used for carrying out orthogonal frequency mixing on the superframe waveform after filtering and sampling, then extracting the waveform after frequency mixing to obtain a complex baseband waveform, wherein the signal frequency after extraction is the baseband sampling rate, and the complex baseband waveform is separated into two parts, namely a pilot signal waveform and a data symbol waveform, according to the position relation of a pilot signal and a data symbol in the superframe;
a matched filtering and peak position estimating module for performing matched filtering on the pilot signal waveform and the local pilot signal and obtaining the maximum matched peak position corresponding to each pilot signal according to the output result of the matched filtering
Carrying out third-order spline interpolation of up-sampling rate on the output result of the matched filtering before and after each maximum matching peak position, improving the estimation precision of the maximum matching peak position, and obtaining N interpolated maximum matching peak positions
Using baseband sampling rate to obtain maximum matching peak position after interpolation
Converting into a first timing position estimated value d1(k) of the pilot signal, wherein the N first timing position estimated values sequentially form a first timing position estimated value sequence;
the Doppler estimation and fuzzy correction module is used for performing third-order spline fitting on the first timing position estimation value sequence, then obtaining a derivative to obtain a Doppler stretching value v (k) corresponding to each guide signal, and obtaining a matching peak position correction d2(k) corresponding to each Doppler stretching value v (k) according to the fuzzy function property of the guide signals;
a resampling and peak position estimating module for estimating the starting position pkThe timing estimation is that the sum of a first timing position estimation value d1(k) and a matching peak position correction value d2(k) is used as a constraint condition, the time offset of the resampling of the pilot signal waveform is obtained by utilizing third-order spline interpolation, then a Farrow filter resampling of time offset correction is carried out on the pilot signal waveform by using a Farrow filter, and the matching peak position estimation is carried out, so that the timing estimation offset is d3 (k);
a timing error combining module for combining the starting position pkThe timing estimation of (2) is the sum of the first timing position estimation value d1(k), the matching peak position correction quantity d2(k) and the timing estimation offset quantity d3(k) as a constraint condition, and the time offset of data symbol resampling is obtained by utilizing third-order spline interpolation.
3. A method of underwater acoustic communication, wherein the symbol timing estimation and compensation method according to claim 1 is used to obtain the time offset of data symbol resampling; then also comprises the following steps:
step 6: data symbol resampling
Resampling the data symbol waveform of the complex baseband waveform by adopting a Farrow filter to obtain a data symbol resampling sequence; first N of data symbolsPilotEach symbol is a fixed pilot symbol sequence, the transmitting end and the receiving end are completely known, except the first NPilotThe symbols except the symbol are information symbol sequences;
and 7: channel estimation based on pilot symbol sequences
Using a pilot frequency symbol sequence part in the data symbol resampling sequence, and obtaining a channel estimation sequence by using a least square method;
and 8: time invariant equalization of information symbol sequences
And 7, obtaining a channel estimation sequence by using the step 7, and performing time-invariant equalization of linear minimum mean square error on the information symbol sequence part in the data symbol resampling sequence to obtain an information symbol sequence to finish the estimation of the symbol sequence.
4. A method for symbol timing estimation and compensation for underwater acoustic communications as claimed in claim 3 wherein the information symbols are modulated using quadrature phase shift keying.
5. A receiving apparatus of underwater acoustic communication based on the method of claim 3, characterized by comprising the symbol timing estimation and compensation apparatus of underwater acoustic communication of claim 2, further comprising:
the resampling module is used for resampling the data symbol waveform of the complex baseband waveform by adopting a Farrow filter to obtain a data symbol resampling sequence; first N of data symbolsPilotEach symbol is a fixed pilot symbol sequence, the transmitting end and the receiving end are completely known, except the first NPilotThe symbols except the symbol are information symbol sequences;
a channel estimation module, which is used for obtaining a channel estimation sequence by using a pilot frequency symbol sequence part in a data symbol resampling sequence and utilizing a least square method;
and the equalization module is used for obtaining a channel estimation sequence by using the channel estimation module based on the pilot frequency symbol sequence, and performing time-invariant equalization of linear minimum mean square error on the information symbol sequence part in the data symbol resampling sequence to obtain an information symbol sequence and finish the estimation of the symbol sequence.
6. The underwater acoustic communication receiving apparatus of claim 5, wherein the information symbols are modulated using quadrature phase shift keying.
7. An underwater acoustic communication system is characterized by comprising a transmitting device and a receiving device,
transmitting means for transmitting a superframe signal, the superframe signal including pilot signals and data symbols, the pilot signals being disposed at a plurality of time positions and used for estimating timing offsets at the respective time positions, the data symbols being disposed between the pilot signals, blank intervals being left between the pilot signals and the data symbols, and the number of the pilot signals being N in total;
receiving apparatus using the underwater acoustic communication receiving apparatus according to claim 5 or 6.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202010386629.XA CN111711492B (en) | 2020-05-09 | 2020-05-09 | Underwater acoustic communication symbol timing estimation and compensation method and device for deep submersible vehicle |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202010386629.XA CN111711492B (en) | 2020-05-09 | 2020-05-09 | Underwater acoustic communication symbol timing estimation and compensation method and device for deep submersible vehicle |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN111711492A true CN111711492A (en) | 2020-09-25 |
| CN111711492B CN111711492B (en) | 2021-07-20 |
Family
ID=72536893
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202010386629.XA Active CN111711492B (en) | 2020-05-09 | 2020-05-09 | Underwater acoustic communication symbol timing estimation and compensation method and device for deep submersible vehicle |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN111711492B (en) |
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN112737996A (en) * | 2020-12-25 | 2021-04-30 | 中国科学院声学研究所 | Time-varying timing offset iterative estimation and compensation method and device for underwater acoustic communication |
| CN114285495A (en) * | 2021-12-30 | 2022-04-05 | 电子科技大学 | Underwater sound OFDM communication system received signal processing method based on message transmission |
| CN116319187A (en) * | 2023-02-20 | 2023-06-23 | 中国人民解放军军事科学院系统工程研究院 | Data processing method and device for satellite Internet of things system |
| CN116846725A (en) * | 2023-08-31 | 2023-10-03 | 为准（北京）电子科技有限公司 | Sampling clock deviation compensation method and device for wireless signals |
Citations (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN104796370A (en) * | 2015-03-20 | 2015-07-22 | 中国电子科技集团公司第三研究所 | Signal synchronization method and system for underwater acoustic communication and underwater acoustic communication system |
| CN104836759A (en) * | 2015-05-18 | 2015-08-12 | 西北工业大学 | Robust underwater acoustic channel estimation method based on joint norm optimization |
| KR101580427B1 (en) * | 2015-01-29 | 2015-12-28 | 한국해양대학교 산학협력단 | Doppler Frequency Estimation and receiving method for Time-varying Underwater Acoustic Communication Channel |
| CN108243138A (en) * | 2018-01-11 | 2018-07-03 | 福建星海通信科技有限公司 | A kind of combination Doppler estimation suitable for underwater sound communication system |
| CN108566354A (en) * | 2018-04-03 | 2018-09-21 | 哈尔滨工程大学 | DPFFT time-varying broadband Doppler Compensation Method in underwater sound OFDM |
| CN109450489A (en) * | 2018-08-30 | 2019-03-08 | 中国船舶重工集团公司第七〇五研究所 | A kind of pilot frequency sequence interference cancellation method of spread-spectrum underwater sound communication |
| CN109743118A (en) * | 2018-12-24 | 2019-05-10 | 哈尔滨工程大学 | A kind of double OFDM underwater acoustic communication methods for expanding the spectral efficient under channel condition of time-varying |
| CN109802911A (en) * | 2019-01-21 | 2019-05-24 | 浙江大学 | A kind of Fast Channel estimation and signal synchronizing method suitable for underwater acoustic modem |
| CN111106877A (en) * | 2019-12-11 | 2020-05-05 | 中国科学院声学研究所 | Underwater acoustic communication transmission method based on Farrow filtering and code word matching |
-
2020
- 2020-05-09 CN CN202010386629.XA patent/CN111711492B/en active Active
Patent Citations (9)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR101580427B1 (en) * | 2015-01-29 | 2015-12-28 | 한국해양대학교 산학협력단 | Doppler Frequency Estimation and receiving method for Time-varying Underwater Acoustic Communication Channel |
| CN104796370A (en) * | 2015-03-20 | 2015-07-22 | 中国电子科技集团公司第三研究所 | Signal synchronization method and system for underwater acoustic communication and underwater acoustic communication system |
| CN104836759A (en) * | 2015-05-18 | 2015-08-12 | 西北工业大学 | Robust underwater acoustic channel estimation method based on joint norm optimization |
| CN108243138A (en) * | 2018-01-11 | 2018-07-03 | 福建星海通信科技有限公司 | A kind of combination Doppler estimation suitable for underwater sound communication system |
| CN108566354A (en) * | 2018-04-03 | 2018-09-21 | 哈尔滨工程大学 | DPFFT time-varying broadband Doppler Compensation Method in underwater sound OFDM |
| CN109450489A (en) * | 2018-08-30 | 2019-03-08 | 中国船舶重工集团公司第七〇五研究所 | A kind of pilot frequency sequence interference cancellation method of spread-spectrum underwater sound communication |
| CN109743118A (en) * | 2018-12-24 | 2019-05-10 | 哈尔滨工程大学 | A kind of double OFDM underwater acoustic communication methods for expanding the spectral efficient under channel condition of time-varying |
| CN109802911A (en) * | 2019-01-21 | 2019-05-24 | 浙江大学 | A kind of Fast Channel estimation and signal synchronizing method suitable for underwater acoustic modem |
| CN111106877A (en) * | 2019-12-11 | 2020-05-05 | 中国科学院声学研究所 | Underwater acoustic communication transmission method based on Farrow filtering and code word matching |
Non-Patent Citations (2)
| Title |
|---|
| ZHU WEIGING， WU YANBO等: "Signal processing in underwater acoustic communication system for manned deep submersible "Jiaolong"", 《INSTITUTE OF ACOUSTICS》 * |
| 朱敏等: "蛟龙号载人潜水器声学系统", 《科学通报》 * |
Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN112737996A (en) * | 2020-12-25 | 2021-04-30 | 中国科学院声学研究所 | Time-varying timing offset iterative estimation and compensation method and device for underwater acoustic communication |
| CN112737996B (en) * | 2020-12-25 | 2022-03-25 | 中国科学院声学研究所 | Time-varying timing offset iterative estimation and compensation method and device for underwater acoustic communication |
| CN114285495A (en) * | 2021-12-30 | 2022-04-05 | 电子科技大学 | Underwater sound OFDM communication system received signal processing method based on message transmission |
| CN116319187A (en) * | 2023-02-20 | 2023-06-23 | 中国人民解放军军事科学院系统工程研究院 | Data processing method and device for satellite Internet of things system |
| CN116319187B (en) * | 2023-02-20 | 2023-10-17 | 中国人民解放军军事科学院系统工程研究院 | Data processing method and device for satellite Internet of things system |
| CN116846725A (en) * | 2023-08-31 | 2023-10-03 | 为准（北京）电子科技有限公司 | Sampling clock deviation compensation method and device for wireless signals |
| CN116846725B (en) * | 2023-08-31 | 2023-11-10 | 为准（北京）电子科技有限公司 | Sampling clock deviation compensation method and device for wireless signals |
Also Published As
| Publication number | Publication date |
|---|---|
| CN111711492B (en) | 2021-07-20 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN111711492B (en) | Underwater acoustic communication symbol timing estimation and compensation method and device for deep submersible vehicle | |
| CN107911133B (en) | A kind of the Doppler factor estimation and compensation method of mobile underwater sound communication | |
| WO2019119778A1 (en) | Anti-interference signal detection and synchronization method for wireless broadband communication system | |
| US7388922B2 (en) | Receiver | |
| CN101547174B (en) | Method for phase and symbol synchronization, channel estimation and frequency domain equalization of SC-FDE system | |
| US8229011B2 (en) | Fine symbol timing synchronization method and apparatus in OFDM system | |
| CN111683034B (en) | OFDM-based large Doppler wireless communication time-frequency synchronization method | |
| CN100448184C (en) | Receiver of quadrature frequency division multiplexing system | |
| CN108768604B (en) | Low-complexity bit synchronization method for PCM/FM multi-symbol detection | |
| CN113259291B (en) | Phase compensation method realized by dynamic Doppler tracking of underwater sound continuous signals | |
| CN113259295A (en) | Signal detection method for underwater acoustic FBMC system | |
| CN101404633B (en) | Carrier wave tracing method for single carrier system based on block transmission | |
| CN109194594A (en) | A kind of phase noise inhibition method based on continuous carrier polymerization | |
| CN113438730B (en) | Wireless positioning method based on GFDM signal | |
| CN116073892B (en) | Demodulation method and device for service channel of low-orbit satellite communication system | |
| EP0993159B1 (en) | Estimation of carrier frequency offset | |
| CN107276654B (en) | Signal processing method and system | |
| CN107276953B (en) | Timing synchronization method, device and system | |
| US6845133B2 (en) | Method and device for processing signals of a digital transmission system | |
| Liu et al. | Non-Uniform Doppler Compensation Method for Staggered Multitone Filter Bank Multicarrier in the Underwater Acoustic Channel | |
| CN107276940B (en) | Timing synchronization method, device and system | |
| CN107276955B (en) | Signal processing method and system | |
| CN107294889B (en) | Carrier synchronization method and device | |
| CN107276943B (en) | Timing synchronization method, device and system | |
| CN107276708B (en) | Timing synchronization method, device and system |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |