CN111669091B - Direct torque control method for motor - Google Patents

Direct torque control method for motor Download PDF

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Publication number
CN111669091B
CN111669091B CN201910178266.8A CN201910178266A CN111669091B CN 111669091 B CN111669091 B CN 111669091B CN 201910178266 A CN201910178266 A CN 201910178266A CN 111669091 B CN111669091 B CN 111669091B
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plane
sub
stator
harmonic
flux linkage
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CN111669091A (en
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李云
诸自强
任远
马雅青
朱世武
詹姆斯·格林
李子健
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Zhuzhou CRRC Times Semiconductor Co Ltd
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Zhuzhou CRRC Times Semiconductor Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2203/00—Indexing scheme relating to controlling arrangements characterised by the means for detecting the position of the rotor
    • H02P2203/03—Determination of the rotor position, e.g. initial rotor position, during standstill or low speed operation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2205/00—Indexing scheme relating to controlling arrangements characterised by the control loops
    • H02P2205/01—Current loop, i.e. comparison of the motor current with a current reference

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  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention provides a direct torque control method of a motor, which obtains a voltage space vector set value related to torque in a double three-phase permanent magnet synchronous motor system by a method of compensating a stator flux linkage vector angle by carrying out proportional integral regulation on a torque error, reduces harmonic voltage caused by inherent asymmetry of a motor and an inverter in the double three-phase permanent magnet synchronous motor system by a proportional integral controller and a double frequency resonance controller, and further effectively inhibits harmonic current generated by the harmonic voltage. Harmonic voltage caused by non-linear factors of an inverter in a double three-phase permanent magnet synchronous motor system is reduced through a six-frequency-multiplication resonance controller, harmonic current generated by the harmonic voltage is effectively restrained, extra motor parameters are not introduced in the calculation process of a reference voltage space vector, and robustness of the motor parameters is effectively enhanced.

Description

Direct torque control method for motor
Technical Field
The invention belongs to the technical field of motor control, and particularly relates to a motor direct torque control method applicable to a double three-phase permanent magnet motor system.
Background
The multi-phase motor alternating current transmission system has the advantages of low torque pulsation, good fault tolerance performance, more control freedom degree and the like. Particularly five-phase and six-phase motor systems. The double three-phase permanent magnet motor system is a six-phase motor system and comprises two sets of stator windings with the difference of 30-degree electrical angles, and the two sets of stator windings can be respectively supplied with power by adopting independent three-phase inverters. Therefore, the double three-phase permanent magnet motor system has the advantages of both the traditional permanent magnet motor and the multi-phase motor, and is widely researched and applied in recent years.
The double three-phase permanent magnet motor is a high-order and strong coupling system under a traditional six-phase coordinate system. Through six-dimensional decoupling coordinate transformation, voltage, current and flux linkage vectors of the double three-phase permanent magnet motor can be projected into three mutually orthogonal two-dimensional sub-planes. Wherein, the electromechanical energy conversion component is projected to alpha-beta sub-plane, and the non-electromechanical energy conversion component is projected to z1-z2Sub-plane and o1-o2A sub-plane. For two sets of double three-phase permanent magnet motor systems with independent winding neutral points, o1-o2The voltage vector of the sub-plane is a zero vector and therefore can be disregarded for control.
In a dual three-phase permanent magnet motor system, although only the voltage and current components of the α - β sub-plane are relevant to electromechanical energy conversion (i.e., torque formation), the z-direction is not the same1-z2The voltage and current components of the sub-planes are effectively controlled, even for small harmonic voltages in z1-z2The sub-planes form larger harmonic currents, which easily causes the heating of the motor to be increased and the system efficiency to be reduced. This feature of the dual three-phase permanent magnet motor system determines that two problems must be considered simultaneously when controlling the dual three-phase motor system: effective control of torque and effective suppression of harmonic currents.
Harmonic current at z from harmonic voltage1-z2Acting on the sub-plane. The formation of harmonic voltages is mainly due to three factors:
1) a method for generating an inappropriate inverter switching signal.
2) Asymmetry of the motor and the inverter themselves, such as phase-to-phase asymmetry of each set of windings of the motor or asymmetry between two sets of windings.
3)The nonlinear factor of the inverter is mainly dead zone action, wherein the dead zone can be regarded as a square wave voltage signal, and the signal is applied to z after decomposition1-z2The main components on the sub-planes are the 5 th and 7 th voltage harmonics.
Therefore, effective suppression of the harmonics of the dual three-phase permanent magnet motor system needs to be performed while considering the above three factors.
The direct torque control of the traditional double three-phase permanent magnet synchronous motor is expanded from the direct torque control of a three-phase permanent magnet motor system, namely, a proper voltage space vector is selected from an off-line optimal switch table to control the state of a switch tube of an inverter according to output signals of a flux linkage hysteresis comparator and a torque hysteresis comparator and position signals of a stator flux linkage. Although the method has simple structure and easy implementation, the hysteresis comparator and the switching table in the traditional direct torque control are adopted to generate the inverter switching signal, so that great torque pulsation is caused, and only the voltage component of the alpha-beta sub-plane is controlled, z1-z2The voltage component of the sub-plane can generate larger harmonic current, so that the motor loss is increased, and the system efficiency is reduced.
The deadbeat direct torque control based on Sine Pulse Width Modulation (SPWM) of the double three-phase permanent magnet synchronous motor is to directly obtain a voltage reference vector from a flux linkage error and a torque error according to a motor model equation and output a switch tube state control signal of an inverter through an SPWM technology. Although the method adopts a Pulse Width Modulation (PWM) strategy, the inherent asymmetry of the motor and the inverter is not considered, and the nonlinear factor of the inverter is not considered, so that more obvious current harmonics exist. At the same time, inaccuracies in the motor parameters may result in inaccuracies in the torque control, due to the strong dependence of the selected control strategy on the motor parameters.
In view of the above, a new direct torque control method for a motor is needed to solve the above technical problems.
Disclosure of Invention
In view of the above technical problems, the present invention provides a new method for controlling direct torque of a motor, so as to solve the technical problem that the prior art cannot effectively control the torque of a dual three-phase permanent magnet synchronous motor system and effectively suppress harmonic current.
The technical scheme of the invention is realized by the following modes:
a method of direct torque control of an electric machine, comprising the steps of:
s10: six-phase stator voltage signals and six-phase stator current signals of the double three-phase motor are obtained, and stator voltage components, stator current components and z of alpha-beta sub-planes are obtained by performing coordinate transformation on the obtained signals1-z2A stator current component of the harmonic sub-plane; acquiring the rotor speed and the rotor position of the motor;
s20: estimating a stator flux linkage, a stator flux linkage position, and an electromagnetic torque from the stator voltage component and the stator current component of the α - β sub-plane obtained in step S10;
s30: determining a given value of the electromagnetic torque by performing proportional integral calculation on the difference between the given rotor speed and the rotor speed obtained in step S10, and further determining a given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque; estimating a given value of a stator flux linkage amplitude corresponding to the rotor speed according to the direct-current bus voltage of the inverter and the rotor speed obtained in the step S10; taking the smaller value of the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque and the given value of the stator flux linkage amplitude corresponding to the rotor rotation speed as the reference value of the stator flux linkage amplitude;
s40: determining an angle increment of the stator flux linkage by performing a proportional integral calculation on a difference between a given value of the electromagnetic torque and the calculated value of the electromagnetic torque obtained in step S20, determining a reference value of the stator flux linkage according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in step S30, and the calculated value of the position of the stator flux linkage obtained in step S20, and determining a given value of a voltage space vector of the α - β sub-plane according to the reference value of the stator flux linkage, the calculated value of the stator flux linkage obtained in step S20, and a stator current component of the α - β sub-plane obtained in step S10;
s50: z obtained based on step S101-z2Stator current component of harmonic sub-plane, obtaining z by harmonic current controller1-z2A given value of the voltage space vector of the harmonic sub-plane;
s60: using space vector pulse width modulation technique to obtain given value of voltage space vector of alpha-beta sub-plane obtained in step S40 and z obtained in step S501-z2The given value of the voltage space vector of the harmonic sub-plane is modulated to generate a switching signal for controlling a switching tube of the inverter, so that the direct torque control of the motor is realized;
the harmonic current controller is configured to reduce harmonic voltages caused by inherent asymmetry of the motor and the inverter and harmonic voltages caused by nonlinear linear factors of the inverter, and further suppress harmonic currents generated by the harmonic voltages.
According to the first embodiment of the present invention, the step S50 further includes the steps of:
s51, based on the rotor position obtained in step S10, for z obtained in step S101-z2Stator current components of the harmonic sub-planes are subjected to synchronous rotation coordinate transformation to obtain z1-z2Stator current components under a harmonic sub-plane synchronous rotation coordinate system;
s52, based on z1-z2The condition that the reference value of the stator current component is zero in the harmonic sub-plane synchronous rotation coordinate system is based on z obtained in step S511-z2Obtaining the stator current component in the harmonic sub-plane synchronous rotation coordinate system by using a harmonic current controller1-z2Given values of voltage space vectors under a sub-plane synchronous rotation coordinate system;
s53, for z obtained in step S521-z2Carrying out synchronous rotation coordinate inverse transformation on a given value of a voltage space vector under a sub-plane synchronous rotation coordinate system to obtain z1-z2A given value of the voltage space vector of the harmonic sub-plane;
the harmonic current controller comprises a proportional integral resonance regulator and a six-frequency-doubling resonance regulator, and the resonance regulator of the proportional integral resonance regulator is a frequency-doubling resonance regulator.
According to the second embodiment of the present invention, the step S50 further includes the steps of:
s51, based on the rotor position obtained in step S10, for z obtained in step S101-z2The stator current component of the harmonic sub-plane is subjected to forward rotation coordinate transformation and reverse rotation coordinate transformation to obtain z1-z2Stator current components under a harmonic sub-plane forward rotation coordinate system and stator current components under a harmonic sub-plane reverse rotation coordinate system;
s52, based on z1-z2The condition that the reference value of the stator current component in the harmonic sub-plane forward rotation coordinate system is zero is based on z obtained in step S511-z2The stator current component under the harmonic sub-plane forward rotation coordinate system is obtained by a proportional-integral regulator and a six-time frequency resonance regulator in the harmonic current controller1-z2A given value of a voltage space vector under a harmonic sub-plane forward rotation coordinate system;
s53, based on z1-z2On condition that the reference value of the stator current component in the harmonic sub-plane reverse rotation coordinate system is zero, z is obtained from step S511-z2Obtaining a stator current component in a harmonic sub-plane reverse rotation coordinate system by using another proportional-integral regulator in the harmonic current controller1-z2A given value of a voltage space vector under a harmonic sub-plane reverse rotation coordinate system;
s54, for z obtained in step S521-z2Performing reverse transformation of forward rotation coordinate on the given value of the voltage space vector in the sub-plane forward rotation coordinate system and performing reverse transformation on the z obtained in the step S531-z2Carrying out reverse rotation coordinate inverse transformation on the given value of the voltage space vector under the sub-plane reverse rotation coordinate system, and adding the inverse transformation results of the given value and the voltage space vector to obtain z1-z2A given value of the voltage space vector of the harmonic sub-plane.
According to the third embodiment of the present invention, the step S50 specifically includes:
based on z1-z2The condition that the reference value of the harmonic sub-plane stator current component is zero is based on z obtained in step S101-z2Stator current component of harmonic sub-plane, obtaining z by harmonic current controller1-z2A given value of the voltage space vector of the sub-plane;
wherein the harmonic current controller comprises a proportional and fundamental resonance adjuster, a quintuple frequency resonance adjuster and a heptatuple frequency resonance adjuster.
According to an embodiment of the present invention, after obtaining the given value of the electromagnetic torque, the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque is determined by looking up the flux linkage-torque table obtained according to the maximum torque current ratio algorithm in step S30.
According to an embodiment of the present invention, in the step S20, the stator flux linkage ψ is calculated from the stator voltage component and the stator current component of the α - β sub-plane obtained in step S10 by the following equationsαβStator flux linkage position thetasAnd electromagnetic torque Te：
Figure BDA0001989438480000051
Figure BDA0001989438480000052
Te＝3P(ψsαisβ-ψsβisα)
Wherein u issα，usβComponents of the stator voltage in the alpha-beta sub-plane on the alpha and beta axes, isα，isβComponents of the stator current in the alpha-beta sub-plane in the alpha-beta axis and beta axis, psisα，ψsβThe components of the stator flux linkage in the α - β sub-plane, the α -axis and the β -axis, respectively, RsIs the stator resistance of the motor, and P is the number of pole pairs of the motor.
According to the embodiment of the present invention, in the step S30, the given value of the stator flux linkage amplitude corresponding to the rotor speed is estimated from the dc bus voltage of the inverter and the rotor speed obtained in the step S10 by the following formula:
|ψs2 *|＝Umax/ωr
Figure BDA0001989438480000053
wherein, | ψs2 *I is the calculated value of stator flux linkage amplitude, omegarAs the rotational speed of the rotor, VdcIs the DC bus voltage of the inverter, UmaxThe maximum amplitude of the phase voltage allowed to be provided by the inverter, η, is a positive coefficient smaller than 1 and close to 1.
According to the embodiment of the present invention, in the step S40, the reference value of the stator flux linkage is determined according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in the step S30, and the calculated value of the position of the stator flux linkage obtained in the step S20 by the following formula:
Figure BDA0001989438480000054
wherein the content of the first and second substances,
Figure BDA0001989438480000055
the components of the reference value of the stator flux linkage in the alpha and beta axes in the alpha-beta sub-plane respectively,
Figure BDA0001989438480000056
is the angle increment of the stator flux linkage,
Figure BDA0001989438480000057
Is a reference value of stator flux linkage amplitude and thetasIs the stator flux linkage position.
According to an embodiment of the present invention, in the step S40, the given value of the voltage space vector of the α - β sub-plane is determined according to the reference value of the stator flux linkage and the calculated value of the stator flux linkage obtained in the step S20 and the stator current component of the α - β sub-plane obtained in the step S10 by the following formula:
Figure BDA0001989438480000061
wherein the content of the first and second substances,
Figure BDA0001989438480000062
components of a given value of the voltage space vector of the alpha-beta sub-plane on the alpha and beta axes, i, respectively, in the alpha-beta sub-planesα，isβThe components of the stator current in the alpha-beta sub-plane on the alpha and beta axes, RsIs the stator resistance, T, of the motorsIs the sampling period of the system.
According to an embodiment of the present invention, in the step S50,
by the formula to z1-z2Stator current component i of the harmonic sub-planesz1，isz2Performing synchronous rotation coordinate transformation to obtain z1-z2Stator current component i under harmonic sub-plane synchronous rotation coordinate systemszd，iszq：
Figure BDA0001989438480000063
By the formula to z1-z2The given value of the voltage space vector under the harmonic sub-plane synchronous rotation coordinate system is subjected to synchronous rotation coordinate inverse transformation to obtain z1-z2Given value of the voltage space vector of the harmonic sub-plane:
Figure BDA0001989438480000064
wherein v isszd *，vszq *Are each z1-z2Of voltage space vectors in a harmonic sub-plane synchronous rotating coordinate systemComponent of given value, vsz1 *，vsz2 *Are each z1-z2Component of a given value of the voltage space vector of the harmonic sub-plane, thetarIs the rotor position.
Compared with the prior art, the direct torque control method of the motor provided by the invention has the following advantages or beneficial effects:
1) the large torque ripple in the traditional direct torque control is effectively reduced.
The method for compensating the stator flux linkage vector angle by carrying out proportional integral adjustment on the torque error is utilized to obtain the given value of the voltage space vector related to the torque, and the switching control of the inverter is realized by combining the Space Vector Pulse Width Modulation (SVPWM) technology, so that the large torque pulsation in the traditional direct torque control is effectively reduced.
2) Effective suppression of harmonic currents
Harmonic voltage caused by inherent asymmetry of a motor and an inverter in a double three-phase permanent magnet synchronous motor system is reduced by utilizing a proportional-integral regulator, a double-frequency resonance regulator and the like, and harmonic current generated by the harmonic voltage can be effectively inhibited.
Harmonic voltage caused by non-linear factors (mainly dead zone influence) of the inverter in a double three-phase permanent magnet synchronous motor system is reduced by utilizing the six-frequency multiplication resonance regulator, and further harmonic current generated by the harmonic voltage can be effectively inhibited.
In summary, by simultaneously considering three main factors (improper inverter switching signal generation method, asymmetry of the motor and the inverter, inverter nonlinearity factor) forming harmonic current, the harmonic current is effectively suppressed.
3) The dependence of a control strategy on motor parameters is effectively reduced.
The voltage space vector related to the torque is obtained by a method of compensating a stator flux linkage vector angle by carrying out Proportional Integral (PI) adjustment on the torque error, and no additional motor parameter is needed except for all stator resistors in a flux linkage observer needed by direct torque control. As no additional motor parameters are introduced in the calculation process of the reference voltage space vector, the robustness of the motor parameters is effectively enhanced.
Additional features and advantages of the invention will be set forth in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. The objectives and other advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.
Drawings
The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention and not to limit the invention.
Fig. 1 is a schematic structural diagram of a double three-phase permanent magnet synchronous motor according to an embodiment of the invention;
fig. 2 is an equivalent circuit diagram of a dual three-phase permanent magnet synchronous motor driving system according to an embodiment of the present invention;
FIG. 3 is a schematic diagram of the direct torque control system of the motor according to the first embodiment of the present invention;
FIG. 4 is a flow chart of an operation for implementing a corresponding motor direct torque control method using the motor direct torque control system of FIG. 3;
FIG. 5 is a schematic diagram of a harmonic current controller in a direct torque control system for an electric machine according to a first embodiment of the present invention;
FIG. 6 is a schematic diagram of a harmonic current controller in a direct torque control system for an electric machine according to a second embodiment of the present invention;
fig. 7 is a design diagram of a harmonic current controller in a direct torque control system of a motor according to a third embodiment of the present invention.
Detailed Description
The following detailed description of the embodiments of the present invention will be provided with reference to the drawings and examples, so that how to apply the technical means to solve the technical problems and achieve the technical effects can be fully understood and implemented. It should be noted that, as long as there is no conflict, the embodiments and the features of the embodiments of the present invention may be combined with each other, and the technical solutions formed are within the scope of the present invention.
In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of the embodiments of the invention. It will be apparent, however, to one skilled in the art that the present invention may be practiced without some of these specific details or with other methods described herein.
It should be noted that the method provided by the present invention is not only applicable to a dual three-phase permanent magnet synchronous motor system, but also applicable to a dual three-phase asynchronous motor system (i.e. a dual three-phase induction motor system). The method provided by the invention is not only suitable for a double three-phase motor system (a permanent magnet synchronous motor or an asynchronous motor) but also suitable for a five-phase motor system (a permanent magnet synchronous motor or an asynchronous motor). The method provided by the invention is not only suitable for the electric running state of a double three-phase or five-phase motor system, but also suitable for the power generation running state of the double three-phase or five-phase motor system. The motor torque control method of the present invention will be described below by taking only a double three-phase permanent magnet synchronous motor as an example.
Fig. 1 is a schematic structural diagram of a dual three-phase permanent magnet synchronous motor in the prior art. It can be seen from the distribution of the stator windings that the stator windings are composed of two sets of conventional three-phase windings ABC and XYZ, each set of windings is connected in a Y shape, the corresponding internal windings are different from each other by 120 degrees in space, and the included angle between the corresponding phases of the two sets of three-phase windings is 30 degrees. Therefore, in terms of the design of a hardware circuit, the double three-phase permanent magnet synchronous motor is a six-phase system, in order to enable the stator flux linkage and the permanent magnet flux linkage to interact to generate constant electromagnetic torque, the phase difference of current of phase windings in each set of Y-shaped windings is 120 degrees, and the phase difference of corresponding phase current between the Y-shaped windings is 30 degrees.
As shown in fig. 2, in the driving system of the dual three-phase permanent magnet synchronous motor, the central points n and n' of the two sets of Y-shaped windings are independent from each other, the dual three-phase permanent magnet synchronous motor is powered by a voltage source inverter, the driving main circuit is formed by connecting two sets of three-phase system driving circuits in parallel with a common direct current bus, wherein 6 mutually independent currents flow in 6-phase stator windings.
Example one
In the present embodiment, the control method of the present invention is implemented by using a direct torque control system of a dual three-phase permanent magnet synchronous motor shown in fig. 3. The system mainly comprises:
a six-dimensional decoupling coordinate transformation module 1 for obtaining six-phase stator voltage signals v of the double three-phase permanent magnet synchronous motorsABCXZYAnd six-phase stator current signal isABCXZYAnd carrying out corresponding coordinate transformation on the stator voltage component v to obtain the stator voltage component v of the alpha-beta sub-planesαβStator current component isαβAnd z1-z2Stator current component i of the harmonic sub-planesz1z2。
Encoder 2 and rotor speed and rotor position acquisition module 3 connected with encoder 2 for acquiring rotor speed omega of double three-phase permanent magnet synchronous motorrAnd rotor position θr. Of course, the rotor speed ω of the dual three-phase permanent magnet synchronous motor can also be obtained by using the position sensorless technologyrAnd rotor position θr. And are not limited herein.
A flux linkage and torque observer module 4 for calculating stator flux linkage psi according to the stator voltage component and the stator current component of the alpha-beta sub-plane obtained by the six-dimensional decoupling coordinate transformation module 1sαβStator flux linkage position thetasAnd electromagnetic torque Te。
A proportional integral module 5 of rotation speed (abbreviated PI module 5 in the figure) for a given rotation speed ω of the rotorr *The rotor speed omega obtained by the rotor speed and rotor position obtaining module 3rThe difference is subjected to proportional integral calculation to obtain a given value T of the electromagnetic torquee *。
A stator flux linkage amplitude setting module 6 for integrating the electromagnetic torque given value T provided by the control module 5 according to the rotation speed proportione *Determining the sum ofe *Given value | Ψ of corresponding stator flux linkage magnitudes1 *And on the other hand, the rotor rotating speed omega obtained by the module 3 is obtained according to the direct current bus voltage and the rotor rotating speed of the inverter and the rotor positionrEstimating the rotor speed omegarGiven value | ψ of corresponding stator flux linkage amplitudes2 *Selecting the smaller value of the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque and the given value of the stator flux linkage amplitude corresponding to the rotor speed as the reference value | psi of the stator flux linkage amplitudes *I.e. | ψs *|＝min(|ψs1 *|，|ψs2 *|)。
A torque proportional-integral module 7 (abbreviated as PI module 7 in the figure) for proportionally integrating the given value T of the electromagnetic torque provided by the control module 5 according to the rotating speede *Calculated value T of the electromagnetic torque estimated by the flux linkage and torque observer module 4eThe difference is subjected to proportional integral calculation to obtain the angle increment delta theta of the stator flux linkages *。
A reference stator flux linkage calculation module 8 for calculating the angular increment Delta theta of the stator flux linkage according to the torque proportional integral module 7s *The reference value | ψ of the stator flux amplitude provided by the stator flux amplitude setting module 6s *The calculated value theta of the stator flux position obtained by the | and flux and torque observer module 4sDetermining a reference value Ψ for a stator flux linkagesαβ *。
A reference voltage calculation module 9 for calculating a reference value psi of the stator flux linkage based on the reference value psi of the stator flux linkage supplied from the reference stator flux linkage calculation module 8sαβ *And the calculated value psi of the stator flux linkage obtained by the flux linkage and torque observer module 4sαβAnd a stator current component i of an alpha-beta sub-plane obtained by the six-dimensional decoupling coordinate transformation module 1sαβDetermining a given value v of a voltage space vector of an alpha-beta sub-planesαβ *。
A PARK conversion module 10 for obtaining the rotor position theta provided by the module 3 based on the rotor speed and the rotor positionrZ provided for the six-dimensional decoupled coordinate transformation module 11-z2Stator current component i of the harmonic sub-planesz1z2Performing synchronous rotation coordinate transformation to obtain z1-z2Stator under harmonic sub-plane synchronous rotation coordinate systemComponent of currentiszdzq。
Harmonic current controller module 11 for controlling the harmonic current at z1-z2Reference value i of stator current component under harmonic sub-plane synchronous rotation coordinate systemszdzq *Is zero (i.e. i)szd *＝iszq *0) according to z provided by PARK transformation module 101-z2Stator current component i under harmonic sub-plane synchronous rotation coordinate systemszdzqOutput corresponding z1-z2Given value v of voltage space vector under sub-plane synchronous rotation coordinate systemszdzq *。
An inverse PARK conversion module 12 for obtaining the rotor position theta provided by the module 3 based on the rotor speed and the rotor positionrZ to harmonic current controller module 11 output1-z2Given value v of voltage space vector under sub-plane synchronous rotation coordinate systemszdz *Inverse transformation of synchronous rotation coordinates is performed to output z1-z2Given value v of voltage space vector of harmonic sub-planesz1z2 *。
An SVPWM module 13 for applying a space vector pulse width modulation technique to a given value v of a voltage space vector of an alpha-beta sub-plane output by the reference voltage calculation module 9sαβ *And z output from inverse PARK transform module 121-z2Given value v of voltage space vector of harmonic sub-planesz1z2 *And modulating to generate a switching signal for controlling a switching tube of the inverter, so that the torque of the double three-phase permanent magnet synchronous motor is controlled.
Fig. 4 is a flowchart illustrating the operation of the direct torque control method of the motor according to the present embodiment. The method mainly comprises the following steps:
step S10: six-phase stator voltage signal u of double three-phase permanent magnet synchronous motor is obtained through current sensor and voltage sensora，ub，uc，ux，uy，uz(for ease of labeling, v is abbreviated in the system shown in FIG. 3sABCXZY) And six-phase stator current signal ia，ib，ic，ix，iy，iz(for ease of labeling, abbreviated i in the system shown in FIG. 3sABCXZY) Obtaining stator voltage component u of alpha-beta sub-plane by respectively carrying out coordinate transformation on six-phase stator voltage signal and six-phase stator current signalsα，usβ(for ease of labeling, v is abbreviated in the system shown in FIG. 3sαβ) Stator current component isα，isβ(for ease of labeling, abbreviated i in the system shown in FIG. 3sαβ) And z is1-z2Stator current component i of the harmonic sub-planesz1，isz2(for ease of labeling, abbreviated i in the system shown in FIG. 3sz1z2)。
Further, the coordinate transformation in this step is a six-dimensional decoupled coordinate transformation.
The current transformation is taken as an example, six-dimensional decoupling coordinate transformation is adopted to carry out coordinate transformation on six-phase stator current, and the specific transformation formula is as follows:
Figure BDA0001989438480000111
wherein isα，isβStator current components i on the alpha and beta axes, respectively, of the stator current in the alpha-beta sub-planesz1，isz2Respectively stator current in z1-z2In the harmonic sub-plane z1Axis and z2Stator current component on shaft, iso1，iso2Respectively stator current at o1-o2In the sub-plane o1Shaft and o2Stator current components on the shaft.
Taking voltage transformation as an example, performing coordinate transformation on six-phase stator voltage by adopting six-dimensional decoupling coordinate transformation, wherein the specific transformation formula is as follows:
Figure BDA0001989438480000112
wherein u issa，usβAre respectively asStator voltage components, u, of the stator voltage on the alpha and beta axes in the alpha-beta sub-planesz1，usz2Respectively stator voltage at z1-z2In the harmonic sub-plane z1Axis and z2Stator voltage component on the shaft, uso1，uso2Respectively stator voltage at o1-o2In the sub-plane o1Shaft and o2Stator voltage component on the shaft.
Meanwhile, the step also comprises the step of obtaining the rotor rotating speed omega of the double three-phase permanent magnet synchronous motorrAnd rotor position θr。
Step S20: the stator voltage component u of the alpha-beta sub-plane obtained according to step S10sα，usβAnd a stator current component isα，isβCalculating the stator flux linkage psi bysαβStator flux linkage position thetasAnd electromagnetic torque Te。
Figure BDA0001989438480000113
Figure BDA0001989438480000121
Te＝3P(ψsαisβ-ψsβisα)
Wherein u issα，usβComponents of the stator voltage in the alpha-beta sub-plane on the alpha and beta axes, isα，isβComponents of the stator current in the alpha-beta sub-plane in the alpha-beta axis and beta axis, psisα，ψsβThe components of the stator flux linkage in the α - β sub-plane, the α -axis and the β -axis, respectively, RsIs the stator resistance of the motor, P is the pole pair number of the motor, and stator flux linkage psisαβIs psisα，ψsβBoth in shorthand form.
Step S30: by setting a given rotor speed omegar *And the rotor rotation speed ω obtained in step S10rThe difference is subjected to proportional integral calculationDetermining a setpoint value T of the electromagnetic torquee *And further determining a set value T of the electromagnetic torquee *Given value | psi of corresponding stator flux linkage amplitudes1 *L, |; according to the DC bus voltage of the inverter and the rotor speed omega obtained in the step S10rEstimating the rotor speed omegarGiven value | psi of corresponding stator flux linkage amplitudes2 *L. Taking a given value | ψ of a stator flux linkage amplitude corresponding to a given value of electromagnetic torques1 *| and given value | ψ of stator flux linkage amplitude corresponding to rotor rotation speeds2 *The smaller value of | is taken as the reference value | ψ of the stator flux linkage amplitudes *|。
On the one hand, the given value T is based on the electromagnetic torquee *The given value T of the electromagnetic torque can be determined by searching a flux linkage-torque table obtained according to a maximum torque current ratio algorithme *Given value | psi of corresponding stator flux linkage amplitudes1 *|。
On the other hand, according to the DC bus voltage V of the inverterdcAnd the rotor rotation speed ω acquired in step S10rThe rotor speed ω can be determined by using the following equationrCalculated value | ψ of corresponding stator flux linkage amplitudes2 *|：
|ψs2 *|＝Umax/ωr，
Figure BDA0001989438480000122
Wherein, UmaxThe maximum amplitude of the phase voltage allowed to be provided by the inverter, η, is a positive coefficient smaller than 1 and close to 1.η may take the value of 0.95.
Obtaining a given value | ψ of a stator flux linkage amplitude corresponding to a given value of electromagnetic torques1 *| and given value | ψ of stator flux linkage amplitude corresponding to rotor rotation speeds2 *After the stator flux linkage amplitude is I, the smaller value of the two is selected as a reference value | psi of the stator flux linkage amplitudes *I.e. | ψs *|＝min(|ψs1 *|，|ψs2 *|)。
Step S40: by setting the electromagnetic torque to a given value Te *The calculated value T of the electromagnetic torque obtained in step S20eThe difference is subjected to proportional integral calculation to determine the angle increment delta theta of the stator flux linkages *So as to be able to increase by delta theta in accordance with the angle of the stator flux linkages *The reference value | ψ of the stator flux linkage amplitude determined in step S30s *| and the calculated value θ of the stator flux linkage position obtained in step S20sDetermining a reference value psi of the stator flux linkagesαβ *And further according to the reference value psi of the stator flux linkagesαβ *And the calculated value ψ of the stator flux linkage obtained in step S20sαβAnd the stator current component i of the alpha-beta sub-plane obtained in step S20sα，isβDetermining a given value of a voltage space vector of an alpha-beta sub-plane
Figure BDA0001989438480000131
Specifically, the angle increment Δ θ according to the stator flux linkages *The reference value | ψ of the stator flux linkage amplitude determined in step S30s *| and the calculated value θ of the stator flux linkage position obtained in step S20sReference value ψ of stator flux linkage determined by the following equationsαβ *。
Figure BDA0001989438480000132
Wherein psisα *Is the component of the reference value of the stator flux linkage in the alpha axis of the alpha-beta sub-plane, psisβ *Is the component of the reference value of the stator flux linkage in the beta axis of the alpha-beta sub-plane, psisαβ *As vectors
Figure BDA0001989438480000133
In which j is the imaginary symbol.
Then, according to the reference value psi of the stator flux linkagesαβ *And the calculated value ψ of the stator flux linkage obtained in step S20sαβAnd the stator current component i of the alpha-beta sub-plane obtained in step S20sα，isβDetermining the given value v of the voltage space vector of the alpha-beta sub-plane using the formulasαβ *。
Figure BDA0001989438480000134
Wherein the content of the first and second substances,
Figure BDA0001989438480000135
is the component of a given value of the voltage space vector on the alpha axis of the alpha-beta sub-plane,
Figure BDA0001989438480000136
is a component of a given value of the voltage space vector on the beta axis of the alpha-beta sub-plane, RsIs stator resistance, TsIs the sampling period of the system.
Figure BDA0001989438480000137
As vectors
Figure BDA0001989438480000138
In which j is the imaginary symbol.
Step S50: z obtained based on step S101-z2Stator current component of harmonic sub-plane, obtaining z by harmonic current controller1-z2A given value of the voltage space vector of the harmonic sub-plane.
Specifically, the steps are divided into the following small steps:
step S51, based on the rotor position θ obtained in step S20rZ obtained in step S10 is calculated by the following equation1-z2Stator current component i of the harmonic sub-planesz1，isz2Performing synchronous rotation coordinate transformation to obtain z1-z2Harmonic sub-plane synchronous rotation coordinate systemStator current component iszd，iszq。
Figure BDA0001989438480000141
Wherein iszd，iszqRespectively stator current in z1-z2Component in a harmonic sub-plane synchronous rotating coordinate system, isz1，isz2Respectively stator current in z1-z2Component of harmonic sub-plane, θrIs the rotor position.
Step S52, based on z1-z2Condition that the reference value of the stator current component is zero in the harmonic sub-plane synchronous rotation coordinate system, i.e.
Figure BDA0001989438480000142
Z obtained according to step S511-z2Stator current component i under harmonic sub-plane synchronous rotation coordinate systemszdAnd iszqObtaining z by means of a harmonic current controller1-z2Given value of voltage space vector under sub-plane synchronous rotation coordinate system
Figure BDA0001989438480000143
Figure BDA0001989438480000144
As vectors
Figure BDA0001989438480000145
In which j is the imaginary symbol.
As shown in fig. 5, the harmonic current controller of the present embodiment is formed by combining a proportional-integral resonance regulator and a six-fold resonance regulator, and the resonance regulator of the proportional-integral resonance regulator is a double-fold resonance regulator.
The transfer function of a proportional-integral resonant regulator is as follows:
Figure BDA0001989438480000146
wherein k isp，ki，kr2The proportional coefficient, the integral coefficient and the double frequency resonance coefficient of the proportional integral resonance regulator are respectively, and omega is frequency.
The transfer function of the six-fold frequency resonance adjuster is as follows:
Figure BDA0001989438480000147
wherein k isr6Is a 6 times frequency resonance regulator coefficient.
Specifically, z obtained in step S511-z2Stator current component i under harmonic sub-plane synchronous rotation coordinate systemszdAnd iszqAnd
Figure BDA0001989438480000148
inputting the harmonic current controller to obtain z1-z2Given value of voltage space vector under sub-plane synchronous rotation coordinate system
Figure BDA0001989438480000149
Figure BDA00019894384800001410
As vectors
Figure BDA00019894384800001411
In which j is the imaginary symbol.
S53, for z obtained in step S521-z2Given value of voltage space vector under sub-plane synchronous rotation coordinate system
Figure BDA00019894384800001412
Carrying out inverse transformation on the synchronous rotation coordinate to obtain z1-z2Given value of voltage space vector of harmonic sub-plane
Figure BDA00019894384800001413
Figure BDA00019894384800001414
As vectors
Figure BDA00019894384800001415
In which j is the imaginary symbol.
Specifically, z is represented by the following formula1-z2Voltage space vector under harmonic sub-plane synchronous rotation coordinate system
Figure BDA0001989438480000151
The given value of the z is subjected to synchronous rotation coordinate inverse transformation to obtain z1-z2Given value of voltage space vector of harmonic sub-plane
Figure BDA0001989438480000152
Figure BDA0001989438480000153
Wherein v isszd *，vszq *Are each z1-z2Component of a given value of a voltage space vector, v, in a harmonic sub-plane synchronous rotation coordinate systemsz1 *，vsz2 *Are each z1-z2The component of a given value of the voltage space vector of the harmonic sub-plane.
Step S60: applying Space Vector Pulse Width Modulation (SVPWM) technique to the given value of the voltage space vector of the alpha-beta sub-plane obtained in step S40
Figure BDA0001989438480000154
And z obtained in step S501-z2Given value of voltage space vector of harmonic sub-plane
Figure BDA0001989438480000155
Modulating, generating for controllingAnd switching signals of a switching tube of the inverter are controlled, so that the control of the torque of the double three-phase permanent magnet synchronous motor is realized.
Example two
In this embodiment, a processing manner different from that of the first embodiment is adopted. Specifically, a harmonic current controller including a dual PI controller is adopted in the dual synchronous rotating coordinate system instead of the harmonic current controller in the first embodiment. As shown in fig. 6, the harmonic current controller includes a proportional-integral regulator and a six-fold resonant regulator.
The transfer function of the proportional-integral regulator is as follows:
Figure BDA0001989438480000156
wherein k isp，kiThe proportional coefficient and the integral coefficient of the proportional-integral regulator are respectively.
The transfer function of the six-fold frequency resonance adjuster is as follows:
Figure BDA0001989438480000157
wherein k isr6Is a 6 times frequency resonance regulator coefficient.
Accordingly, steps S10 to S40 and S60 of the control method of the present embodiment are the same as those of the first embodiment, except for step S50. In this embodiment, a double-synchronous rotating coordinate transformation different from the first embodiment is used to transform the harmonic current signal to a forward rotating coordinate system and a reverse rotating coordinate system, and then two sets of current errors are controlled by two PI regulators respectively to realize the regulation of the fundamental frequency positive sequence component and the fundamental frequency negative sequence component, and simultaneously, a frequency doubling resonance regulator 6 times under the forward rotating coordinate system is used to regulate the 5 th harmonic and the 7 th harmonic brought by the dead zone of the inverter (as shown in fig. 6).
Specifically, step S50 is divided into the following small steps:
s51, based on the rotor position obtained in step S20,for z obtained in step S101-z2The stator current component of the harmonic sub-plane is subjected to forward rotation coordinate transformation and reverse rotation coordinate transformation to obtain z1-z2Stator current components under a harmonic sub-plane forward rotation coordinate system and stator current components under a harmonic sub-plane reverse rotation coordinate system;
s52, based on z1-z2The condition that the reference value of the stator current component in the harmonic sub-plane forward rotation coordinate system is zero is based on z obtained in step S511-z2The stator current component under the harmonic sub-plane forward rotation coordinate system is obtained by a proportional-integral regulator and a six-time frequency resonance regulator in the harmonic current controller1-z2A given value of a voltage space vector under a harmonic sub-plane forward rotation coordinate system;
s53, based on z1-z2On condition that the reference value of the stator current component in the harmonic sub-plane reverse rotation coordinate system is zero, z is obtained from step S511-z2The stator current component in harmonic sub-plane reverse rotation coordinate system is obtained by a proportional-integral regulator in the harmonic current controller1-z2A given value of a voltage space vector under a harmonic sub-plane reverse rotation coordinate system;
s54, for z obtained in step S521-z2Performing reverse transformation of forward rotation coordinate on the given value of the voltage space vector in the sub-plane forward rotation coordinate system and performing reverse transformation on the z obtained in the step S531-z2Carrying out reverse rotation coordinate inverse transformation on the given value of the voltage space vector under the sub-plane reverse rotation coordinate system, and adding the inverse transformation results of the given value and the voltage space vector to obtain z1-z2A given value of the voltage space vector of the harmonic sub-plane.
EXAMPLE III
In this embodiment, a processing manner different from those of the first and second embodiments is adopted. In particular, another harmonic current controller is employed in a stationary coordinate system (i.e., without coordinate transformation).
Specifically, the harmonic current controller is composed of a proportional and fundamental resonance adjuster, a quintuple resonance adjuster, and a heptatuple resonance adjuster (as shown in fig. 7).
The transfer functions of the proportional and fundamental resonance adjusters are as follows:
Figure BDA0001989438480000161
the transfer function of a quintupled resonant regulator is as follows:
Figure BDA0001989438480000162
the transfer function of the seven-fold frequency resonance adjuster is as follows:
Figure BDA0001989438480000171
wherein k ispIs the proportionality coefficient, k, of a proportional-integral regulatorrIs the fundamental resonance regulator coefficient, kr5Is a 5 frequency multiplication resonance regulator coefficient, kr7Is a factor of 7 frequency doubling resonance regulator.
Similarly, steps S10 to S40 and S60 of the control method of the present embodiment are the same as those of the first embodiment and the second embodiment, except for step S50. In this embodiment, no coordinate transformation is required, i.e. no rotor position is required, but three resonant adjusters of different frequencies are required: the proportion and fundamental frequency resonance regulator is used for regulating positive sequence components and negative sequence components of fundamental frequency, the 5-time frequency resonance regulator is used for regulating 5-time harmonic waves brought by an inverter dead zone, and the 7-time frequency resonance regulator is used for regulating 7-time harmonic waves brought by the inverter dead zone.
Specifically, the step S50 is:
based on z1-z2The condition that the reference value of the harmonic sub-plane stator current component is zero is based on z obtained in step S101-z2Stator current component of harmonic sub-plane, obtaining z by harmonic current controller1-z2Given values of the voltage space vector of the sub-plane.
It should be noted that, although the embodiments of the present invention are described above, the descriptions are only for the convenience of understanding the present invention and are not intended to limit the present invention. For example, the method provided by the invention is not only applicable to a double three-phase permanent magnet synchronous motor system, but also applicable to a double three-phase asynchronous motor system (namely a double three-phase induction motor system). The method provided by the invention is not only suitable for a double three-phase motor system (a permanent magnet synchronous motor or an asynchronous motor) but also suitable for a five-phase motor system (a permanent magnet synchronous motor or an asynchronous motor). The method provided by the invention is not only suitable for the electric running state of a double three-phase or five-phase motor system, but also suitable for the power generation running state of the double three-phase or five-phase motor system. Thus, it will be apparent to persons skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention as disclosed, and the scope of the invention as defined in the appended claims is to be accorded the full scope of the invention.

Claims (5)

1. A method of direct torque control of an electric machine, comprising the steps of:
s10: six-phase stator voltage signals and six-phase stator current signals of the double three-phase motor are obtained, and stator voltage components, stator current components and z of alpha-beta sub-planes are obtained by performing coordinate transformation on the obtained signals1-z2A stator current component of the harmonic sub-plane; acquiring the rotor speed and the rotor position of the motor;
s20: estimating a stator flux linkage, a stator flux linkage position, and an electromagnetic torque from the stator voltage component and the stator current component of the α - β sub-plane obtained in step S10; wherein the stator flux linkage ψ is calculated from the stator voltage component and the stator current component of the α - β sub-plane obtained in step S10 by the following equationsαβStator flux linkage position thetasAnd electromagnetic torque Te：
Figure FDA0003321837000000011
Figure FDA0003321837000000012
Te＝3P(ψsαisβ-ψsβisα)
Wherein u issα,usβComponents of the stator voltage in the alpha-beta sub-plane on the alpha and beta axes, isα,isβComponents of the stator current in the alpha-beta sub-plane in the alpha-beta axis and beta axis, psisα,ψsβThe components of the stator flux linkage in the α - β sub-plane, the α -axis and the β -axis, respectively, RsIs the stator resistance of the motor, P is the pole pair number of the motor, psisαβA stator flux linkage;
s30: determining a given value of the electromagnetic torque by performing proportional integral calculation on the difference between the given rotor speed and the rotor speed obtained in step S10, and further determining a given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque; estimating a given value of a stator flux linkage amplitude corresponding to the rotor speed according to the direct-current bus voltage of the inverter and the rotor speed obtained in the step S10; taking the smaller value of the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque and the given value of the stator flux linkage amplitude corresponding to the rotor rotation speed as the reference value of the stator flux linkage amplitude; wherein, the given value of the stator flux linkage amplitude corresponding to the rotor speed is estimated according to the direct-current bus voltage of the inverter and the rotor speed obtained in step S10 by the following formula:
|ψs2 *|＝Umax/ωr
Figure FDA0003321837000000013
wherein, | ψs2 *I is stator flux linkage amplitudeCalculated value of (a), ωrAs the rotational speed of the rotor, VdcIs the DC bus voltage of the inverter, UmaxThe maximum amplitude value of the phase voltage allowed to be provided for the inverter is provided, and eta is a positive coefficient;
s40: determining an angle increment of the stator flux linkage by performing a proportional integral calculation on a difference between a given value of the electromagnetic torque and the calculated value of the electromagnetic torque obtained in step S20, determining a reference value of the stator flux linkage according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined in step S30, and the calculated value of the position of the stator flux linkage obtained in step S20, and determining a given value of a voltage space vector of the α - β sub-plane according to the reference value of the stator flux linkage, the calculated value of the stator flux linkage obtained in step S20, and a stator current component of the α - β sub-plane obtained in step S10; wherein the reference value of the stator flux linkage is determined according to the angle increment of the stator flux linkage, the reference value of the amplitude of the stator flux linkage determined at step S30, and the calculated value of the position of the stator flux linkage obtained at step S20 by the following formula:
Figure FDA0003321837000000021
wherein the content of the first and second substances,
Figure FDA0003321837000000022
the components of the reference value of the stator flux linkage in the alpha and beta axes in the alpha-beta sub-plane respectively,
Figure FDA0003321837000000023
in increments of the angle of the stator flux linkage,
Figure FDA0003321837000000024
is a reference value of stator flux linkage amplitude, thetasIs the stator flux linkage position;
determining a given value of a voltage space vector of the α - β sub-plane from the reference value of the stator flux linkage and the calculated value of the stator flux linkage obtained at step S20 and the stator current component of the α - β sub-plane obtained at step S10 by:
Figure FDA0003321837000000025
wherein the content of the first and second substances,
Figure FDA0003321837000000026
components of a given value of the voltage space vector of the alpha-beta sub-plane on the alpha and beta axes, i, respectively, in the alpha-beta sub-planesα,isβThe components of the stator current in the alpha-beta sub-plane on the alpha and beta axes, RsIs the stator resistance, T, of the motorsIs the sampling period of the system;
s50: z obtained based on step S101-z2Stator current component of harmonic sub-plane, obtaining z by harmonic current controller1-z2A given value of the voltage space vector of the harmonic sub-plane; wherein z is represented by the formula1-z2Stator current component i of the harmonic sub-planesz1,isz2Performing synchronous rotation coordinate transformation to obtain z1-z2Stator current component i under harmonic sub-plane synchronous rotation coordinate systemszd,iszq：
Figure FDA0003321837000000027
By the formula to z1-z2The given value of the voltage space vector under the harmonic sub-plane synchronous rotation coordinate system is subjected to synchronous rotation coordinate inverse transformation to obtain z1-z2Given value of the voltage space vector of the harmonic sub-plane:
Figure FDA0003321837000000031
wherein v isszd *,vszq *Are each z1-z2Voltage space under harmonic sub-plane synchronous rotation coordinate systemComponent of a given value of an intervector, vsz1 *,vsz2 *Are each z1-z2Component of a given value of the voltage space vector of the harmonic sub-plane, thetarIs the rotor position;
s60: using space vector pulse width modulation method to obtain given value of voltage space vector of alpha-beta sub-plane obtained in step S40 and z obtained in step S501-z2The given value of the voltage space vector of the harmonic sub-plane is modulated to generate a switching signal for controlling a switching tube of the inverter, so that the direct torque control of the motor is realized;
the harmonic current controller is configured to reduce harmonic voltages caused by inherent asymmetry of the motor and the inverter and harmonic voltages caused by nonlinear linear factors of the inverter, and further suppress harmonic currents generated by the harmonic voltages.
2. The motor direct torque control method according to claim 1, wherein the step S50 further includes the steps of:
s51, based on the rotor position obtained in step S10, for z obtained in step S101-z2Stator current components of the harmonic sub-planes are subjected to synchronous rotation coordinate transformation to obtain z1-z2Stator current components under a harmonic sub-plane synchronous rotation coordinate system;
s52, based on z1-z2The condition that the reference value of the stator current component is zero in the harmonic sub-plane synchronous rotation coordinate system is based on z obtained in step S511-z2Obtaining the stator current component in the harmonic sub-plane synchronous rotation coordinate system by using a harmonic current controller1-z2Given values of voltage space vectors under a sub-plane synchronous rotation coordinate system;
s53, for z obtained in step S521-z2Carrying out synchronous rotation coordinate inverse transformation on a given value of a voltage space vector under a sub-plane synchronous rotation coordinate system to obtain z1-z2A given value of the voltage space vector of the harmonic sub-plane;
the harmonic current controller comprises a proportional integral resonance regulator and a six-frequency-doubling resonance regulator, and the resonance regulator of the proportional integral resonance regulator is a frequency-doubling resonance regulator.
3. The motor direct torque control method according to claim 1, wherein the step S50 further includes the steps of:
s51, based on the rotor position obtained in step S10, for z obtained in step S101-z2The stator current component of the harmonic sub-plane is subjected to forward rotation coordinate transformation and reverse rotation coordinate transformation to obtain z1-z2Stator current components under a harmonic sub-plane forward rotation coordinate system and stator current components under a harmonic sub-plane reverse rotation coordinate system;
s52, based on z1-z2The condition that the reference value of the stator current component in the harmonic sub-plane forward rotation coordinate system is zero is based on z obtained in step S511-z2The stator current component under the harmonic sub-plane forward rotation coordinate system is obtained by a proportional-integral regulator and a six-time frequency resonance regulator in the harmonic current controller1-z2A given value of a voltage space vector under a harmonic sub-plane forward rotation coordinate system;
s53, based on z1-z2On condition that the reference value of the stator current component in the harmonic sub-plane reverse rotation coordinate system is zero, z is obtained from step S511-z2Obtaining a stator current component in a harmonic sub-plane reverse rotation coordinate system by using another proportional-integral regulator in the harmonic current controller1-z2A given value of a voltage space vector under a harmonic sub-plane reverse rotation coordinate system;
s54, for z obtained in step S521-z2Performing reverse transformation of forward rotation coordinate on the given value of the voltage space vector in the sub-plane forward rotation coordinate system and performing reverse transformation on the z obtained in the step S531-z2The given value of the voltage space vector under the sub-plane reverse rotation coordinate system is subjected to reverse rotation coordinate inverse transformation, and the given value is subjected to reverse rotation coordinate inverse transformationThe inverse transformation results of the two are added to obtain z1-z2A given value of the voltage space vector of the harmonic sub-plane.
4. The direct torque control method of an electric motor according to claim 1, wherein the step S50 is specifically:
based on z1-z2The condition that the reference value of the harmonic sub-plane stator current component is zero is based on z obtained in step S101-z2Stator current component of harmonic sub-plane, obtaining z by harmonic current controller1-z2A given value of the voltage space vector of the sub-plane;
wherein the harmonic current controller comprises a proportional and fundamental resonance adjuster, a quintuple frequency resonance adjuster and a heptatuple frequency resonance adjuster.
5. The motor direct torque control method according to any one of claims 1 to 4, characterized in that:
in step S30, after the given value of the electromagnetic torque is obtained, the given value of the stator flux linkage amplitude corresponding to the given value of the electromagnetic torque is determined by looking up the flux linkage-torque table obtained according to the maximum torque current ratio algorithm.
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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103441726A (en) * 2013-08-25 2013-12-11 浙江大学 Double three-phase permanent magnet motor vector control method based on proportional resonance regulator
CN107222146A (en) * 2017-07-31 2017-09-29 福州大学 The Direct Torque Control of double three-phase permanent-magnetic synchronous motor high load capability
CN108683377A (en) * 2018-04-17 2018-10-19 西安理工大学 Five leg inverters drive double three-phase machine system Strategy of Direct Torque Control

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103441726A (en) * 2013-08-25 2013-12-11 浙江大学 Double three-phase permanent magnet motor vector control method based on proportional resonance regulator
CN107222146A (en) * 2017-07-31 2017-09-29 福州大学 The Direct Torque Control of double three-phase permanent-magnetic synchronous motor high load capability
CN108683377A (en) * 2018-04-17 2018-10-19 西安理工大学 Five leg inverters drive double three-phase machine system Strategy of Direct Torque Control

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
双三相永磁同步电机直接转矩控制;张华强;《电气传动》;20170217;第3-8页 *

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