CN111416635A

CN111416635A - Angle measuring method and angle measuring equipment

Info

Publication number: CN111416635A
Application number: CN202010198392.2A
Authority: CN
Inventors: 张文彦; 张玉龙; 李铮
Original assignee: Sensethink Technology Shenzhen Co ltd; Shanghai Sensethink Communications R&d Co ltd
Current assignee: Sensethink Technology Shenzhen Co ltd; Shanghai Sensethink Communications R&d Co ltd
Priority date: 2020-03-19
Filing date: 2020-03-19
Publication date: 2020-07-14
Anticipated expiration: 2040-03-19
Also published as: CN111416635B

Abstract

The application discloses an angle measurement method and angle measurement equipment. The angle measuring method comprises the following steps: receiving radio frequency signals by using M antennas, wherein the M antennas comprise H angle measuring antenna pairs; selecting radio frequency signals received by one antenna in turn according to a preset mode to carry out radio frequency processing, detecting whether a lead code exists in baseband signals in a current time window, responding to the fact that the lead code is not detected, sliding the time window forwards, and repeating the steps until the lead code is detected; estimating a signal phase difference for each goniometric antenna pair in response to detecting the preamble; and determining azimuth angle estimated values of the transmitting antennas based on the signal phase difference estimation results of the H angle measurement antenna pairs. The method has the advantage of low complexity.

Description

Angle measuring method and angle measuring equipment

Technical Field

The present application relates to the field of wireless communication technologies, and in particular, to an angle measurement method and an angle measurement device.

Background

The CSS (Chirp Spread Spectrum, CSS) is a Spread Spectrum technology, and the CSS is mainly applied to radar, and at present, in technologies such as L oRa (L ong Range Radio) and some other low Power Wide Area networks (L ow Power Wide-Area networks, L PWAN), the CSS technology is also used for data transmission.

In wireless communication systems, providing services based on location is becoming an increasing demand. Azimuth based positioning systems, or combined range and azimuth positioning systems, typically require a device to be able to measure the angle of incidence of the signal. Receivers of azimuth-based positioning systems are generally equipped with multiple antennas, and the angle of incidence of signals is measured by processing signals received by the multiple antennas. In a conventional multi-antenna receiver, a radio frequency signal received by each antenna is processed by a respective radio frequency module to obtain a plurality of paths of baseband signals, and the baseband modules process the plurality of paths of baseband signals, so that the implementation complexity of the whole receiver is increased exponentially along with the increase of the number of the antennas.

Disclosure of Invention

The method comprises the steps of selecting a radio-frequency signal received by one of a plurality of antennas in turn to carry out radio-frequency processing to obtain a baseband signal, processing the baseband signal by utilizing the characteristic that a lead code contains repeated chirp spread spectrum modulation symbols, dividing the plurality of antennas into a plurality of angle measurement antenna pairs, estimating the signal phase difference of each angle measurement antenna pair, calculating a candidate azimuth angle set according to the phase difference estimation result, and obtaining an azimuth angle estimation value of a transmitting antenna based on the candidate azimuth angle set. Compared with the traditional angle measurement method, the angle measurement method disclosed by the invention has the beneficial effect of low hardware implementation complexity.

Specifically, the present disclosure provides an angle measurement method, which may be applied to an angle measurement apparatus, the method including: receiving a radio frequency signal transmitted from a transmitting antenna by using M antennas, wherein the radio frequency signal carries a wireless data frame, the wireless data frame comprises a frame header modulated by using chirp spread spectrum, and the frame header comprises N_pThe antenna array comprises a lead code consisting of repeated preset chirp spread spectrum modulation symbols, wherein the phases of two adjacent chirp spread spectrum modulation symbols of the lead code are continuous, the M antennas comprise H angle measurement antenna pairs, each angle measurement antenna pair comprises a first antenna and a second antenna, the distance between the first antenna and the second antenna is not more than lambda/2, H and Np are integers more than or equal to 2, and lambda is the carrier wavelength of the radio-frequency signal; selecting a radio frequency signal received by one of the M antennas in turn according to a preset mode to perform radio frequency processing so as to obtain a baseband signal corresponding to each antenna, detecting whether a lead code exists in the baseband signal in a current time window, and sliding the time window forward for a preset time length in response to the fact that the lead code is not detected; repeating this step until the preamble is detected; in response to detecting the preamble, estimating a signal phase difference of each of the H angle-measuring antenna pairs, wherein the signal phase difference estimation result of the H-th angle-measuring antenna pair is Δ θ_h(ii) a And determining the azimuth angle estimated value of the transmitting antenna based on the signal phase difference estimation results of the H angle measurement antenna pairs.

In some preferred embodiments, the predetermined time length is a multiple of the period of the chirped spread spectrum modulation symbol.

In some embodiments, the M antennas are located in the same plane.

In some embodiments, the step of detecting whether the baseband signal within the current time window includes the preamble comprises: dividing the baseband signal corresponding to each antenna in the current time window into a plurality of receiving symbols according to the chirp spread spectrum modulation symbol period; calculating corresponding matching symbols for each received symbol of each antenna to obtain each antennaThe matching signal of (1); calculating discrete Fourier transform for the matching symbol of each antenna to obtain a corresponding frequency domain matching symbol; calculating corresponding frequency domain matching autocorrelation symbols for adjacent frequency domain matching symbols of each antenna; accumulating the frequency-based points of all the frequency-domain matching autocorrelation symbols of the M antennas to obtain a total frequency-domain matching autocorrelation symbol; determining a frequency domain peak E of the summed frequency domain matched autocorrelation symbols^maxAnd whether a preset threshold value is exceeded, if the preset threshold value is exceeded, the preamble is considered to be detected, otherwise, the preamble is considered not to be detected.

In some embodiments, the step of estimating the signal phase difference of each of the H goniometric antenna pairs comprises: frequency domain matching of the kth of the autocorrelation symbol for each of the M antennas_maxAccumulating the frequency points to obtain a decimal frequency multiplication offset compensation accumulation result of each antenna; and estimating the signal phase difference of each antenna pair of the H angle measurement antenna pairs based on the decimal frequency multiplication deviation compensation accumulation result of each antenna.

In some further embodiments, the step of estimating the signal phase difference of each of the H goniometric antenna pairs comprises: estimating fractional frequency offset_f(ii) a Frequency domain matching of the kth of the autocorrelation symbol for each of the M antennas_maxCarrying out decimal frequency multiplication offset compensation on each frequency point; frequency domain matching of the kth of the autocorrelation symbol for each of the M antennas_maxAccumulating the decimal frequency multiplication deviation compensation results of the frequency points to obtain decimal frequency multiplication deviation compensation accumulation results of each antenna; and estimating the signal phase difference of each angle measuring antenna pair of the H angle measuring antenna pairs based on the decimal frequency multiplication deviation compensation accumulation result of each antenna.

In some embodiments, the step of determining an azimuth estimate for the transmit antenna based on the signal phase difference estimates for the H goniometric antenna pairs comprises: calculating candidate azimuth angle pairs based on each of the H angle-measuring antenna pairs based on the signal phase difference estimation result of the angle-measuring antenna pair, wherein the candidate azimuth angle pairs comprise a first candidate azimuth angle and a second candidate squareAn azimuth angle; forming a candidate azimuth set based on first candidate azimuths and second candidate azimuths of all angle-measuring antenna pairs of the H angle-measuring antenna pairs

From the set of candidate azimuths

Determining a subset of azimuth angles phi; calculating an azimuth estimate for the transmit antenna based on the azimuth subset Φ.

The present disclosure also provides an angle measurement device, which includes M antennas for receiving radio frequency signals, a multiplexer, a radio frequency module, and a baseband module. The radio frequency signal carries a wireless data frame, the wireless data frame comprises a frame header modulated by using chirp spread spectrum, the M antennas comprise H angle measuring antenna pairs, each angle measuring antenna pair comprises a first antenna and a second antenna, the distance between the first antenna and the second antenna is not more than lambda/2, H is an integer larger than or equal to 2, and lambda is the carrier wavelength of the radio frequency signal; the multiplexer is configured to select one of the radio frequency signals received by the M antennas for output according to an antenna selection control signal; the radio frequency module is configured to perform radio frequency processing on the radio frequency signal output by the multiplexer to obtain a corresponding baseband signal; the baseband module is coupled to the multiplexer and configured to process the baseband signal to generate the antenna selection control signal and determine an azimuth estimate for the transmit antenna.

In some embodiments, the M antennas are located in the same plane.

In some embodiments, the processing of the baseband signal by the baseband module comprises: detecting whether a preamble exists in the baseband signal; in response to detecting the preamble, estimating a signal phase difference of each of the H angle-measuring antenna pairs, wherein the signal phase difference estimation result of the H-th angle-measuring antenna pair is Δ θ_h(ii) a Based on each of the H pairs of angle-measuring antennasAnd determining an azimuth angle estimated value of the transmitting antenna according to the signal phase difference estimation result.

In some embodiments, the baseband module detecting whether a preamble is present in the baseband signal comprises: generating the antenna selection control signal for alternately selecting the radio frequency signal received by one of the M antennas to be output according to a preset mode, detecting whether the baseband signal in the current time window comprises the preamble, responding to the fact that the preamble is not detected, sliding the time window forward for a preset time length, and then repeatedly executing the steps until the preamble is detected.

The foregoing is a summary of the application that may be simplified, generalized, and details omitted, and thus it should be understood by those skilled in the art that this section is illustrative only and is not intended to limit the scope of the application in any way. This summary is not intended to identify key features or essential features of the claimed subject matter, nor is it intended to be used as an aid in determining the scope of the claimed subject matter.

Drawings

The above-described and other features of the present disclosure will become more fully apparent from the following description and appended claims, taken in conjunction with the accompanying drawings. It is appreciated that these drawings depict only several embodiments of the disclosure and are therefore not to be considered limiting of its scope. The present disclosure will be described more clearly and in detail by using the accompanying drawings.

Fig. 1 illustrates a structure of a data frame 10 of a wireless communication system of the present disclosure;

FIG. 2 illustrates an angle measurement method 20 of an embodiment of the present disclosure;

FIG. 3 schematically illustrates a diagram of angle measurement using a 3-antenna receiver;

FIG. 4 is a schematic diagram illustrating an exemplary calculation of a candidate azimuth angle based on a pair of angle-measuring antennas;

fig. 5 schematically illustrates a schematic diagram of the present disclosure for alternately selecting rf signals received by M antennas for rf processing;

fig. 6 illustrates an angle measurement device 60 of an embodiment of the present disclosure.

Detailed Description

To make the objects, technical solutions and advantages of the present disclosure more apparent, the present disclosure will be described in detail below with reference to the accompanying drawings. In the drawings, like reference numerals generally refer to like parts throughout the various views unless the context dictates otherwise. The illustrative embodiments described in the detailed description, drawings, and claims are only a few examples of the disclosure, and not all examples. It will be understood that aspects of the present disclosure, as generally described in the present disclosure and illustrated in the figures herein, may be arranged, substituted, combined, and designed in a wide variety of different configurations, all of which form part of the present disclosure. Other embodiments may be utilized, and other changes may be made, without departing from the spirit or scope of the subject matter of this application, and any such modifications, equivalents, improvements, etc., that are made based on this disclosure are intended to be included within the scope of this disclosure.

The present disclosure provides an angle measurement method for measuring an azimuth angle of a transmitted signal using multiple antennas for a wireless communication system using Chirp Spread Spectrum (CSS) modulation, which does not require the use of multiple rf modules, reduces the number of baseband modules and the number of baseband signals to be processed, and reduces the cost of a receiving device.

Fig. 1 shows a structure of a data frame 10 of a wireless communication system of the present disclosure, which includes a frame header 11 and a frame body 12, wherein the frame body 12 is temporally located behind the frame header 11, the frame header 11 is mainly used for synchronization, and the frame body 12 is mainly used for carrying data. The frame header 11 uses CSS modulation, and the frame body 12 may use CSS modulation, or may use other modulation methods, such as Direct Sequence Spread Spectrum (DSSS) modulation. The frame header 11 comprises a preamble 111, and the preamble 111 is composed of N_pA plurality of repeated predetermined chirp spread spectrum modulation symbols, wherein N_pIs greater thanOr an integer equal to 2, preferably N_pIs an integer greater than or equal to 4, N_pThe larger the value of (c), the more data the receiver can use to process, the better the performance. The wireless communication system of the present disclosure includes a plurality of communication devices, each of which communicates with each other using a structure of a data frame 10, each of which can function as a transmitter, and transmits a signal to other wireless communication devices using the data frame 10; or as a receiver, to receive signals transmitted by other wireless communication devices using the data frame 10.

Chirp Spread Spectrum (CSS) modulation is a technique for modulating the frequency of a signal, the frequency of which varies linearly with time from an initial frequency to a final frequency within a predetermined time interval T, and the frequency variation ranges from-B/2 to B/2, where B is the bandwidth of the chirp spread spectrum signal. If the frequency of the chirp spread spectrum signal increases with time, it is called an up-chirp signal; conversely, if the frequency decreases with increasing time, it is called a down-chirp signal. The initial frequency may be a value between-B/2 and B/2, and the chirped spread spectrum signal may be modulated by mapping different symbols in the symbol set to different initial frequency values, this process is referred to as chirped spread spectrum modulation in this disclosure, the modulated chirped spread spectrum signal is referred to as a chirped spread spectrum modulation signal, the chirped spread spectrum modulation signal within a predetermined time interval T is referred to as a chirped spread spectrum modulation symbol, and the predetermined time interval T is referred to as a chirped spread spectrum modulation symbol period.

For an up-chirped signal, the symbol "ζ" may be mapped to a chirped spread spectrum modulation symbol whose frequency variation over time may be represented as:

wherein the content of the first and second substances,

in order to be the initial frequency of the frequency,

representing frequencyThe rate of change of the speed,

for the time of frequency hopping, "ζ" is referred to as the modulated symbol, ζ ═ 0,1,2, …, S-1, and S is referred to as the spreading factor. When ζ is 0, the corresponding chirp spread spectrum modulation symbol is called basic chirp. It will be appreciated that equation (1) above corresponds the symbol ζ to different initial frequencies f_u,ζ,0Frequency f_u,ζ(t) increases linearly with time, over time

When f is present_u,ζ(t) is equal to B/2, when the frequency f_u,ζ(t) jumps to-B/2 and then increases linearly.

Similarly, for a down-chirp signal, the symbol "ζ" may be mapped to a chirp spread spectrum modulation symbol whose frequency variation over time may be expressed as:

wherein the content of the first and second substances,

in order to be the initial frequency of the frequency,

at the time of frequency hopping, ζ is 0,1,2, …, and S-1, S is a spreading factor.

While the above equations (1) and (2) only show the case where frequency hopping occurs once in the time interval T, it can be understood that the frequency hopping times of the chirped spread spectrum modulation symbol in the time interval T are more than once by increasing the rate of frequency change, and at this time, the function of the frequency change with time can be expressed as a piecewise linear function including a plurality of segments.

The phase of the chirp spread spectrum modulation signal can be controlled by a continuous function

Can be represented byThe frequency is integrated to obtain, for example, for an up-chirped signal, the phase of the chirped spread spectrum modulation symbol corresponding to symbol "ζ" can be expressed as

Corresponding chirp spread spectrum modulation signal s_u,ζ(t) may be expressed in complex signal form as

The phases of adjacent chirped spread spectrum modulation symbols are continuous, that is, the phase of the previous chirped spread spectrum modulation symbol at the termination time is equal to the phase of the next chirped spread spectrum modulation symbol at the start time.

Fig. 2 illustrates an angle measurement method 20 of an embodiment of the present disclosure, which may be applied in a receiver using multi-antenna reception of a wireless communication system of the present disclosure. Fig. 3 is a schematic diagram illustrating an angle measurement performed by a receiver using 3 antennas, and fig. 4 is a schematic diagram illustrating an example of calculating a candidate azimuth angle based on an angle measurement antenna pair. The method 20 includes the following steps 210 through 240.

In step 210, radio frequency signals are received using M antennas.

Wherein, M is the number of receiving antennas of the multi-antenna receiver, and M antennas are positioned in the same plane. It will be appreciated by those skilled in the art that considering that an actual implementation may not be an absolute plane, the M antennas referred to herein may be located in the same plane with a certain tolerance, for example, an included angle between planes formed by any 3 antennas is smaller than a certain angle, for example, 30 °, and the antennas may be considered to be in the same plane. In the schematic diagram of fig. 3, the transmitting antenna 320 is located at point a, the number of receiving antennas is 3, and the receiving antennas include 310, 311, and 313. Those skilled in the art will appreciate that the number of antennas M may be any integer greater than or equal to 3.

The M antennas comprise H angle measuring antenna pairs, each angle measuring antenna pair comprises a first antenna and a second antenna, and the space between the first antenna and the second antennaThe distance is not more than lambda/2, wherein H is an integer greater than or equal to 2, and lambda is the carrier wavelength of the radio frequency signal. Different goniometric antenna pairs may share the same antenna, for example, for the case where M is 3 as shown in fig. 3, the 0 th goniometric antenna pair may include

antennas

310, 311, the 1 st goniometric antenna pair may include

antennas

311, 312, and the 2 nd goniometric antenna pair may include

antennas

312, 310. The distance between the antenna 310 and the antenna 311 is d₀The distance between the antenna 311 and the antenna 312 is d₁The distance between the antenna 312 and the antenna 310 is d₂。

It is contemplated that in a practical scenario, the transmitter and receiver are far enough apart that the signal transmitted from the transmit antenna can be considered to be incident parallel at any point on the receiver, and all antennas of the transmit antenna and the receiver can be considered to be in the same plane. In actual use, all antennas of the receiver can be horizontally placed, so that the aforementioned requirements can be met.

Radio frequency signals are transmitted by a transmitting antenna 320 of a transmitter in a wireless communication system using data frames 10, pointing from a receiver in the direction of the transmitting antenna 320

For transmitting antenna direction, transmitting antenna direction

Relative to a reference direction

Is defined as the azimuth angle α of the transmitting antenna 320, wherein the reference direction

The direction is preset, and the direction can be set arbitrarily according to actual needs. For convenience of description, the present disclosure refers to an orientation

Relative to the other direction

Angle of rotation of, mean from direction

Rotate in a counterclockwise direction to a direction

If the angle is greater than or equal to pi, 2 pi is subtracted. Thus, according to the preceding definition, the rotation angle in both directions has a value in the range of [ - π, π), if the direction is oriented in one direction

Relative to the direction

Is less than 0, corresponding to the direction

Rotate clockwise to the direction

The angle of rotation is equal to the absolute value of its angle of rotation.

In the present disclosure, the radio frequency signals received by the M antennas are transmitted by the transmitter through a single antenna. It should be noted that the single antenna transmission referred to in this disclosure means that the transmitter transmits using only one antenna, or if the transmitter transmits exactly the same signal using multiple antennas that are close enough in space, so that the transmission signals of these antennas are not substantially different from the transmission using only one antenna after being spatially combined.

In step 221, one of the M antennas is selected in turn to receive the rf signal for rf processing, so as to obtain a baseband signal corresponding to each antenna.

In this step, the rf module processes the rf signals received by the antennas in a time division multiplexing manner, that is, each antenna is alternately allocated a certain time interval, and in this time interval, the rf module only processes the rf signals received by the antennaThe radio frequency signal is processed. Fig. 5 exemplarily shows a schematic diagram of the present disclosure for selecting radio frequency signals received by M antennas to perform radio frequency processing alternately, in a time interval (t)_uM+m,t_uM+m+1) And selecting the radio frequency signal received by the mth antenna for processing, wherein u is an integer. It should be noted that the length of the time interval of each antenna may be the same or different, depending on the specific implementation. In addition, the selection of the rf processing time intervals for each antenna shown in fig. 5 is merely exemplary, and other selection schemes may be used, for example, the antennas may be selected in different orders at different times.

The processing of the radio frequency module comprises the steps of amplifying, down-converting, sampling and the like of the radio frequency signal to obtain a corresponding baseband signal. Representing the radio frequency signal received by the mth antenna as

The received signal obtained after down-conversion is denoted as r_m(T) sampling the received signal with a sampling period Δ T ═ T/S to obtain a baseband signal r_m(n Δ T), where T is the chirp spread spectrum modulation symbol period, S is the spreading factor, n denotes the sample number, and M is 0,1, …, and M-1.

In a conventional multi-antenna receiver, radio frequency signals received by each antenna are usually processed by respective radio frequency modules to obtain multiple baseband signals, and then the multiple baseband signals are processed. The method needs to allocate a radio frequency module for each antenna, so that the cost is high, and the baseband needs to process multiple paths of signals, so that the calculation amount is large and the power consumption is high. According to the method, only one radio frequency module is used, the radio frequency signal received by one of the M antennas is selected in turn to be subjected to radio frequency processing through the radio frequency module, and one radio frequency module does not need to be arranged for each antenna, so that the cost of the receiver can be reduced.

In step 222, it is detected whether a preamble is present in the baseband signal within the current time window.

Processing the baseband signal obtained after being processed by the radio frequency module within a period of timeAnd detecting whether the preamble exists in the baseband signal in the period of time. For convenience of description, the period of time is referred to as a time window, and each time window may be represented by a start position of the time window and a length of the time window. For the example shown in FIG. 5, the starting position of the time window of the qth time window is w_qThe time window length is W_qWherein q is an integer. It should be noted that the lengths of the different time windows may be the same or different, and may be selected according to actual needs. In addition, although fig. 5 shows adjacent time windows overlapping each other, they may not overlap. In each time window, the number of time periods in which the received signal of each antenna is subjected to radio frequency processing may be the same or different. In other words, within a time window, some antennas may be allocated several discrete time periods for rf processing.

And the baseband module sequentially processes the baseband signals in the time window, if a lead code is detected in a certain time window, the subsequent steps are executed, otherwise, if the lead code is not detected, the baseband signals in the next time window are continuously processed until the lead code is detected.

For convenience of description, the baseband signal corresponding to each antenna in the current time window may be divided into a plurality of received symbols according to the chirp spread spectrum modulation symbol period, and an nth sample of an l-th received symbol of an m-th antenna may be represented as r_m,l[n]Where 0. ltoreq. n<And S. It is then detected whether a preamble is present within the current time window in the manner described below.

Firstly, calculating a corresponding matching symbol for each receiving symbol of each antenna according to the following formula to obtain a matching signal of each antenna:

c_m,l[n]＝r_m,i[n]*p^*[n]， (4)

wherein, c_m,l[n]，0≤n<S, is the l matching symbol of the m antenna, p [ n ]]The nth sampling value of the local signal constructed according to the preset chirp spread spectrum modulation symbol in the lead code of the frame header is obtained, namely if the preset chirp spread spectrum modulation symbol in the lead code is

0≤t<T, then

Then, calculating discrete Fourier transform for each matching symbol of each antenna to obtain frequency domain matching symbols:

wherein C is_m,l[k]K is 0,1, …, S-1, the ith frequency-domain matching symbol of the mth antenna, and k is the frequency point number.

Next, for the adjacent frequency domain matching symbols of each antenna, calculating corresponding frequency domain matching autocorrelation symbols according to the following formula:

wherein D is_m,l[k]And k is 0,1, …, S-1, which is the l-th frequency-domain matched autocorrelation symbol of the m-th antenna. It should be noted that adjacent frequency-domain matching symbols refer to two frequency-domain matching symbols that are consecutive in time. If a certain antenna is allocated with two time periods of radio frequency processing in a time window, the last frequency-domain matching symbol of the previous time period is not continuous in time with the first frequency-domain matching symbol of the next time period, and the two frequency-domain matching symbols do not belong to adjacent frequency-domain matching symbols.

Then, accumulating all frequency domain matching autocorrelation symbols of the M antennas according to frequency points according to the following formula to obtain a summed frequency domain matching autocorrelation symbol:

E[k]＝∑_m∑_lD_m,l[k],k＝0,1,…,S-1， (7)

where E [ k ], k ═ 0,1, …, S-1, are the summed matching autocorrelation symbols.

Finally, the frequency domain peak value E of the summed frequency domain matched autocorrelation symbols is judged^maxAnd whether a preset threshold value is exceeded, if the preset threshold value is exceeded, the preamble is considered to be detected, otherwise, the preamble is considered not to be detected. Wherein E is^maxFor maximum value of the modulus of the summed matched autocorrelation symbols at each frequency point, i.e. E^max＝max{|E[k]I, k is 0,1, …, S-1, and the frequency point number corresponding to the frequency domain peak is k_max。

If no preamble is detected, the time window is slid forward by a preset time length in step 223, and then steps 221 and 222 are repeatedly performed until a preamble is detected. The preset time length of the forward sliding of the time window can be selected according to actual needs so as to balance the calculation complexity and the detection speed. Preferably, the preset time length may be a multiple of the chirp spread spectrum modulation symbol period.

In step 230, after the preamble is detected, a signal phase difference is estimated for each of the H angle-measuring antenna pairs.

Since the distance between each antenna and the transmitting antenna may be different, there may be a certain phase difference between signals received by different antennas. In the present disclosure, the signal phase difference of each goniometric antenna pair refers to a difference between the phases of the first antenna and the second antenna of the goniometric antenna pair.

In addition, because the clocks of the receiver and the transmitter are independent of each other, the clock frequencies of the receiver and the transmitter usually have a deviation, and after the receiver converts the received radio frequency signal into a baseband signal, the baseband signal has a frequency deviation f relative to the transmitter_o. Frequency deviation f_oCan be expressed as the sum of integer multiples and decimal multiples of 1/T, i.e. f_o＝(_i+_f) a/T, wherein_iIs an integer, referred to as an integer frequency offset,_fis small, and-0.5 is less than or equal to_f<0.5, called fractional frequency offset.

In some embodiments, a fractional frequency offset estimation may be performed based on the calculation of step 222 with respect to the summed frequency domain matched autocorrelation symbols and further the signal phase difference for each goniometric antenna pair is estimated.

First, the estimated fractional frequency offset is estimated by the following formula_f：

_f＝angle(E[k_max])/(2π)， (8)

Wherein, E [ k_max]Matching the kth autocorrelation symbol for the summed frequency domain_maxThe value of each frequency bin, function angle (x), represents the phase angle of complex number x, which ranges from [ - π, π).

Estimating fractional frequency offset_fThen, the k-th of the autocorrelation symbol is matched to the frequency domain of each antenna according to the following formula_maxAnd (3) carrying out decimal frequency multiplication deviation compensation on each frequency point to obtain decimal frequency multiplication deviation compensation results:

wherein, D'_m,l[k_max]Matching the kth frequency domain of the autocorrelation symbol for the ith antenna_maxThe result, p, obtained by fractional frequency offset compensation at each frequency point_m,lThe starting position of the l-th received symbol for the m-th antenna is offset from the number of symbols of the starting position of the current time window, i.e., by how many chirp spread spectrum modulation symbol periods.

Then, the frequency domain of each antenna is matched with the k-th of the self-correlation symbol_maxAccumulating the decimal frequency multiplication deviation compensation results of the frequency points to obtain decimal frequency multiplication deviation compensation accumulation results of each antenna:

F_m[k_max]＝∑_lD′_m,l[k_max]， (10)

wherein, F_m[k_max]And compensating and accumulating the result of the decimal frequency multiplication offset of the mth antenna. In equation (10), the summation range may be several frequency-domain matched autocorrelation symbols of the mth antenna, and preferably, the summation range is all frequency-domain matched autocorrelation symbols of the mth antenna.

Finally, the signal phase difference of each angle measuring antenna pair is estimated according to the following formula:

wherein h0 is the serial number of the first antenna in the h angle-measuring antenna pair, h1 is the serial number of the second antenna in the h angle-measuring antenna pair, and delta theta_hAnd estimating the signal phase difference of the h-th angle measuring antenna pair.

In some embodiments, the fractional frequency offset estimation is not performed, and the k < th > of the autocorrelation symbol is directly matched based on the frequency domain of each antenna_maxAnd estimating the signal phase difference of each angle measuring antenna pair by each frequency point. The method can be suitable for fractional frequency offset_fThe smaller the case.

First, the k-th of the autocorrelation symbol is matched to the frequency domain of each antenna according to the following formula_maxAccumulating the frequency points to obtain a decimal frequency multiplication offset compensation accumulation result of each antenna:

F_m[k_max]＝∑_lD_m,l[k_max]， (12)

wherein, F_m[k_max]Frequency domain matching of the kth of the autocorrelation symbol for the mth antenna_maxAnd accumulating results of the frequency points.

Then, the signal phase difference of each angle measuring antenna pair is estimated according to the following formula:

wherein h0 and h1 are respectively the serial numbers of the first antenna and the second antenna of the h angle-measuring antenna pair, and delta theta_hAnd estimating the signal phase difference of the h-th angle measuring antenna pair.

In some other embodiments, the baseband signals of multiple time windows may be utilized to perform decimal frequency offset estimation, and then the decimal frequency offset estimation result is utilized to perform signal compensation, and the signal phase difference of each angle measuring antenna pair is estimated based on the compensated signals. The method can be suitable for the condition that the signal-to-noise ratio of the received signal is low, and the estimation result of the signal phase difference of each angle measuring antenna pair is more accurate in the mode.

Firstly, continuously sliding Q time windows, and selecting a radio frequency signal received by one of M antennas in each time window in turn to perform radio frequency processing to obtain a baseband signal corresponding to each antenna, wherein Q is a preset positive integer, and the time window sliding and the antenna selection are performed so that the baseband signal corresponding to each antenna in the current time window comprises at least 2 continuous chirp spread spectrum modulation symbol periods. For each time window Q, Q ═ 0,1, …, Q-1, the process proceeds as follows in steps 231-:

in step 231, the baseband signal corresponding to each antenna in the qth time window is divided into a plurality of received symbols according to the chirped spread spectrum modulation symbol period, where the nth sample value of the ith received symbol of the mth antenna is r_q,m,l[n]，0≤n<S，m＝0,1,…,M-1。

In step 232, for each received symbol of each antenna in the qth time window, a corresponding matching symbol is calculated according to the following formula to obtain a matching signal of each antenna:

c_q,m,l[n]＝r_q,m,l[n]*p^*[n]， (14)

wherein, c_q,m,l[n]N sample value of the l matching symbol for the m antenna, p n]And the nth sampling value of the local signal is constructed according to the preset chirp spread spectrum modulation symbol in the preamble of the frame header.

In step 233, a discrete fourier transform is computed for the matched symbols of each antenna to obtain corresponding frequency domain matched symbols:

wherein C is_q,m,l[k]And matching the value of the k frequency point of the symbol for the l frequency domain of the m antenna.

In step 234, for the frequency-domain matched symbol of each antenna in the qth time window, the corresponding frequency-domain matched autocorrelation symbol is calculated according to the following formula:

wherein D is_q,m,l[k]And matching the value of the k frequency point of the autocorrelation symbol for the l frequency domain of the m antenna.

In step 235, accumulating all frequency-domain matched autocorrelation symbols of the M antennas in the qth time window according to frequency points according to the following formula to obtain a summed frequency-domain matched autocorrelation symbol:

E_q[k]＝∑_m∑_lD_q,m,l[k],k＝0,1,…,S-1， (17)

wherein E is_q[k]Is the value of the k-th bin of the summed matched autocorrelation symbols.

After obtaining the summed matched autocorrelation symbols for each time window, the frequency domain peak E of the summed frequency domain matched autocorrelation symbols for Q time windows is then searched^maxI.e. E^nax＝max{|E_q[k]0,1, …, S-1; q is 0,1, …, Q-1, wherein the frequency domain peak is located at the frequency point with the number k_maxThe sequence number of the time window is q_max。

Then, the decimal frequency deviation is estimated according to the following formula_f：

_f＝angle(E_qmax[k_max])/(2π)。 (18)

Then, the q-th equation is expressed as follows_maxFrequency domain matched autocorrelation symbol kth for each antenna within a time window_maxCarrying out decimal frequency multiplication offset compensation on each frequency point:

wherein, D'_m,l[k_max]Is the q th_maxThe kth frequency domain of the ith frequency domain matched autocorrelation symbol of the mth antenna in each time window_maxThe result, p, obtained by fractional frequency offset compensation at each frequency point_m,lIs the q th_maxThe starting position of the ith received symbol of the mth antenna within the time window is offset from the number of symbols at the starting position of the current time window.

Then, the q-th equation is expressed as follows_maxWithin a time windowIs matched to the k-th of the autocorrelation symbol_maxAccumulating the decimal frequency multiplication deviation compensation results of the frequency points to obtain decimal frequency multiplication deviation compensation accumulation results of each antenna:

F_m[k_max]＝∑_lD′_m,l[k_max]， (20)

wherein, F_m[k_max]Is the q th_maxAnd compensating and accumulating the result of the decimal frequency multiplication offset of the mth antenna in the time window.

Referring now to FIG. 4, the h-th goniometric antenna pair includes a spacing d_hA first antenna 410 and a second antenna 411. Defining the direction of the h-th goniometric antenna pair

The direction of the normal to the h-th goniometric antenna pair is the direction in which the first antenna 410 points towards the second antenna 411

To be composed of

In the direction obtained after a 90 ° rotation anticlockwise. The transmitting antenna 420 is located at position A, the transmitting antenna direction

Relative to a reference directionIs α (not shown in fig. 4, see also fig. 3),direction of transmitting antenna

Relative to the normal direction

Has a rotation angle of psi_h。

Due to the different distances between the transmitting antenna 420 and the first antenna 410 and the second antenna 411, there is a time difference between the arrival of the signals transmitted by the transmitting antenna 420 at the first antenna 410 and the arrival of the signals at the second antenna 411, so that there is a phase difference Δ θ between the signals received by the first antenna 410 and the second antenna 411_hAnd Δ θ_h＝2πd_hsin(ψ_h) And/lambda. According to the formula, can obtain

Wherein arcsin (x) represents an arcsine function of x.

However, the goniometric antenna pair cannot distinguish the mirror position B at position A of the transmit antenna 420₀In other words, from the mirror position B₀The phase difference of the transmitted signal arriving at the first antenna 410 and the second antenna 411 of the h-th goniometric antenna pair is the same as the phase difference of the signal arriving at both antennas from position a. Mirror 421 direction of the transmitting antenna relative to the normal direction

Is of phi'_h＝π-ψ_h. Therefore, the transmit antenna direction cannot be uniquely determined directly from equation (22)

The method 20 of the present disclosure obtains an azimuth estimate for the transmit antenna by eliminating the mirror position of the transmit antenna using multiple pairs of angle-measuring antennas, which is described in detail in step 240.

Specifically, after the signal phase difference of each angle-measuring antenna pair is estimated, in step 240, the azimuth angle estimation value of the transmitting antenna is determined based on the signal phase difference estimation results of H angle-measuring antenna pairs. This step may be implemented by sub-steps 241 to 244.

In step 241, candidate azimuth angle pairs based on each pair of angle-measuring antennas are calculated based on the signal phase difference estimation result of the pair.

Each candidate azimuth pair comprises a first candidate azimuth and a second candidate azimuth, and the first candidate azimuth and the second candidate azimuth based on each goniometric antenna pair are calculated according to the following formula:

wherein, ξ_h0And ξ_h1A first candidate azimuth angle and a second candidate azimuth angle, gamma, based on the h-th goniometric antenna pair, respectively_hFor the normal direction of the h-th angle-measuring antenna pair

Relative to a reference direction

The angle of rotation of.

In step 242, a candidate azimuth set will be formed based on the first candidate azimuth and the second candidate azimuth of all H goniometric antenna pairs

In other words, the set of candidate azimuth angles

As a union of sets consisting of a first candidate azimuth and a second candidate azimuth based on each goniometric antenna pair, i.e.

Wherein (ξ)_h0,ξ_h1Denotes a first candidate position based on h goniometric antenna pairsCorner ξ_h0And a first candidate azimuth ξ_h1A set of compositions.

In step 243, from the set of candidate azimuths

Determining a subset of azimuths Φ.

As can be seen from fig. 3, for each goniometric antenna pair there is a mirror image antenna of the transmit antenna, for example 321 for the 0 th goniometric antenna pair, of the transmit antenna 320, which is located at position B₀(ii) a For the 1 st goniometric antenna pair, the mirror antenna of the transmit antenna 320 is 322, which is located at position B₁. Thus, a set of candidate azimuth angles

Including the orientations of these mirror antennas, it is necessary to exclude the orientations of the mirror antennas to obtain an azimuth subset Φ of the transmit antennas in order to obtain an estimate of the orientation of the transmit antennas.

In some embodiments, the set of candidate azimuth angles may be selected from

Is equal to the predetermined number of subsets, the subset whose density of elements is higher than the predetermined threshold is selected as the azimuth subset Φ.

In the present disclosure, the element density of an azimuth set refers to the density of all elements in the set, and the higher the density, the smaller the gap between the elements is, or the closer the gap is. The density of the set Φ can be measured by a density function D (Φ), the smaller the value of the density function D (Φ), the higher the density of the set elements. The density function D (Φ) can be defined in various ways as long as the aforementioned properties are reflected. For example, in some embodiments, a density function may be defined as

Wherein phi is_iThe i-th element of the set phi is represented,

representing the mean of all elements of the set phi. In other embodiments, the intensity function may be defined as D (Φ) ═ max (Φ) -min (Φ) |, (25)

Where max (Φ) and min (Φ) represent the maximum and minimum values of the elements of the set, respectively. There are various implementations of the intensity function, and the present disclosure does not limit the specific manner of the intensity function D (Φ).

Preferably, in some further embodiments, the set of candidate azimuth angles may be selected from

The number of elements of (1) is equal to the preset number of subsets, and the subset with the highest concentration of the elements is selected as the azimuth subset Φ.

In some other embodiments, one of the candidate azimuths may be excluded from the subset of azimuths Φ, e.g., one of the candidate azimuths may be randomly excluded, or the candidate azimuths excluded from having a larger average distance to all elements of the subset of azimuths Φ, considering that when the direction of the transmit antenna is close to the direction of the pair of azimuths, the transmit antenna is very close to the direction of its mirror antenna, resulting in two candidate azimuths based on the same pair of azimuths in the subset of azimuths Φ, thereby increasing the weight of the pair of azimuths.

In step 244, an azimuth estimate for the transmit antenna is calculated based on the azimuth subset Φ.

In other embodiments, an element may be randomly selected from the subset of azimuth angles Φ as the estimate of the azimuth angle for the transmit antenna.

Preferably, in some embodiments, an average value of all elements in the azimuth subset Φ may be calculated, and the average value is used as the azimuth estimate for the transmit antenna. This method results in estimates with smaller variance.

Besides, there are other methods for calculating the azimuth angle estimation value of the transmitting antenna, and any method for calculating the estimation value of the azimuth angle of the transmitting antenna based on the azimuth angle subset Φ should be considered to fall within the scope of the present disclosure.

As can be seen from the above steps, the angle measurement method 20 disclosed herein, for a communication system based on chirped spread spectrum modulation, utilizes the characteristic that a preamble contains repeated chirped spread spectrum modulation symbols, selects one received radio frequency signal of multiple antennas in turn to perform radio frequency processing to obtain a baseband signal, then divides the baseband signal of each antenna into multiple received symbols, matches each received symbol with a local signal reconstructed by using the preamble, and performs autocorrelation on the matched symbols, divides the multiple antennas into multiple angle measurement antenna pairs, estimates a signal phase difference of each angle measurement antenna pair, then estimates a candidate azimuth angle set based on each angle measurement antenna pair according to a phase difference estimation result, and then obtains an azimuth angle estimation of a transmitting antenna based on the candidate azimuth angle set. The method 20 of the present disclosure has a lower hardware implementation complexity compared to conventional multi-antenna goniometry methods.

The present disclosure also provides an angle measuring device 60, as shown in fig. 6, the angle measuring device 60 includes M antennas 601, a Multiplexer (MUX)602, a radio frequency module 603, and a baseband module 604. The angle measuring apparatus 60 may be used to implement the angle measuring method 20 of the embodiment of the present disclosure.

Each antenna 601 is used for receiving a radio frequency signal transmitted by a transmitter in a wireless communication system using a data frame 10; the multiplexer 602 is configured to select one of the radio frequency signals received by the M antennas for output according to the antenna selection control signal; the rf module 603 is configured to perform rf processing on the rf signal output by the multiplexer 602 to obtain a corresponding baseband signal; the baseband module 604 is coupled to the multiplexer 602 and the radio frequency module 603 and configured to process the baseband signal to generate an antenna selection control signal and estimate an azimuth angle of the transmit antenna.

In some embodiments, the goniometer device 60 may further include M low Noise amplifiers (L ow Noise Amplifier, L NA)605, each low Noise Amplifier 605 being coupled between the antenna 601 of a respective signal branch and the multiplexer 602, configured to amplify the radio frequency signal received by the antenna 601.

The processing of the baseband signal by the baseband module 604 includes performing steps 610 through 630.

In step 610, it is detected whether a preamble is present in the baseband signal.

Generating an antenna selection control signal which selects one of the M paths of radio frequency signals after phase compensation in turn to output according to a preset mode, detecting whether a baseband signal in a current time window comprises the lead code, and if the lead code is not detected, sliding the time window forward for a preset time length; this step is then repeated until a preamble is detected.

The process of detecting whether the baseband signal in the current time window includes the preamble may employ various embodiments of step 222 of the angle measurement method 20 of the present disclosure, and is not described herein again.

In step 620, in response to detecting the preamble, a signal phase difference is estimated for each of the H angle-measuring antenna pairs.

The process of estimating the signal phase difference for each angle-measuring antenna pair may employ various embodiments of step 230 of the angle-measuring method 20 of the present disclosure, and will not be described herein again.

In step 630, an azimuth angle estimate for the transmit antenna is determined based on the signal phase difference estimates for each of the H angle-measuring antenna pairs.

The process of determining the azimuth angle estimation value of the transmitting antenna based on the signal phase difference estimation result of each of the H angle measurement antenna pairs may adopt various embodiments of step 240 of the angle measurement method 20 of the present disclosure, and details are not repeated here.

Other variations to the disclosed embodiments can be understood and effected by those skilled in the art from a review of the specification, the disclosure, the drawings, and the appended claims. In the claims, the word "comprising" does not exclude other elements or steps, and the words "a" or "an" do not exclude a plurality. In the practical application of the present application, one element may perform the functions of several technical features recited in the claims. Any reference signs in the claims shall not be construed as limiting the scope.

Claims

1. A method of measuring an angle, the method comprising:

receiving a radio frequency signal transmitted from a transmitting antenna by using M antennas, wherein the radio frequency signal carries a wireless data frame, the wireless data frame comprises a frame header modulated by using chirp spread spectrum, and the frame header comprises N_pThe antenna array comprises a lead code consisting of repeated preset chirp spread spectrum modulation symbols, wherein the phases of two adjacent chirp spread spectrum modulation symbols of the lead code are continuous, the M antennas comprise H angle measurement antenna pairs, each angle measurement antenna pair comprises a first antenna and a second antenna, the distance between the first antenna and the second antenna is not more than lambda/2, H and Np are integers more than or equal to 2, and lambda is the carrier wavelength of the radio-frequency signal;

selecting a radio frequency signal received by one of the M antennas in turn according to a preset mode to perform radio frequency processing so as to obtain a baseband signal corresponding to each antenna, detecting whether a lead code exists in the baseband signal in a current time window, and sliding the time window forward for a preset time length in response to the fact that the lead code is not detected; repeating this step until the preamble is detected;

in response to detecting the preamble, estimating a signal phase difference of each of the H angle-measuring antenna pairs, wherein the signal phase difference estimation result of the H-th angle-measuring antenna pair is Δ θ_h(ii) a And

and determining the azimuth angle estimated value of the transmitting antenna based on the signal phase difference estimation results of the H angle measurement antenna pairs.

2. The angle measurement method according to claim 1, wherein the predetermined time length is a multiple of a period of the chirp spread spectrum modulation symbol.

3. The method of claim 1, wherein the M antennas are located in the same plane.

4. The method of claim 1, wherein the step of detecting whether the baseband signal within the current time window includes the preamble comprises:

dividing the baseband signal corresponding to each antenna in the current time window into a plurality of receiving symbols according to the chirp spread spectrum modulation symbol period, wherein the nth sample of the ith receiving symbol of the mth antenna is r_m，l[n]N is more than or equal to 0 and is less than S, S is the number of samples included in a chirp spread spectrum modulation symbol period, and M is 0, 1.

Calculating a corresponding matching symbol for each received symbol of each antenna according to the following formula to obtain a matching signal of each antenna: c. C_m，l[n]＝r_m，l[n]*p^*[n]Wherein c is_m，l[n]N sample value of the l matching symbol for the m antenna, p n]The nth sampling value of the preset chirp spread spectrum modulation symbol is obtained;

calculating discrete Fourier transform for the matched symbols of each antenna to obtain corresponding frequency-domain matched symbols:

wherein C is_m，l[k]Matching the value of the kth frequency point of the symbol for the ith frequency domain of the mth antenna;

calculating corresponding frequency domain matching autocorrelation symbols for the adjacent frequency domain matching symbols of each antenna according to the following formula:

wherein D is_m，l[k]Matching the value of the kth frequency point of the autocorrelation symbol for the ith frequency domain of the mth antenna;

accumulating the frequency-based points of all the frequency-domain matched autocorrelation symbols of the M antennas according to the following formula to obtain a summed frequency-domain matched autocorrelation symbol: e [ k ]]＝∑_m∑_lD_m，l[k]K is 0, 1.., S-1, wherein E [ k [ ] is]The value of the k frequency point of the summed matched autocorrelation symbols; and

determining a frequency domain peak E of the summed frequency domain matched autocorrelation symbols^maxWhether a preset threshold value is exceeded, if the preset threshold value is exceeded, the preamble is considered to be detected, otherwise, the preamble is considered not to be detected, wherein E^max＝max{|E[k]0,1,., S-1}, and the frequency point number corresponding to the frequency domain peak is k_max。

5. The angle measurement method according to claim 4, wherein the step of estimating the signal phase difference of each of the H angle measurement antenna pairs comprises:

matching the frequency domain of the autocorrelation symbol for each of the M antennas by the following equation_maxAccumulating the frequency points to obtain a decimal frequency multiplication offset compensation accumulation result of each antenna: f_m[k_max]＝∑_lD_m，l[k_max]Wherein F is_m[k_max]Frequency domain matching of the kth of the autocorrelation symbol for the mth antenna_maxAccumulating results of the frequency points;

estimating the signal phase difference of the H antenna pair in the H angle measuring antenna pairs according to the following formula:

h0 is the serial number of the first antenna of the h angle-measuring antenna pair, and h1 is the serial number of the second antenna of the h angle-measuring antenna pair.

6. The angle measurement method according to claim 4, wherein the step of estimating the signal phase difference of each of the H angle measurement antenna pairs comprises:

estimating fractional frequency offset according to the following formula_f：_f＝angle(E[k_max])/(2π)；

Press and press withMatching the frequency domain of the autocorrelation symbol for each of the M antennas by the following equation_maxCarrying out decimal frequency multiplication offset compensation on each frequency point:

wherein, D'_m，l[k_max]Matching the kth frequency domain of the autocorrelation symbol for the ith antenna_maxThe result, p, obtained by fractional frequency offset compensation at each frequency point_m，lThe number of symbols is offset from the starting position of the current time window for the starting position of the ith received symbol of the mth antenna;

matching the frequency domain of the autocorrelation symbol for each of the M antennas by the following equation_maxAccumulating the decimal frequency multiplication deviation compensation result of each frequency point to obtain the decimal frequency multiplication deviation compensation accumulation result of each antenna: f_m[k_max]＝∑_lD′_m，l[k_max]Wherein F is_m[k_max]Compensating and accumulating the result of the decimal frequency multiplication offset of the mth antenna;

estimating the signal phase difference of the H angle measuring antenna pair in the H angle measuring antenna pairs according to the following formula:

7. The angle measurement method according to claim 1, wherein the step of estimating the signal phase difference of each of the H angle measurement antenna pairs comprises:

continuously sliding Q time windows, and selecting a radio frequency signal received by one of the M antennas in each time window in turn to perform radio frequency processing to obtain a baseband signal corresponding to each antenna, wherein Q is a preset positive integer, and the time window sliding and the antenna selection are performed so that the baseband signal corresponding to each antenna of the M antennas in the current time window comprises at least 2 continuous chirp spread spectrum modulation symbol periods;

dividing the baseband signal corresponding to each antenna in the qth time window into a plurality of receiving symbols according to the chirp spread spectrum modulation symbol period, wherein the nth sample of the ith receiving symbol of the mth antenna is r_q，m，l[n]N is more than or equal to 0 and less than S, S is the number of samples included in one chirp spread spectrum modulation symbol period, M is 0, 1.

For each received symbol of each antenna in the qth time window, calculating a corresponding matching symbol according to the following formula to obtain a matching signal of each antenna: c. C_q，m，l[n]＝r_q，m，l[n]*p^*[n]Wherein c is_q，m，l[n]N sample value of the l matching symbol for the m antenna, p n]The nth sampling value of the preset chirp spread spectrum modulation symbol is obtained;

wherein C is_q，m，l[k]Matching the value of the kth frequency point of the symbol for the ith frequency domain of the mth antenna;

calculating the corresponding frequency domain matching autocorrelation symbol for the frequency domain matching symbol of each antenna in the qth time window according to the following formula:

wherein D is_q，m，l[k]Matching the value of the kth frequency point of the autocorrelation symbol for the ith frequency domain of the mth antenna;

accumulating the frequency-point-based autocorrelation symbols of all the frequency-domain matched autocorrelation symbols of the M antennas in the qth time window according to the following formula to obtain a summed frequency-domain matched autocorrelation symbol: e_q[k]＝∑_m∑_lD_q，m，l[k]K is 0, 1.., S-1, wherein E_q[k]The value of the k frequency point of the summed matched autocorrelation symbols;

searching for a frequency domain peak E of the summed frequency domain matched autocorrelation symbols for the Q time windows^maxWherein E is^max＝max{|E_q[k]I, k ═ 0,1,. ·, S-1; q is 0,1,., Q-1}, and the frequency point number corresponding to the frequency domain peak is k_maxSequence number of time window q_max；

Estimating fractional frequency offset according to the following formula_f：

According to the following formula for the q_maxFrequency domain matched autocorrelation symbol kth for each of the M antennas within a time window_maxCarrying out decimal frequency multiplication offset compensation on each frequency point:

wherein, D'_m，l[k_max]Is the q th_maxThe kth frequency domain of the ith frequency domain matched autocorrelation symbol of the mth antenna in each time window_maxThe result, p, obtained by fractional frequency offset compensation at each frequency point_m，lIs the q th_maxThe starting position of the ith receiving symbol of the mth antenna in the time window is offset relative to the symbol number of the starting position of the current time window;

according to the following formula for the q_maxFrequency domain matched autocorrelation symbol kth for each of the M antennas within a time window_maxAccumulating the decimal frequency multiplication deviation compensation result of each frequency point to obtain the decimal frequency multiplication deviation compensation accumulation result of each antenna: f_m[k_max]＝∑_lD′_m，l[k_max]Wherein F is_m[k_max]Is the q th_maxThe decimal frequency multiplication offset compensation accumulation result of the mth antenna in each time window;

h0 is the serial number of the first antenna in the h angle measuring antenna pair, and h1 is the serial number of the second antenna in the h angle measuring antenna pair.

8. The angle measurement method according to any one of claims 1 to 7, wherein the step of determining the azimuth angle estimation values of the transmission antennas based on the signal phase difference estimation results of the H angle measurement antenna pairs comprises:

calculating a candidate azimuth angle pair based on each of the H angle-measuring antenna pairs based on the signal phase difference estimation result of the angle-measuring antenna pair, wherein the candidate azimuth angle pair comprises a first candidate azimuth angle and a second candidate azimuth angle, and the first candidate azimuth angle calculated based on the H angle-measuring antenna pair is ξ_h0And a second candidate azimuth of ξ_h1；

Forming a candidate azimuth set based on first candidate azimuths and second candidate azimuths of all angle-measuring antenna pairs of the H angle-measuring antenna pairs

Wherein the content of the first and second substances,

from the set of candidate azimuths

Determining a subset of azimuth angles phi; and

calculating an azimuth estimate for the transmit antenna based on the azimuth subset Φ.

9. The method of claim 8, wherein said deriving from said set of candidate azimuths

The step of determining the subset of azimuths Φ comprises:

from the set of candidate azimuths

Is equal to the preset number of subsets, the subset whose element density is higher than the preset threshold is selected as the azimuth subset Φ.

10. The method of claim 8, wherein said deriving from said set of candidate azimuths

The step of determining the subset of azimuths Φ comprises:

from the set of candidate azimuths

11. The angle measurement method according to claim 8, wherein the step of calculating a candidate azimuth angle pair based on each of the H angle measurement antenna pairs based on the signal phase difference estimation result of the angle measurement antenna pair comprises:

for each of the H goniometric antenna pairs, calculating a first candidate azimuth angle and the second candidate azimuth angle based on the goniometric antenna pair according to the following formulas:

wherein d is_hIs the distance, gamma, between the first and second antennas of the h-th goniometric antenna pair_hIs the rotation angle of the normal direction of the h-th goniometric antenna pair relative to the reference direction.

12. An angle measuring apparatus, characterized in that the angle measuring apparatus comprises:

the M antennas are used for receiving radio frequency signals transmitted from transmitting antennas, the radio frequency signals carry wireless data frames, the wireless data frames comprise frame headers modulated by using chirp spread spectrum, H angle measuring antenna pairs are included in the M antennas, each angle measuring antenna pair comprises a first antenna and a second antenna, the distance between the first antenna and the second antenna is not more than lambda/2, H is an integer larger than or equal to 2, and lambda is the carrier wavelength of the radio frequency signals;

the multiplexer is configured to select one of the radio frequency signals received by the M antennas to output according to an antenna selection control signal;

a radio frequency module configured to perform radio frequency processing on the radio frequency signal output by the multiplexer to obtain a corresponding baseband signal; and

a baseband module coupled to the multiplexer, the baseband module configured to process the baseband signal to generate the antenna selection control signal and determine an azimuth estimate for the transmit antenna.

13. The goniometric device of claim 12, wherein the M antennas are located in the same plane.

14. An apparatus according to claim 12, wherein the frame header comprises a reference number N_pA preamble composed of a plurality of repeated preset chirp spread spectrum modulation symbols, wherein phases between two adjacent chirp spread spectrum modulation symbols of the preamble are continuous, wherein N is_pThe baseband module is an integer greater than or equal to 2, and the processing the baseband signal by the baseband module includes:

detecting whether a preamble exists in the baseband signal;

and determining an azimuth angle estimated value of the transmitting antenna based on the signal phase difference estimation result of each angle measuring antenna pair in the H angle measuring antenna pairs.

15. The goniometric device of claim 14, wherein the baseband module detecting whether a preamble is present in the baseband signal comprises:

generating the antenna selection control signal for alternately selecting the radio frequency signal received by one of the M antennas to be output according to a preset mode, detecting whether the baseband signal in the current time window comprises the preamble, responding to the fact that the preamble is not detected, sliding the time window forward for a preset time length, and then repeatedly executing the steps until the preamble is detected.

16. The angle measurement device of claim 15, wherein the predetermined time duration is a multiple of a period of a chirped spread spectrum modulation symbol.

17. The goniometric device of claim 15, wherein the baseband module detects whether the baseband signal within a current time window includes the preamble comprises:

18. The goniometric device of claim 17, wherein the baseband module to estimate the signal phase difference for each of the H goniometric antenna pairs comprises:

19. The goniometric device of claim 17, wherein the baseband module to estimate the signal phase difference for each of the H goniometric antenna pairs comprises:

Matching the frequency domain of the autocorrelation symbol for each of the M antennas by the following equation_maxCarrying out decimal frequency multiplication offset compensation on each frequency point:

matching the frequency domain of the autocorrelation symbol for each of the M antennas by the following equation_maxAccumulating the decimal frequency multiplication offset compensation results of each frequency point to obtain each frequency pointThe decimal frequency multiplication offset compensation accumulation result of the antenna is as follows: f_m[k_max]＝∑_lD′_m，l[k_max]Wherein F is_m[k_max]Compensating and accumulating the result of the decimal frequency multiplication offset of the mth antenna;

20. The goniometric device of claim 14, wherein the baseband module to estimate the signal phase difference for each of the H goniometric antenna pairs comprises:

continuously sliding Q time windows, and generating an antenna selection control signal for alternately selecting and outputting a radio frequency signal received by one of the M antennas in each time window, wherein Q is a preset positive integer, and the time window sliding and the antenna selection should enable a baseband signal corresponding to each antenna of the M antennas in the current time window to comprise at least 2 continuous chirp spread spectrum modulation symbol periods;

Estimating fractional frequency offset according to the following formula_f：

estimating the signal phase difference of each angle measuring antenna pair in the H angle measuring antenna pairs according to the following formula:

21. The goniometric device of any one of claims 14 to 20, wherein the baseband module determines the azimuth angle estimate for the transmit antenna based on the signal phase difference estimates for the H goniometric antenna pairs comprises:

calculating a candidate based on each of the H angle-measuring antenna pairs based on a signal phase difference estimation result of the angle-measuring antenna pairSelecting an azimuth pair comprising a first candidate azimuth and a second candidate azimuth, wherein the first candidate azimuth based on the h-th goniometric antenna pair is ξ_h0And a second candidate azimuth of ξ_h1；

Wherein the content of the first and second substances,

from the set of candidate azimuths

Determining a subset of azimuth angles phi; and

22. The goniometric device of claim 21, wherein the baseband module derives from the set of candidate azimuths

Determining the subset of azimuths Φ comprises:

from the set of candidate azimuths

23. The goniometric device of claim 21, wherein the baseband module derives from the set of candidate azimuths

Determining the subset of azimuths Φ comprises:

from the set of candidate azimuths

24. The goniometric device of claim 21, wherein the baseband module to compute candidate azimuth angle pairs based on each of the H goniometric antenna pairs based on the signal phase difference estimates for that pair comprises: