CN111416540B - Multi-level converter midpoint potential rapid balance control system and method - Google Patents

Multi-level converter midpoint potential rapid balance control system and method Download PDF

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CN111416540B
CN111416540B CN202010342336.1A CN202010342336A CN111416540B CN 111416540 B CN111416540 B CN 111416540B CN 202010342336 A CN202010342336 A CN 202010342336A CN 111416540 B CN111416540 B CN 111416540B
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coordinate system
capacitor
modulation signal
abc coordinate
ftc
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CN111416540A (en
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张承慧
陈志远
邢相洋
段彬
付有良
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Shandong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H02M7/487—Neutral point clamped inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/66—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal
    • H02M7/68—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters
    • H02M7/72—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/79—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/797—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output with possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only

Abstract

The invention provides a system and a method for controlling the rapid balance of midpoint potential of a multilevel converter, which belong to the technical field of multilevel converters, and are characterized in that the rapid balance control of the midpoint potential is finally realized by detecting the voltage values of upper and lower capacitors in real time, sending the difference value to a finite time controller, outputting a zero sequence component for adjusting the action time of a P/N type redundant small vector in real time, and assisting a zero sequence injection simplified SVPWM (space vector pulse width modulation) strategy to adjust the action time of a small vector with strong midpoint potential adjusting capability in real time; the method can realize the rapid balance control of the midpoint potential, effectively reduce the voltage stress borne by the direct-current side capacitor and the power loop switch tube, and greatly reduce the harmonic content of the alternating-current side voltage and current and the harmonic pollution to the power grid; meanwhile, the method does not depend on a system model, and is strong in universality, high in flexibility, higher in midpoint balancing speed, smaller in steady-state error and stronger in anti-interference capability.

Description

Multi-level converter midpoint potential rapid balance control system and method
Technical Field
The disclosure relates to the technical field of multi-level converters, and in particular relates to a system and a method for controlling rapid balancing of a midpoint potential of a multi-level converter.
Background
The statements in this section merely provide background information related to the present disclosure and may not necessarily constitute prior art.
Compared with the traditional three-phase two-level converter topology, the T-type three-level converter has the advantages of high efficiency, low output harmonic, small electromagnetic interference and the like; compared with a Neutral Point Clamped (NPC) type three-level converter, the T type three-level converter has the advantages of less device quantity (no need of diodes), lower switching loss, more uniform heat distribution, less number of independent driving power supplies and the like, and has become a hotspot of research and application. The method has a wide application prospect in the fields of electric vehicles or charging and discharging equipment for energy storage, new energy power conversion systems, motor dragging converter systems and the like.
However, due to the inconsistency of the capacitance characteristics of the switching device and the dc side and external disturbance, the dc deviation of the midpoint potential at the dc side of the T-type three-level converter, i.e. midpoint imbalance, is easy to occur. The harmonic content of the alternating current output voltage and current can be increased, the voltage stress of a power circuit switching device can be increased, the switching device can be damaged, and a direct current side capacitor can be broken down. Therefore, in order to ensure safe and reliable operation of the T-type three-level converter, it is necessary to ensure that the midpoint potential of the converter is half of the dc-side voltage.
The inventor of the present disclosure finds that, in order to effectively solve the problem of the unbalanced midpoint potential of the T-type three-level converter system, there are two solutions, namely hard solution and soft solution. The hardware scheme mainly comprises: (1) the upper capacitor voltage and the lower capacitor voltage are respectively and independently supplied by two independent direct current power supplies; (2) an additional current transformer is used to inject or draw current into the midpoint of the capacitor. Clearly, hardware solutions can increase system cost and complexity. The software scheme mainly comprises the following steps: (1) a midpoint potential balance control method based on a proportional P controller; (2) a neutral point potential balance control method based on a proportional-integral PI controller; (3) a neutral point potential balance control method based on a dead beat controller; (4) a midpoint potential balance control method based on model prediction control. However, the methods (1) and (2) have the disadvantages of slow dynamic response speed, large steady-state error, poor interference resistance and the like; the methods (3) and (4) have the problems of strong dependence on system hardware parameters, poor universality and the like due to system models.
Disclosure of Invention
In order to overcome the defects of the prior art, the invention provides a system and a method for controlling the rapid balance of the midpoint potential of the multi-level converter, which are independent of a system model, strong in universality and high in flexibility, and have the advantages of high midpoint balance speed, small steady-state error and strong anti-interference capability compared with the traditional proportional P controller and proportional-integral PI controller.
In order to achieve the purpose, the following technical scheme is adopted in the disclosure:
the first aspect of the disclosure provides a point potential rapid balance control system in a multi-level converter.
A multi-level converter midpoint potential rapid balance control system comprises an FTC controller, a common mode component calculation module and an SVPWM modulation module;
coordinate transformation is carried out on a d-axis modulation signal and a q-axis modulation signal of the multi-level converter to respectively obtain alternating current modulation signals under an abc coordinate system;
the common-mode component calculation module outputs a common-mode component signal according to the received alternating current modulation signal under the abc coordinate system, and the common-mode component signal is respectively superposed with the alternating current modulation signal under the abc coordinate system to respectively obtain a first modulation signal under the abc coordinate system;
acquiring voltage values of a first capacitor and a second capacitor on the direct current side of the multilevel converter, outputting an adjusting quantity by the FTC controller according to a difference value of the first capacitor and the second capacitor, and obtaining a second modulating signal under an abc coordinate system after the adjusting quantity is respectively superposed with a first modulating signal under the abc coordinate system;
and the SVPWM (Space Vector Pulse Width Modulation) Modulation module outputs a driving signal of the T-type multilevel converter according to a second Modulation signal and a triangular carrier wave under the abc coordinate system.
The second aspect of the disclosure provides a method for controlling rapid balancing of a midpoint potential in a multilevel converter.
A method for controlling rapid balance of a midpoint potential of a multilevel converter comprises the following steps:
coordinate transformation is carried out on a d-axis modulation signal and a q-axis modulation signal of the multi-level converter to respectively obtain alternating current modulation signals under an abc coordinate system;
obtaining a common-mode component signal according to the received alternating current modulation signal under the abc coordinate system, and respectively obtaining a first modulation signal under the abc coordinate system after the common-mode component signal is respectively superposed with the alternating current modulation signal under the abc coordinate system;
acquiring voltage values of a first capacitor and a second capacitor on a direct current side of the multilevel converter, and outputting adjustment quantities according to difference values of the first capacitor and the second capacitor, wherein the adjustment quantities are respectively superposed with a first modulation signal under an abc coordinate system to obtain a second modulation signal under the abc coordinate system;
and obtaining driving signals of switching tubes of a phase a, a phase b and a phase c of the T-type multi-level converter according to the second modulation signal and the triangular carrier wave in the abc coordinate system.
The third aspect of the present disclosure provides an electronic device including the system for controlling rapid balancing of a midpoint potential in a multilevel converter according to the first aspect of the present disclosure.
Compared with the prior art, the beneficial effect of this disclosure is:
1. the system and the method for controlling the rapid balance of the midpoint potential of the multi-level converter do not depend on a system model, get rid of the serious dependence on system hardware parameters, and have strong universality and high flexibility.
2. Compared with the traditional proportional P controller and the proportional-integral PI controller, the multi-level converter midpoint potential rapid balance control system and method have the advantages of rapid midpoint balance speed, small steady-state error and strong anti-interference capability.
3. According to the system and the method for controlling the rapid balance of the midpoint potential of the multilevel converter, the action time of P/N type redundancy small vectors is adjusted in real time by constructing an upper capacitor voltage closed-loop control system and a lower capacitor voltage closed-loop control system based on a finite time controller, the rapid balance control of the midpoint potential is realized, the voltage stress borne by a direct current side capacitor and a power circuit switch tube is reduced, the harmonic content of voltage and current at an alternating current side is reduced, and the harmonic pollution to a power grid is reduced.
4. Compared with the existing hardware scheme, the system and the method for controlling the rapid balance of the midpoint potential of the multi-level converter have the advantages of low cost, small system volume and high efficiency.
Drawings
Fig. 1 is a power topology diagram of a T-type three-level converter system provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 2 is a space voltage vector diagram of a T-type three-level converter system provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 3 is a block diagram of a point potential rapid balance control strategy in an FTC-based T-type three-level converter provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 4 is a schematic diagram of three-level PWM modulation under Up > Un working conditions provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 5 is a schematic diagram of three-level PWM modulation under an Up < Un condition provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 6 is a diagram illustrating a simulation effect of the proportional P controller-based midpoint balance control provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 7 is a diagram of simulation effect of the proportional-integral PI controller-based midpoint balance control provided in embodiment 1 or embodiment 2 of the present disclosure.
Fig. 8 is a diagram of a midpoint balance control simulation effect based on the method according to the present disclosure provided in embodiment 1 or embodiment 2 of the present disclosure.
Detailed Description
It should be noted that the following detailed description is exemplary and is intended to provide further explanation of the disclosure. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs.
It is noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of example embodiments according to the present disclosure. As used herein, the singular forms "a", "an" and "the" are intended to include the plural forms as well, and it should be understood that when the terms "comprises" and/or "comprising" are used in this specification, they specify the presence of stated features, steps, operations, devices, components, and/or combinations thereof, unless the context clearly indicates otherwise.
The embodiments and features of the embodiments in the present application may be combined with each other without conflict.
Example 1:
the embodiment 1 of the present disclosure provides a multi-level converter midpoint potential fast balance control system, including an FTC controller, a common mode component calculation module, and an SVPWM (Space Vector Pulse Width Modulation) Modulation module;
d-axis modulation signal m for multi-level converterdAnd q-axis modulation signal mqAfter coordinate transformation, alternating current modulation signals m under the abc coordinate system are respectively obtaineda，mb，mc；
The common-mode component calculation module outputs a common-mode component signal according to the received alternating current modulation signal under the abc coordinate system, and the common-mode component signal is respectively superposed with the alternating current modulation signal under the abc coordinate system to respectively obtain a first modulation signal m under the abc coordinate systema1，mb1，mc1；
Acquiring voltage values of a first capacitor and a second capacitor on a direct current side of a multilevel converter, outputting an adjustment quantity by an FTC (fiber Time Controller) according to a difference value of the first capacitor and the second capacitor, and respectively superposing the adjustment quantity with a first modulation signal m under an abc coordinate system to obtain a second modulation signal m under the abc coordinate systema2，mb2，mc2；
The SVPWM modulation module modulates the signal m according to a second modulation signal m under an abc coordinate systema2，mb2，mc2And a triangular carrier outputting a driving signal S of the T-type three-level convertera，Sb，Sc。
The following is further described with reference to the accompanying drawings:
the present embodiment is primarily directed to a T-type three-level converter system. As shown in fig. 1, the system comprises, from left to right: three-phase AC network eabcLCL filter, T-type three-phase full bridge, DC bus upper and lower side capacitor C1And C2DC side load RLDC side voltage source Vdc. Wherein, the T-shaped three-phase full bridge comprises A, B, C three phases, each phase comprises 4 power circuit switch tubes S1、S2、S3、S4. In addition, the T-type three-level converter system can operate in two working modes of rectification and inversion, wherein when the switch S is switched on1When closed, the system can operate in a commutation mode, when switch S is closed2When closed, the system may operate in an inverter mode.
Specifically, each phase bridge arm of the T-type three-level converter system comprises P, O, N three working states, and an upper capacitor C and a lower capacitor C on a direct current side are taken1And C2The midpoint of (b) is a reference potential point O, the output voltages of the bridge arms corresponding to P, O, N in three states are respectively: 1/2Udc, 0, -1/2 Udc.
Specifically, the three-phase output voltage of the T-type three-level converter system has 27 level combinations, that is, 27 voltage vectors, each including: 3 zero vectors (OOO, NNN, PPP), 12 small vectors (ONN, OON, NON, NOO, NNO, ONO; POO, PPO, OPO, OPP, OOP, POP), 6 medium vectors (PON, OPN, NPO, NOP, ONP, PNO) and 6 large vectors (PNN, PPN, NPN, NPP, NNP, PNP), wherein redundant vectors exist for both the zero and small vectors. The spatial distribution of the voltage vectors and their redundancy relationship are shown in fig. 2.
Based on the foregoing analysis, the basic steps of the FTC-based midpoint potential fast balance control method proposed in this embodiment are given based on the T-type three-level converter system shown in fig. 1:
(1) real-time acquisition of upper and lower capacitor voltages of a DC bus of the converter is carried out by utilizing a Hall voltage sensor, and instantaneous analog quantity of the capacitor voltages is transmitted to a DSP controller to complete AD conversion, wherein an upper capacitor C1Voltage value of is noted as UpA lower capacitor C2Voltage value of is noted as Un。
(2) Upper and lower capacitors C1、C2Voltage U ofpAnd UnCalculating the difference, and recording the difference as delta UpnNamely: delta Upn＝Up-UnThen, delta U is addedpnThe input quantity of the finite time control algorithm is input into a finite time controller, wherein the finite time controller is represented by the formula (1):
y＝k1*sign(e(t))*|e(t)|α (1)
in the formula (1), e (t) is the input quantity of the finite time controller, namely, Delta Upn(ii) a y is the output quantity of the controller, and the y refers to the regulating quantity of the P/N type redundancy small vector action time with stronger adjustment capacity on the center potential; k is a radical of1And alphaIs a controller parameter, where k1>0,0<α<And 1, setting specific values of the parameters according to a control target of a control system.
(3) Performing a finite time control method to reference the voltage vector VrefThe 6 th small sector located in the IIth large sector is taken as an example and discussed in the following two cases:
in the first case: u shapep>Un
When the upper capacitor voltage UpGreater than the lower capacitor voltage UnTime, difference DeltaU between upper and lower capacitor voltagespnInput to the finite time controller as input e (t) to the finite time controller, namely: e (t) ═ Δ UpnAt this time, the value of e (t) is greater than 0, and then the function relation (1) of the finite time controller is executed to calculate the output value y of the finite time controllerFTCNamely:
yFTC＝k1*sign(ΔUpn)*|ΔUpn|α (2)
wherein, yFTCRefers to the adjustment amount of P/N type redundancy small vector action time with stronger adjustment capability to the center potential, at the moment, yFTCIs also greater than 0, and then zero sequence component y is addedFTCAre respectively superposed on the modulated waves m shown in FIG. 3a1、mb1、mc1To form a final T-type three-level converter modulation wave ma2、mb2、mc2As shown in formula (3):
Figure BDA0002468972370000081
from the three-level PWM modulation scheme given in FIG. 4, the final modulated wave ma2、mb2、mc2Equivalent to modulating wave ma1、mb1、mc1Is moved up on the basis of | yFTC|(yFTCAbsolute value of (d) of the carrier, it is apparent that, in the 7-segment switching timing generated after comparison with the upper and lower carriers, the leftmost and rightmost 2-segment N-type small vector (OON) action time will be reduced, and thus the P-type small vector action time will be reducedAnd is increased. Wherein the action time of the N-type small vector is from tN1Is reduced to tN2Reduction of action time by Δ tN(ii) a The action time of P type small vector is from tP1Increase to tP2Increase in action time Δ tP。
And according to the inherent influence rule of the P/N type small vectors on the upper and lower capacitor voltages (the upper capacitor voltage U is reduced when the P type small vectors actpLowering low capacitor voltage U under action of N-type small vectorn) It is obvious that the effect of increasing the P-type small vector action time and decreasing the N-type small vector action time is: upper capacitor C1Voltage U acrosspWill reduce the lower capacitance C2Voltage U acrossnWill increase, therefore, for Up>UnFor the working condition of (1), the voltages of the upper capacitor and the lower capacitor of the T-type three-level converter system finally tend to be balanced, namely: u shapep＝Un。
In the second case: u shapep<Un
When the upper capacitor voltage UpLess than the lower capacitor voltage UnTime, difference DeltaU between upper and lower capacitor voltagespnInput to the finite time controller as input e (t) to the finite time controller, namely: e (t) ═ Δ UpnAt this time, the value of e (t) is less than 0, and then the function relation (1) of the finite time controller is executed to calculate the output value y of the finite time controllerFTCAs shown in formula (2). Wherein, yFTCRefers to the adjustment amount of P/N type redundancy small vector action time with stronger adjustment capability to the center potential, at the moment, yFTCIs also less than 0, and then zero sequence component y is addedFTCAre respectively superposed on the modulated waves m shown in FIG. 3a1、mb1、mc1To form a final T-type three-level converter modulation wave ma2、mb2、mc2As shown in formula (3).
From the three-level PWM modulation scheme given in FIG. 5, the final modulation waveform ma2、mb2、mc2Equivalent to modulating wave ma1、mb1、mc1Based on up-down moving | yFTC|(yFTCAbsolute value of (d) of the P-type small vectors, it is apparent that the leftmost and rightmost 2-segment N-type small vectors (OON) have increased and thus decreased on-times in the 7-segment switching sequences generated after comparison with the upper and lower carriers. Wherein the action time of the N-type small vector is from tN1Increase to tN2Increase in action time Δ tN(ii) a The action time of P type small vector is from tP1Is reduced to tP2Reduction of action time by Δ tP。
And according to the inherent influence rule of the P/N type small vectors on the upper and lower capacitor voltages (the upper capacitor voltage U is reduced when the P type small vectors actpLowering low capacitor voltage U under action of N-type small vectorn) It is obvious that the effect of the decrease of the P-type small vector action time and the increase of the N-type small vector action time is as follows: voltage U across upper capacitor C1pWill increase the voltage U across the lower capacitor C2nWill be reduced, therefore, for Up<UnFor the working condition of (1), the voltages of the upper capacitor and the lower capacitor of the T-type three-level converter system finally tend to be balanced, namely: u shapep＝Un。
(4) Based on the superposed zero-sequence component y given in the step (3)FTCRear modulated wave ma2、mb2、mc2And obtaining driving signals Sa, Sb and Sc of the T-type three-level converter system by assisting a double-carrier PWM (pulse-Width modulation) strategy. Wherein Sa is a driving signal of a-phase switching tube of the T-type three-level converter, Sb is a driving signal of b-phase switching tube of the T-type three-level converter, and Sc is a driving signal of c-phase switching tube of the T-type three-level converter.
In summary, a point potential rapid balance control strategy block diagram of the FTC-based T-type three-level converter is provided, as shown in fig. 3. Wherein id and iq are given artificially, wherein id is given according to the system power level, iq is 0, id and iq are dq axis components of alternating-current side grid-connected currents ia, ib and ic, and ia, ib and ic are obtained through a coordinate transformation module (abc/dq).
The input and output of the common mode component calculation module are respectively: m isa、mb、mcAnd vcomAnd satisfies formula (4):
vcom＝-0.5(vmax+vmin) (4)
in the formula, vmaxAnd vminThe following relationship is satisfied:
Figure BDA0002468972370000101
(5) simulation verification of the method disclosed by the invention is carried out through MATLAB-Simulink 2013b simulation software, and simulation parameters are shown in Table 1. The simulation results are shown in fig. 6, 7, and 8.
TABLE 1 simulation parameters Table
Grid voltage eabc(effective value) 110V
Voltage V at DC sidedc 350V
DC side capacitor C1、C2 2000uF
Filter inductance L 3mH
Switching frequency fs 10KHz
Simulation step length 2.5us
Wherein, fig. 6 is the simulation result of the midpoint balance control based on the proportional P controller, and it can be known that the dynamic response time of the system is about 0.05s, and the steady-state error is about 1.2V; fig. 7 is a simulation result of the midpoint balance control based on the proportional-integral PI controller, and it can be seen from the graph that the dynamic response time of the system is about 0.05s, the steady-state error is about 1.1V, and the control effect is substantially consistent with that of the proportional P controller.
Fig. 8 shows the simulation result of the midpoint balance control based on the method of the present embodiment, and it can be seen from the figure that the dynamic response time of the system is about 0.008s, and the steady-state error is about 0.2V. Obviously, compared with the proportional P and proportional-integral PI control methods, the midpoint potential fast balance control method provided by the embodiment greatly improves the dynamic response speed of the system and simultaneously significantly reduces the steady-state error of the system.
Example 2:
the embodiment 2 of the disclosure provides a method for controlling the rapid balance of a midpoint potential of a multi-level converter, which comprises an FTC controller, a common-mode component calculation module and an SVPWM modulation module;
acquiring an alternating current modulation signal under an abc coordinate system of a multilevel converter;
the common-mode component calculation module outputs a common-mode component signal according to the received alternating current modulation signal under the abc coordinate system, and the common-mode component signal is respectively superposed with the alternating current modulation signal under the abc coordinate system to respectively obtain a first modulation signal under the abc coordinate system;
acquiring voltage values of a first capacitor and a second capacitor on the direct current side of the multilevel converter, outputting an adjusting quantity by the FTC controller according to a difference value of the first capacitor and the second capacitor, and obtaining a second modulating signal under an abc coordinate system after the adjusting quantity is respectively superposed with a first modulating signal under the abc coordinate system;
and the SVPWM module respectively outputs driving signals of switching tubes of a phase a, a phase b and a phase c of the T-type three-level converter according to a second modulation signal and a triangular carrier wave in an abc coordinate system.
The following is further described with reference to the accompanying drawings:
the present embodiment is primarily directed to a T-type three-level converter system. As shown in FIG. 1, the systemThe system comprises the following components in sequence from left to right: three-phase AC network eabcLCL filter, T-type three-phase full bridge, DC bus upper and lower side capacitor C1And C2DC side load RLDC side voltage source Vdc. Wherein, the T-shaped three-phase full bridge comprises A, B, C three phases, each phase comprises 4 power circuit switch tubes S1、S2、S3、S4. In addition, the T-type three-level converter system can operate in two working modes of rectification and inversion, wherein when the switch S is switched on1When closed, the system can operate in a commutation mode, when switch S is closed2When closed, the system may operate in an inverter mode.
Specifically, each phase bridge arm of the T-type three-level converter system comprises P, O, N three working states, and an upper capacitor C and a lower capacitor C on a direct current side are taken1And C2The midpoint of (b) is a reference potential point O, the output voltages of the bridge arms corresponding to P, O, N in three states are respectively: 1/2Udc, 0, -1/2 Udc.
Specifically, the three-phase output voltage of the T-type three-level converter system has 27 level combinations, that is, 27 voltage vectors, each including: 3 zero vectors (OOO, NNN, PPP), 12 small vectors (ONN, OON, NON, NOO, NNO, ONO; POO, PPO, OPO, OPP, OOP, POP), 6 medium vectors (PON, OPN, NPO, NOP, ONP, PNO) and 6 large vectors (PNN, PPN, NPN, NPP, NNP, PNP), wherein redundant vectors exist for both the zero and small vectors. The spatial distribution of the voltage vectors and their redundancy relationship are shown in fig. 2.
Based on the foregoing analysis, the basic steps of the FTC-based midpoint potential fast balance control method proposed in this embodiment are given based on the T-type three-level converter system shown in fig. 1:
(1) real-time acquisition of upper and lower capacitor voltages of a DC bus of the converter is carried out by utilizing a Hall voltage sensor, and instantaneous analog quantity of the capacitor voltages is transmitted to a DSP controller to complete AD conversion, wherein an upper capacitor C1Voltage value of is noted as UpA lower capacitor C2Voltage value of is noted as Un。
(2) Upper and lower capacitors C1、C2Voltage U ofpAnd UnCalculating the difference, and recording the difference as delta UpnNamely: delta Upn＝Up-UnThen, delta U is addedpnThe input quantity of the finite time control algorithm is input into a finite time controller, wherein the finite time controller is represented by the formula (1):
y＝k1*sign(e(t))*|e(t)|α (1)
in the formula (1), e (t) is the input quantity of the finite time controller, namely, Delta Upn(ii) a y is the output quantity of the controller, and the y refers to the regulating quantity of the P/N type redundancy small vector action time with stronger adjustment capacity on the center potential; k is a radical of1And α is a controller parameter, where k1>0,0<α<And 1, setting specific values of the parameters according to a control target of a control system.
(3) Performing a finite time control method to reference the voltage vector VrefThe 6 th small sector located in the IIth large sector is taken as an example and discussed in the following two cases:
in the first case: u shapep>Un
When the upper capacitor voltage UpGreater than the lower capacitor voltage UnTime, difference DeltaU between upper and lower capacitor voltagespnInput to the finite time controller as input e (t) to the finite time controller, namely: e (t) ═ Δ UpnAt this time, the value of e (t) is greater than 0, and then the function relation (1) of the finite time controller is executed to calculate the output value y of the finite time controllerFTCNamely:
yFTC＝k1*sign(ΔUpn)*|ΔUpn|α (2)
wherein, yFTCRefers to the adjustment amount of P/N type redundancy small vector action time with stronger adjustment capability to the center potential, at the moment, yFTCIs also greater than 0, and then zero sequence component y is addedFTCAre respectively superposed on the modulated waves m shown in FIG. 3a1、mb1、mc1To form a final T-type three-level converter modulation wave ma2、mb2、mc2As shown in formula (3)
Figure BDA0002468972370000131
From the three-level PWM modulation scheme given in FIG. 4, the final modulated wave ma2、mb2、mc2Equivalent to modulating wave ma1、mb1、mc1Is moved up on the basis of | yFTC|(yFTCAbsolute value of (d) of the P-type small vectors, it is apparent that the leftmost and rightmost 2-segment N-type small vectors (OON) have a reduced action time and thus an increased action time in the 7-segment switching timing generated after comparison with the upper and lower carriers. Wherein the action time of the N-type small vector is from tN1Is reduced to tN2Reduction of action time by Δ tN(ii) a The action time of P type small vector is from tP1Increase to tP2Increase in action time Δ tP。
And according to the inherent influence rule of the P/N type small vectors on the upper and lower capacitor voltages (the upper capacitor voltage U is reduced when the P type small vectors actpLowering low capacitor voltage U under action of N-type small vectorn) It is obvious that the effect of increasing the P-type small vector action time and decreasing the N-type small vector action time is: upper capacitor C1Voltage U acrosspWill reduce the lower capacitance C2Voltage U acrossnWill increase, therefore, for Up>UnFor the working condition of (1), the voltages of the upper capacitor and the lower capacitor of the T-type three-level converter system finally tend to be balanced, namely: u shapep＝Un。
In the second case: u shapep<Un
When the upper capacitor voltage UpLess than the lower capacitor voltage UnTime, difference DeltaU between upper and lower capacitor voltagespnInput to the finite time controller as input e (t) to the finite time controller, namely: e (t) ═ Δ UpnAt this time, the value of e (t) is less than 0, and then the function relation (1) of the finite time controller is executed to calculate the output value y of the finite time controllerFTCAs shown in formula (2). Wherein, yFTCRefers to the adjustment of the potential of the center pointThe adjustment amount of the action time of the P/N type redundant small vector with strong node capacity, at this time, yFTCIs also less than 0, and then zero sequence component y is addedFTCAre respectively superposed on the modulated waves m shown in FIG. 3a1、mb1、mc1To form a final T-type three-level converter modulation wave ma2、mb2、mc2As shown in formula (3).
From the three-level PWM modulation scheme given in FIG. 5, the final modulation waveform ma2、mb2、mc2Equivalent to modulating wave ma1、mb1、mc1Based on up-down moving | yFTC|(yFTCAbsolute value of (d) of the P-type small vectors, it is apparent that the leftmost and rightmost 2-segment N-type small vectors (OON) have increased and thus decreased on-times in the 7-segment switching sequences generated after comparison with the upper and lower carriers. Wherein the action time of the N-type small vector is from tN1Increase to tN2Increase in action time Δ tN(ii) a The action time of P type small vector is from tP1Is reduced to tP2Reduction of action time by Δ tP。
And according to the inherent influence rule of the P/N type small vectors on the upper and lower capacitor voltages (the upper capacitor voltage U is reduced when the P type small vectors actpLowering low capacitor voltage U under action of N-type small vectorn) It is obvious that the effect of the decrease of the P-type small vector action time and the increase of the N-type small vector action time is as follows: voltage U across upper capacitor C1pWill increase the voltage U across the lower capacitor C2nWill be reduced, therefore, for Up<UnFor the working condition of (1), the voltages of the upper capacitor and the lower capacitor of the T-type three-level converter system finally tend to be balanced, namely: u shapep＝Un。
(4) Based on the superposed zero-sequence component y given in the step (3)FTCRear modulated wave ma2、mb2、mc2And obtaining driving signals Sa, Sb and Sc of the T-type three-level converter system by assisting a double-carrier PWM (pulse-Width modulation) strategy. Wherein Sa is a driving signal of a-phase switching tube of the T-type three-level converter, and Sb is a driving signal of b-phase switching tube of the T-type three-level converterAnd Sc is a driving signal of a c-phase switching tube of the T-type three-level converter.
In summary, a point potential rapid balance control strategy block diagram of the FTC-based T-type three-level converter is provided, as shown in fig. 3. Wherein id and iq are given artificially, wherein id is given according to the system power level, iq is 0, id and iq are dq axis components of alternating-current side grid-connected currents ia, ib and ic, and ia, ib and ic are obtained through a coordinate transformation module (abc/dq).
The input and output of the common mode component calculation module are respectively: m isa、mb、mcAnd vcomAnd satisfies formula (4):
vcom＝-0.5(vmax+vmin) (4)
in the formula, vmaxAnd vminThe following relationship is satisfied:
Figure BDA0002468972370000151
(5) simulation verification of the method disclosed by the invention is carried out through MATLAB-Simulink 2013b simulation software, and simulation parameters are shown in Table 1. The simulation results are shown in fig. 6, 7, and 8.
TABLE 1 simulation parameters Table
Grid voltage eabc(effective value) 110V
Voltage V at DC sidedc 350V
DC side capacitor C1、C2 2000uF
Filter inductance L 3mH
Switching frequency fs 10KHz
Simulation step length 2.5us
Wherein, fig. 6 is the simulation result of the midpoint balance control based on the proportional P controller, and it can be known that the dynamic response time of the system is about 0.05s, and the steady-state error is about 1.2V; fig. 7 is a simulation result of the midpoint balance control based on the proportional-integral PI controller, and it can be seen from the graph that the dynamic response time of the system is about 0.05s, the steady-state error is about 1.1V, and the control effect is substantially consistent with that of the proportional P controller.
Fig. 8 shows the simulation result of the midpoint balance control based on the method of the present embodiment, and it can be seen from the figure that the dynamic response time of the system is about 0.008s, and the steady-state error is about 0.2V. Obviously, compared with the proportional P and proportional-integral PI control methods, the midpoint potential fast balance control method provided by the embodiment greatly improves the dynamic response speed of the system and simultaneously significantly reduces the steady-state error of the system.
Example 3:
the embodiment 3 of the present disclosure provides an electronic device, including the system for controlling rapid balancing of midpoint potentials in a multilevel converter according to the embodiment 1 of the present disclosure.
The above description is only a preferred embodiment of the present disclosure and is not intended to limit the present disclosure, and various modifications and changes may be made to the present disclosure by those skilled in the art. Any modification, equivalent replacement, improvement and the like made within the spirit and principle of the present disclosure should be included in the protection scope of the present disclosure.

Claims (6)

1. A multi-level converter midpoint potential rapid balance control system is characterized by comprising an FTC controller, a common mode component calculation module and an SVPWM modulation module;
coordinate transformation is carried out on a d-axis modulation signal and a q-axis modulation signal of the multi-level converter to respectively obtain alternating current modulation signals under an abc coordinate system;
the common-mode component calculation module outputs a common-mode component signal according to the received alternating current modulation signal under the abc coordinate system, and the common-mode component signal is respectively superposed with the alternating current modulation signal under the abc coordinate system to respectively obtain a first modulation signal under the abc coordinate system;
acquiring voltage values of a first capacitor and a second capacitor on the direct current side of the multilevel converter, outputting an adjusting quantity by the FTC controller according to a difference value of the first capacitor and the second capacitor, and obtaining a second modulating signal under an abc coordinate system after the adjusting quantity is respectively superposed with a first modulating signal under the abc coordinate system;
the FTC controller specifically comprises:
yFTC＝k1*sign(ΔUpn)*|ΔUpn|α
wherein, yFTCIs the output quantity of the controller; k is a radical of1And α is a controller parameter, k1>0,0<α<1；ΔUpnThe voltage difference value of a first capacitor and a second capacitor on the direct current side of the multilevel converter is obtained;
the second modulation signal in the abc coordinate system is specifically:
Figure FDA0002923622620000011
wherein, yFTCAn output quantity of the FTC controller, in particular an adjustment quantity of P/N type redundancy small vector action time for adjusting the midpoint potential, ma1，mb1，mc1Respectively a first modulation signal under an abc coordinate system;
and the SVPWM module outputs a driving signal of the T-shaped multi-level converter according to the second modulation signal and the triangular carrier wave in the abc coordinate system.
2. The system for controlling rapid balancing of a midpoint potential in a multilevel converter according to claim 1, wherein the common mode component signal is specifically:
vcom＝-0.5(vmax+vmin)
wherein v ismaxIs the maximum value, v, of the AC modulated signal in the abc coordinate systemminIs the minimum value of the alternating current modulation signal under the abc coordinate system.
3. The system for controlling rapid balancing of midpoint potentials of multilevel converters according to claim 1, wherein the SVPWM modulation module outputs driving signals of switching tubes of a phase, b phase and c phase of the T-type multilevel converter respectively.
4. A method for controlling the rapid balance of the point potential in a multilevel converter is characterized by comprising the following steps:
coordinate transformation is carried out on a d-axis modulation signal and a q-axis modulation signal of the multi-level converter to respectively obtain alternating current modulation signals under an abc coordinate system;
obtaining a common-mode component signal according to the received alternating current modulation signal under the abc coordinate system, and respectively obtaining a first modulation signal under the abc coordinate system after the common-mode component signal is respectively superposed with the alternating current modulation signal under the abc coordinate system;
acquiring voltage values of a first capacitor and a second capacitor on a direct current side of the multilevel converter, and outputting adjustment quantities according to difference values of the first capacitor and the second capacitor, wherein the adjustment quantities are respectively superposed with a first modulation signal under an abc coordinate system to obtain a second modulation signal under the abc coordinate system;
the method comprises the steps of obtaining voltage values of a first capacitor and a second capacitor on a direct current side of the multilevel converter, and outputting an adjustment quantity according to a difference value of the first capacitor and the second capacitor, and specifically comprises the following steps:
yFTC＝k1*sign(ΔUpn)*|ΔUpn|α
wherein, yFTCIs the output quantity of the controller; k is a radical of1And α is a controller parameter, k1>0,0<α<1；ΔUpnThe voltage difference value of a first capacitor and a second capacitor on the direct current side of the multilevel converter is obtained;
the second modulation signal in the abc coordinate system is specifically:
Figure FDA0002923622620000031
wherein, yFTCAn output quantity of the FTC controller, in particular an adjustment quantity of P/N type redundancy small vector action time for adjusting the midpoint potential, ma1，mb1，mc1Respectively a first modulation signal under an abc coordinate system;
and obtaining driving signals of switching tubes of a phase a, a phase b and a phase c of the T-type multi-level converter according to the second modulation signal and the triangular carrier wave in the abc coordinate system.
5. The method for controlling rapid balancing of a midpoint potential in a multilevel converter according to claim 4, wherein the common mode component signal is specifically:
vcom＝-0.5(vmax+vmin)
wherein v ismaxIs the maximum value, v, of the AC modulated signal in the abc coordinate systemminIs the minimum value of the alternating current modulation signal under the abc coordinate system.
6. An electronic device comprising the system for rapid balancing of midpoint potentials in a multilevel converter according to any of claims 1 to 3.
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