CN111277186A - Induction motor field weakening control method based on optimized six-beat operation - Google Patents

Induction motor field weakening control method based on optimized six-beat operation Download PDF

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CN111277186A
CN111277186A CN201811481087.3A CN201811481087A CN111277186A CN 111277186 A CN111277186 A CN 111277186A CN 201811481087 A CN201811481087 A CN 201811481087A CN 111277186 A CN111277186 A CN 111277186A
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current
voltage
omega
weak magnetic
induction motor
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CN111277186B (en
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于泳
张旭
王勃
张静
徐殿国
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Harbin Institute of Technology
Delta Greentech China Co Ltd
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Harbin Institute of Technology
Delta Greentech China Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/20—Estimation of torque
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/085—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency

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  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

The invention provides an induction motor field weakening control method based on six-beat operation optimization, and belongs to the technical field of motor control. The control method firstly sets the given voltage value in the flux-weakening controller to be 2UdcThe frequency of the feedback current resonance filter is set to be omega* cAnd 6. omegasThe greater between; when the rotating speed exceeds the rated rotating speed, the motor enters a weak magnetic region, the inverter outputs six-beat voltage, the six-beat voltage causes the generation of the current sixth harmonic, and the feedback current with the sixth harmonic passes throughAfter passing through the resonant filter, making a difference with a given value, entering a complex vector current regulator, generating a given voltage by the current regulator, and generating an inverter trigger pulse through an SVPWM (space vector pulse width modulation) strategy to control the motor to operate in a weak magnetic region. The invention is applied to the six-beat operation flux weakening control of the induction motor.

Description

Induction motor field weakening control method based on optimized six-beat operation
Technical Field
The invention relates to an induction motor field weakening control method based on six-beat operation optimization, and belongs to the technical field of motor control.
Background
With the wide application and continuous development of vector control in the field of speed regulation systems of alternating-current motors, the driving capability, dynamic response, speed regulation range, precision and other aspects of the alternating-current motors are remarkably improved, and therefore the vector control system is gradually used for replacing a direct-current speed regulation system to be applied to various industrial application occasions including household appliances, elevator traction, automobile machine tools and the like. Meanwhile, the requirement of high precision, high efficiency and high quality in the fields of electric automobiles, numerical control machine tool spindle drives, electric locomotives and the like particularly in high-performance industrial control occasions prompts the development of motor drive variable frequency speed control systems towards high speed. In the field of electric vehicles, in the face of complex road traffic conditions, the drive of electric vehicles needs to have the ability to operate under a variety of operating conditions. The device can realize smooth high-speed running in a medium/high-speed area so as to meet the requirement of people on comfort and safety. In the main shaft driving system for the numerical control machine tool, as a core component, the quality of the main shaft driving control performance directly determines the overall level of the numerical control machine tool. The requirements of high efficiency, high precision and high reliability can be met by adopting a direct drive mode, so that the driving motor gradually develops towards high speed. Similarly, in high power transmission applications such as electric locomotives, the high speed of train operation requires stable operation of the traction transmission system of the electric locomotive at high rotational speeds, and also requires the ability to consistently output maximum torque to overcome the large resistance of the train during high speed operation.
The requirements for motor speed regulation in high-performance control occasions include: wide speed regulating range (especially high speed region), high speed stable running capability, maximum torque output capability, full utilization of limited voltage and current resources, high dynamic response and the like.
However, the conventional ac motor has two problems in high-speed driving:
1. traditionally, only the output amplitude of three-phase voltage can reach the voltage of a direct current bus by adopting an SVPWM (space vector pulse width modulation) method
Figure BDA0001893398970000011
And the maximum fundamental wave voltage amplitude value which can be output by the SVPWM module is 2/pi of the direct current bus voltage, which indicates that the waste of direct current bus voltage resources exists in the traditional method.
2. If the overmodulation technology is adopted to carry out limit expansion on the output voltage of the inverter, current distortion is inevitable, a filter has to be used for filtering, and the traditional filter can reduce the current bandwidth and influence the control performance of the whole system.
Disclosure of Invention
The invention provides an induction motor field weakening control method based on optimized six-beat operation, which aims to solve the problems that harmonic wave is increased and system performance is poor due to nonlinear expansion of motor stator voltage, and simultaneously aims at the problem that the traditional filter can reduce the current loop bandwidth.
The invention relates to an induction motor weak magnetic control method based on optimized six-beat operation, which is used for an induction motor system in a high-speed weak magnetic operation state, and adopts the following technical scheme:
an induction motor weak magnetic control method based on optimized six-beat operation, the control method comprises the following steps:
firstly, setting the given voltage value in the field weakening controller to be 2UdcThe/pi ensures that the system can reach a six-beat running mode, and a feedback current resonant filter is built based on a generalized second-order integrator;
secondly, filtering the feedback current with the sixth harmonic by using the feedback current resonant filter, and generating a command voltage by using a complex vector current regulator after the difference is made between the feedback current and a current set value;
step three: secondly, the instruction voltage generated by the complex vector current regulator enters an SVPWM module to generate an inverter trigger pulse on one hand, and on the other hand, the instruction voltage is used as a feedback signal of the flux weakening first-zone controller to complete flux weakening control according to a voltage constraint equation;
step four, according to the weak magnetic control process in step three, the exciting current component produced by the weak magnetic controller gives isd，refThen, a torque current component given i is generated according to the current constraintsq，ref(ii) a Wherein isd，refAnd isq，refRespectively representing the d-axis and q-axis given currents of the stator;
step five: as the rotating speed continues to rise, in order to meet the maximum slip limit, d-axis command voltage output by the complex vector current regulator is used as a flux weakening two-zone controller feedback signal, and dq-axis command voltage is met
Figure BDA0001893398970000021
u* sd、u* sqThe dq-axis command voltages are respectively applied until a given rotation speed is reached.
Further, the process of building the feedback current resonant filter based on the generalized second-order integrator in the step one comprises the following steps:
will resonate at a frequency omegahSet to ω* cAnd 6. omegaeThe larger value in between, namely: e.g. omega* cAnd 6. omegaeThe larger value therebetween is ω* cThen the resonant frequency ωhSet to ω* cE.g. omega* cAnd 6. omegaeThe larger value between is 6. omegaeThen the resonant frequency ωhIs set to be 6 omegaeWherein, ω is* cFor the design value of the current loop bandwidth, ωeIs an instantaneous synchronous angular frequency and is 6 · ω in the fundamental regione<ω* cFilter resonant frequency omegah＝ω* cThe motor stator voltage is a standard sine wave, and no sixth harmonic or harmonic exists in the feedback stator currentThe vibration filter does not function.
Further, the process of generating the command voltage in the second step is:
the detailed process of the control method comprises the following steps: when the induction motor enters a weak magnetic area along with the increase of the rotating speed, the weak magnetic control of the induction motor is started; according to the condition that the inverter outputs six-beat voltage in the weak magnetic region and generates a large amount of sixth harmonic in the stator current of the motor, the frequency of the sixth harmonic in the weak magnetic region is 6 omegae>ω* cSetting the resonant frequency omega of the resonant filterh＝6·ωe(ii) a And attenuating the sixth harmonic through a resonant filter to obtain a current signal only with a fundamental current component, inputting the fundamental current component into a current closed-loop controller, and entering a complex vector current regulator to generate an instruction voltage after making a difference with a current given value.
The resonant frequency of the resonant filter is selected in a mode that the filter does not influence the bandwidth of a current loop in a full-speed domain range; at the same time, under the action of resonant filter, the current regulator outputs command voltage u* sdq No 6 th harmonic exists in a synchronous rotating coordinate system, no 5 th harmonic and 7 th harmonic exist in a two-phase static coordinate system, and the control module (a current regulator and a voltage regulator) is free from harmonic interference while the output torque of a weak magnetic area is improved under six-beat operation.
The invention has the beneficial effects that:
the weak magnetic control method of the induction motor based on the optimized six-beat operation can realize the non-harmonic operation in the controller and keep the current bandwidth unaffected while the induction motor is in the weak magnetic area for six-beat operation, realizes the stable six-beat operation of the weak magnetic area of the induction motor, improves the maximum torque of the weak magnetic area and optimizes the dynamic and static performances. The weak magnetic control method of the induction motor based on the optimized six-beat operation provided by the invention realizes the improvement of the maximum torque under the condition that a control module is not influenced by effectively improving the weak magnetic controller and the current closed loop. And according to the characteristics of the current loop, a feedback current filter which does not influence the bandwidth of the current loop is designed, and the dynamic performance of the current loop is ensured.
Drawings
FIG. 1: a voltage closed loop-based weak magnetic control system block diagram of a traditional induction motor;
FIG. 2: the maximum torque of the induction motor outputs a voltage current track;
FIG. 3: overmodulating a first region stator voltage vector trajectory;
FIG. 4: overmodulating a corresponding graph of a stator phase voltage space vector locus and a time domain waveform of a region;
FIG. 5: overmodulating a stator voltage vector locus of a second region;
FIG. 6: overmodulation a corresponding graph of a space vector locus and a time domain waveform of a stator phase voltage of a second region;
FIG. 7: a weak magnetic control system block diagram of an induction motor based on optimized six-beat operation;
FIG. 8: a block diagram of a feedback current resonant filter based on a generalized second order integrator (SOGI);
FIG. 9: a feedback current resonant filter bode diagram;
FIG. 10: the experimental result of the comparison of the method and the traditional method in the three-fold base speed step acceleration process is shown, wherein (a) is an experimental result graph of the traditional inscribed circle method, and (b) is an experimental result graph of the method;
FIG. 11: the method outputs the torque condition under 1, 2, 3 and 4 times of basic speed;
FIG. 12: the steady state torque frequency spectrum of the method under the basic speed of 1, 2, 3 and 4 times, wherein, (a) is a steady state torque frequency spectrum graph under the basic speed of 1 time, (b) is a steady state torque frequency spectrum graph under the basic speed of 2 times, (c) is a steady state torque frequency spectrum graph under the basic speed of 3 times, and (d) is a steady state torque frequency spectrum graph under the basic speed of 4 times;
FIG. 13: the traditional method and the invented method have the conditions of rotating speed, torque current, phase current and phase current THD under the load of 0%, 20%, 40% and 60%, wherein, (a) is a waveform diagram of the traditional inscribed circle method, (b) is a waveform diagram of the method of the invention, and (c) is the total distortion rate of the phase current waveform in the method of the invention under different loads;
FIG. 14: feedback current and frequency spectrum before and after filtering, wherein, (a) is feedback current and frequency spectrum of a harmonic-free filter, and (b) is feedback current and frequency spectrum of the harmonic-free filter;
FIG. 15: the voltage time domain waveform under the two-phase static coordinate system is output by the current regulator without the feedback current filter, wherein, (a) is a voltage time domain waveform under the two-phase static coordinate system and the output voltage track of the current regulator without the feedback current filter, and (b) is a voltage time domain waveform under the two-phase static coordinate system and the output voltage track of the current regulator with the feedback filter.
Detailed Description
The embodiments of the present invention will be described in detail and fully with reference to the accompanying drawings in the patent embodiments, but the present invention is not limited by the embodiments, and the embodiments described below are only a part of the embodiments of the present invention, and not all of the embodiments. All other examples, which can be obtained by a person skilled in the art without making any creative effort based on the embodiments of the present invention, belong to the protection scope of the present invention.
Example 1:
the embodiments of the present invention will be described in detail and fully with reference to the accompanying drawings, which are included in the embodiments of the present invention. All other examples, which can be obtained by a person skilled in the art without making any creative effort based on the embodiments of the present invention, belong to the protection scope of the present invention.
Based on the traditional flux weakening control strategy based on the voltage closed loop, the invention firstly explains the flux weakening control strategy of the voltage closed loop with reference to fig. 1:
the operating range of the induction motor in the voltage closed loop flux weakening control strategy can be divided into three intervals: a constant torque area, a constant power area (a weak magnetic area I) and a constant voltage area (a weak magnetic area II). In the interval where the motor speed is lower than the rated speed, it is called a constant torque zone. In the constant torque region, the stator current is rated current ismaxVoltage and rotation speed are increased in fixed proportion, inducted electricityThe back electromotive force of the machine operation is smaller than the rated voltage value, and the rated torque is output and kept constant. When the voltage rises to the rated voltage usmaxWhen the induction motor reaches the rated rotating speed, the induction motor enters a constant power area. Due to the maximum voltage usmaxAnd limitation of the maximum current ismaxIn this interval, the rated power can be output. In the constant voltage region, the motor can not keep the maximum current due to the limitation of the maximum slip rate of the stable operation, and is simultaneously subjected to the maximum voltage usmaxOutput torque and power decrease sharply with increasing speed.
The mathematical model of the induction machine oriented by the rotor flux linkage is as follows:
Figure BDA0001893398970000041
in the formula: u. ofsd，usqIs a d-q shafting stator voltage component; i.e. isd，isqIs a d-q shafting stator current component; omegaeThe synchronous rotating speed of the electrical angle; rsIs a stator resistor; l iss，LrThe self-inductance of the stator and the rotor is obtained; l ismIs mutual inductance; sigma is a leakage inductance factor, and the leakage inductance factor,
Figure BDA0001893398970000042
ψris the rotor flux linkage.
During the operation of an induction machine, it is mainly constrained by the maximum voltage and the maximum current. The constraints are comprehensively determined by considering the self limitations of the motor and the inverter and the operating environment, such as the maximum output voltage allowed by the inverter, the maximum current allowed by the motor, the working time under different working conditions, the heat dissipation condition and the like. Therefore, the maximum torque control under the constraint can be described as:
maximum value:
Figure BDA0001893398970000051
constraint conditions are as follows:
Figure BDA0001893398970000052
if the stator resistance and the transient part in the formula (1) are omitted, the constraint condition of the weak magnetic area expressed in the form of voltage is obtained by combining the stator resistance and the transient part with the formula (3):
Figure BDA0001893398970000053
consider u to besmaxAnd ismaxAre set to a constant value, then as the speed of the induction machine increases, the ideal constrained trajectory curve is shown in fig. 2, where ABO represents the current trajectory and OCD represents the voltage trajectory. Thus the whole operating interval can be divided into three zones: the constant torque area corresponds to a point A and a line segment OC, the constant power area (a weak magnetic area I) corresponds to a curve segment AB and CD, and the constant voltage area (a weak magnetic area II) corresponds to a line segment BO and a point D. The maximum torque control in the flux weakening zone of the induction motor is to ensure that the current vector trajectory can run along ABO and the voltage vector trajectory can run along the OCD segment and finally stay at point D as the rotating speed increases.
FIG. 1 shows a voltage closed loop flux weakening control block diagram based on indirect rotor flux linkage orientation. On the basis of the commonly used double closed loop structure, two PI voltage closed loops are additionally used for forcing the voltage vector to operate according to the ideal maximum torque track shown in FIG. 2. Wherein the controller I is used for weakening d-axis current given, so the controller is also called a flux weakening controller, and the given is set to be
Figure BDA0001893398970000054
Namely the radius of an inscribed circle of the hexagonal space vector pulse width modulation maximum output voltage, the feedback value is the amplitude of a voltage vector output by a current loop, and the amplitude limiting value is set to be 0. Controller II is used for weakening q-axis current setting in constant voltage area, and the setting is set
Figure BDA0001893398970000055
(corresponding to point D in FIG. 2), the feedback value is the D-axis voltage amplitude, and the clipping value is 0.
After the motor is started, the exciting current is maintained at a given value to ensure proper magnetic field size, and the torque current passes through
Figure BDA0001893398970000056
Defined so as to guarantee a maximum current condition. In the stage, the motor is only limited by the maximum current, the voltage controller does not work, the output is 0, the rotating speed is increased rapidly, and the dq axis voltage | usd|usqThe rapid rise, the back emf quickly approaches the inverter maximum output voltage. From the torque equation (2), this stage can ensure a constant torque output capability, and is called a constant torque region.
When the back electromotive force rises to the weak magnetic controller (controller I) to start acting, the motor enters a constant power area. i.e. idFollowing id,refThe d-axis flux linkage is always kept in an ideal weak magnetic state, and the output voltage is always kept at the magnitude of the restraint voltage. At this time, the torque current iqTo be received
Figure BDA0001893398970000061
The effect is increased, and the maximum current state is still maintained. Knowing the speed omega from the equation of voltagerAnd torque current iqWhile increasing so that | usdI rapidly rises, uqIt is forced to start dropping. The whole constant power area motor is simultaneously subjected to the double constraints of maximum voltage and maximum current, and the flux weakening controller is always in action.
When usdL is raised to
Figure BDA0001893398970000062
The voltage controller II is activated, meaning that the motor enters the constant voltage region, and the decrease in q-axis current causes u to godIs maintained at
Figure BDA0001893398970000063
U and uqWill naturally adjust to
Figure BDA0001893398970000064
At this stage, the maximum current state cannot be satisfied, the motor is only constrained by the maximum voltage, and the two voltage controllers are always on.
Secondly, the invention is applied to a six-beat operation mode, and since the six-beat operation mode is the final state of the overmodulation operation, the overmodulation algorithm is explained as follows:
defining the modulation factor M as:
Figure BDA0001893398970000065
wherein
Figure BDA0001893398970000066
Given voltage vector modular length, 2U, for current regulator outputdcAnd/pi is the maximum fundamental voltage amplitude which can be output by the inverter.
The overmodulation algorithm adopts the principle that the amplitude of the fundamental wave of the stator voltage is equivalent, and is divided into an overmodulation first region and an overmodulation second region along with the increase of a modulation coefficient. Next, first, overmodulation region is described, as shown by the dashed line in FIG. 3, when u* sdqAfter the regular hexagon is exceeded, the trace of the voltage vector of the exceeding part is limited to the boundary of the regular hexagon, thereby inevitably causing the reduction of the amplitude of the fundamental wave of the stator voltage, and the amplitude of the voltage vector close to the vertex of the hexagon needs to be increased to compensate the loss, as shown by the solid line in fig. 3.
Overmodulation-one region modulation angle α is defined in FIG. threerThen, according to FIG. 4, a quarter period (0) can be obtained<θ<Pi/2) stator phase voltage instantaneous values u and αrThe following relationships exist:
Figure BDA0001893398970000071
where θ is the phase angle of the current regulator output voltage vector.
The fundamental component U of U can be obtained from equation (7)1And αrThe following relationships exist:
Figure BDA0001893398970000072
α can be obtained according to the principle of equivalent fundamental amplituderThe relationship with the modulation M is as follows:
Figure BDA0001893398970000073
with following
Figure BDA0001893398970000074
A further increase of αrAnd the amplitude of the fundamental wave of the stator voltage cannot be further increased by the method if the amplitude is continuously reduced from pi/6 to 0, and the amplitude of the fundamental wave of the stator voltage can only be further increased by increasing the dwell time of the voltage vector at the vertex of the hexagon to enter an overmodulation second region.
Defining a two-zone modulation angle α as shown in FIG. 5h
The overmodulation two region algorithm is illustrated by taking the voltage vector in sector one as an example. When a voltage vector is given, as shown in FIG. 5
Figure BDA0001893398970000075
When rotating from A to B, the actual voltage vector stays at the point A when
Figure BDA0001893398970000076
Rotating from B to C, the actual voltage vector rotates to point D at a faster speed
Figure BDA0001893398970000077
Rotating from C to D, the actual voltage vector stays at point D, thus increasing the voltage magnitude with a given voltage vector phase change. Modulated voltage vector phase angle theta0The relationship to a given voltage vector phase angle θ is as follows:
Figure BDA0001893398970000078
the overmodulation two-phase voltage instantaneous value u can likewise be obtained from fig. 62And αhThe relationship of (a) to (b) is as follows:
Figure BDA0001893398970000081
α can be obtainedhThe relationship with the modulation M is as follows:
Figure BDA0001893398970000082
when αhWhen the voltage vector of the stator jumps in six vertexes of the hexagon in sequence at pi/6, the inverter is called to be in a six-beat mode under the state, the amplitude of the fundamental wave of the output voltage also reaches the maximum value, and the state is the operation state of the inverter in the weak magnetic region.
Based on the above description of the voltage closed-loop theory and the overmodulation algorithm, the present invention is described in detail below:
the method comprises the following steps: referring to fig. 7, the reference voltage is set to 2UdcThe reference voltage and the feedback voltage are subjected to difference and then enter a PI regulator, and a stator current excitation component given value i is obtained through PI regulation and amplitude limitingsd,ref(ii) a Simultaneously, a speed measuring coded disc is adopted to obtain the rotating speed omega of the rotorrObtaining a stator current torque component given value i through PI after the given rotating speed and the rotor rotating speed are differedsq，refThe torque component limiter value is determined by the current constraints. i.e. isq，refAnd isd,refAnd a feedback current isd-fdbAnd isq-fdbMaking difference, respectively obtaining u through complex vector current regulator* sqAnd u* sd。isd-fdb、isq-fdbFrom detected three-phase currents i of the machinea、ibAnd performing coordinate transformation and filtering on the obtained product.
Step two: given voltage vector u from current regulator outputsd，usqAnd outputting trigger pulses of a switching tube of the inverter by adopting an SVPWM (space vector pulse width modulation) strategy at a rotor magnetic chain angle. Wherein the angle of flux linkage is represented by isd-fdb、isq-fdbAnd rotor speed omegarAnd calculating. After the induction motor is started, the induction motor firstly runs in a constant torque area, and the rotating speed omega at the momentrAnd the SVPWM modulation is in a linear region, and the generated magnetic field is a uniform rotating magnetic field without current distortion.
Step three: with the speed of rotation omegarThe temperature of the mixture rises and the temperature of the mixture rises,the synchronous angular frequency is continuously increased, and the amplitude of the stator voltage is continuously increased when the stator voltage is increased
Figure BDA0001893398970000083
Equal to a given voltage of 2UdcAt/pi, the induction motor enters a constant power region (a weak magnetic region) from a constant torque region. As shown in FIG. 7, since the feedback voltage is equal to the reference 2U in the entire weak magnetic regiondcAnd/pi when the modulation is equal to 1, overmodulation binary modulation angle α when the modulation is 1 according to the above description of the overmodulation algorithmhAnd (3) after SVPWM overmodulation, the given voltage output by the current regulator is changed into a six-step wave, and the overmodulation process keeps the fundamental wave of the voltage unchanged, but changes the phase of the voltage vector according to the modulation angle, so that the inverter operates in a six-beat mode. Compared with the traditional flux weakening method of the inscribed circle voltage closed loop, the output fundamental wave voltage of the inverter is improved in limit, the stator current fundamental wave at the same rotating speed is improved, the flux weakening depth is weakened, and the maximum output torque of the motor in a flux weakening area is improved.
Step four: according to the previous description of the overmodulation algorithm, the six-beat operation is performed to boost the fundamental voltage by adjusting the phase, so that the six-beat operation causes severe phase voltage distortion, which further causes torque ripple and current harmonics. Aiming at the problem, the invention designs a band-stop resonant filter to reduce the influence of feedback current harmonics on a current regulator. The actual bandwidth of the current loop is equal to the design value omega of the bandwidth of the current loop* cResonant frequency omega of resonant filterhThe smaller value in between. While setting the filter resonant frequency to 6 omegae(sixth harmonic current frequency) and design value of current loop bandwidth (omega)* c) The larger value therebetween, as shown in fig. 8, when the motor is operated at the fundamental speed region, the filter resonance frequency is ωh＝ω* c>6ωeThe actual current loop bandwidth is equal to ω* cThe filter does not affect the current loop bandwidth; after entering the weak magnetic region, the resonant frequency of the filter is 6 omegae＝ωh>ω* cThus the actual current loop bandwidth is equal to ω* cThe filter still does not influence the bandwidth of the current loop, and the actual bandwidth value of the current loop is always equal to the design bandwidth value omega* cTherefore, the effect that the actual current loop bandwidth is not influenced when the harmonic wave is filtered by the resonance filter is achieved. Fig. 8 and 9 show a block diagram and a bode diagram of a designed resonant filter based on a generalized second-order integrator. It can be seen that the feedback current resonant filter with the self-adaptive rotating speed does not need motor parameters, has high robustness, and simultaneously reduces the phase delay generated by the filter according to the Baud diagram and the strong band-stop characteristic.
Step five: with the speed of rotation omegarAnd the current continues to rise, and the induction motor cannot keep the maximum current due to the limitation of the maximum slip rate and enters a constant voltage area (flux weakening area II). i.e. isd,refDecrease isq,refAlso reduced, the system is now only constrained by voltage and slip, and the dq axis voltages are satisfied
Figure BDA0001893398970000091
The motor will run to a given rotational speed in this operating region. Because the motor is always in a six-beat running mode in the weak magnetic region, the fundamental voltage and the output torque of the whole weak magnetic region are improved.
The experimental effect is as follows: according to fig. 10, the three-fold base speed step speed-up time is reduced from 0.796s at the inscribed circle voltage to 0.695s at six beats under six-beat operation, which shows that the torque output is increased, and the effectiveness of the invention is verified. Comparing the torque and a-phase stator current waveforms shows that the torque increases and the fluctuation of the torque and the stator current increases. Illustrating that phase current distortion is caused by voltage distortion caused by overmodulation. Fig. 11 shows the output torque at 100%, 200%, 300%, 400% of the rated speed using the algorithm of the present invention, and fig. 12 shows the output torque spectrum at the corresponding speed. From the torque spectrum analysis in fig. 12, it is understood that torque ripple occurs only in the weak magnetic region, and torque ripple caused by six beat operation is mainly torque ripple of 6 times fundamental frequency.
According to fig. 13, when the load torque is increased from no load to 20%, 40%, 60% of the rated load, the torque current and the stator current are increased as the load is increased. When the torque is increased to 20% and 40%, the traditional inscribed circle voltage and the proposed system under six-beat voltage can follow the given rotating speed, but when the load is increased to 60% of rated torque, the rotating speed under the voltage output of the inscribed circle can not follow the given rotating speed, and the rotating speed is reduced until the system enters a new steady state. In contrast, the rotating speed can still follow the given value under the six-beat operation, which shows that the system loading capacity is stronger under the six-beat operation, and the effectiveness of the invention is verified.
With reference to fig. 14, harmonic analysis is performed before and after the feedback current filtering, it can be seen that the harmonic current is mainly 6 th harmonic (the synchronous rotation speed is 150Hz and corresponds to 900Hz), and the resonance filter can effectively filter the sixth harmonic, thereby verifying the effectiveness of the resonance filter of the present invention. As can be seen from the vector locus of the output voltage of the current regulator shown in fig. 15, when there is no feedback current filter, the current regulator is affected by the feedback current at 6 th harmonic in the synchronous rotating coordinate system to output a given voltage containing 6 th harmonic, and the given voltage is converted into 5 th and 7 th harmonics after being converted into the two-phase stationary coordinate system, so that the vector locus of the voltage becomes hexagonal. The hexagonal voltage causes instability of a modulation degree parameter in an overshoot algorithm, and finally causes unstable six-beat operation, such as a six-beat output voltage time-domain waveform without a filter in fig. 15. In contrast, after the feedback current filter is added, because almost no current harmonic wave enters the current regulator, the regulator outputs a given voltage to be recovered to be circular, so that the modulation degree is stable, the overmodulation process becomes stable, and finally a stable six-beat voltage is output, thereby further verifying the effectiveness of the resonant filter in the invention.
Although the present invention has been described with reference to the preferred embodiments, it should be understood that various changes and modifications can be made therein by those skilled in the art without departing from the spirit and scope of the invention as defined in the appended claims.

Claims (3)

1. A weak magnetic control method for an induction motor based on optimized six-beat operation is characterized by comprising the following steps:
first of allStep, setting the given voltage value in the flux-weakening controller to be 2UdcThe feedback current resonant filter is built based on the generalized second-order integrator;
secondly, filtering the feedback current with the sixth harmonic by using the feedback current resonant filter, and generating a command voltage by using a complex vector current regulator after the difference is made between the feedback current and a current set value;
step three: secondly, the instruction voltage generated by the complex vector current regulator enters an SVPWM module to generate an inverter trigger pulse on one hand, and on the other hand, the instruction voltage is used as a feedback signal of the flux weakening first-zone controller to complete flux weakening control according to a voltage constraint equation;
step four, according to the weak magnetic control process in step three, the exciting current component produced by the weak magnetic controller gives isd，refThen, a torque current component given i is generated according to the current constraintsq，ref(ii) a Wherein isd，refAnd isq，refRespectively representing the d-axis and q-axis given currents of the stator;
step five: d-axis command voltage output by the complex vector current regulator is used as a flux weakening two-zone controller feedback signal as the rotating speed continues to rise, and dq-axis command voltage is satisfied
Figure FDA0001893398960000011
u* sd、u* sqThe dq-axis command voltages are respectively applied until a given rotation speed is reached.
2. The weak magnetic control method of the induction motor based on the optimized six-beat operation is characterized in that the step one of building a feedback current resonant filter based on the generalized second-order integrator comprises the following steps of:
will resonate at a frequency omegahSet to ω* cAnd 6. omegaeThe larger value in between, namely: e.g. omega* cAnd 6. omegaeThe larger value therebetween is ω* cThen the resonant frequency ωhSet to ω* cE.g. omega* cAnd 6. omegaeThe larger value between is 6. omegaeThen the resonant frequency ωhIs set to be 6 omegaeWherein, ω is* cFor the design value of the current loop bandwidth, ωeIs an instantaneous synchronous angular frequency and is 6 · ω in the fundamental regione<ω* cFilter resonant frequency omegah＝ω* cThe voltage of the motor stator is a standard sine wave, no sixth harmonic exists in the feedback stator current, and the resonant filter does not work.
3. The weak magnetic control method of the induction motor based on the optimized six-beat operation is characterized in that the process of generating the command voltage in the second step is as follows:
the detailed process of the control method comprises the following steps: when the induction motor enters a weak magnetic area along with the increase of the rotating speed, the weak magnetic control of the induction motor is started; according to the condition that the inverter outputs six-beat voltage in the weak magnetic region and generates a large amount of sixth harmonic in the stator current of the motor, the six-order current harmonic frequency in the weak magnetic region is 6 omegae>ω* cSetting the resonant frequency omega of the resonant filterh＝6·ωe(ii) a And attenuating the sixth harmonic through a resonant filter to obtain a current signal only with a fundamental current component, inputting the fundamental current component into a current closed-loop controller, and entering a complex vector current regulator to generate an instruction voltage after making a difference with a current given value.
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