CN111277186A - Induction motor field weakening control method based on optimized six-beat operation - Google Patents

Induction motor field weakening control method based on optimized six-beat operation Download PDF

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CN111277186A
CN111277186A CN201811481087.3A CN201811481087A CN111277186A CN 111277186 A CN111277186 A CN 111277186A CN 201811481087 A CN201811481087 A CN 201811481087A CN 111277186 A CN111277186 A CN 111277186A
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current
voltage
omega
field weakening
induction motor
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CN111277186B (en
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于泳
张旭
王勃
张静
徐殿国
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Delta Greentech China Co Ltd
Harbin Institute of Technology Shenzhen
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Harbin Institute of Technology Shenzhen
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    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/20Estimation of torque
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22Current control, e.g. using a current control loop
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters
    • H02P27/08Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters with pulse width modulation
    • H02P27/085Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency

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  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

本发明提出了一种基于优化六拍运行的感应电机弱磁控制方法,属于电机控制技术领域。所述控制方法首先将弱磁控制器中电压给定值设定为2Udc/π以保证系统可达到六拍运行模式,并将反馈电流谐振滤波器频率设置为ω* c与6·ωs之间的较大值;当转速超过额定转速后电机进入弱磁区,逆变器将输出六拍电压,六拍电压将造成电流六次谐波的产生,带有六次谐波的反馈电流经过谐振滤波器后与给定值做差后进入复矢量电流调节器,电流调节器产生给定电压,经SVPWM调制策略产生逆变器触发脉冲控制电机运行在弱磁区。本发明应用于感应电机的六拍运行弱磁控制。

Figure 201811481087

The invention provides a field weakening control method for an induction motor based on optimized six-beat operation, which belongs to the technical field of motor control. The control method first sets the voltage given value in the field weakening controller to 2U dc /π to ensure that the system can reach the six-beat operation mode, and sets the feedback current resonant filter frequency to ω * c and 6 ω s When the speed exceeds the rated speed, the motor enters the field weakening area, the inverter will output six-beat voltage, and the six-beat voltage will cause the generation of the sixth harmonic of the current, and the feedback current with the sixth harmonic will pass through After the resonant filter makes a difference with the given value, it enters the complex vector current regulator, the current regulator generates a given voltage, and the inverter trigger pulse is generated by the SVPWM modulation strategy to control the motor to run in the field weakening area. The invention is applied to the field weakening control of the six-beat operation of the induction motor.

Figure 201811481087

Description

一种基于优化六拍运行的感应电机弱磁控制方法A Field Weakening Control Method for Induction Motor Based on Optimal Six-beat Operation

技术领域technical field

本发明涉及一种基于优化六拍运行的感应电机弱磁控制方法,属于电机控制技术领域。The invention relates to a field weakening control method of an induction motor based on optimized six-beat operation, and belongs to the technical field of motor control.

背景技术Background technique

随着矢量控制在交流电机调速系统领域的广泛应用和不断发展,交流电机无论在驱动能力、动态响应或是调速范围与精度等方面都得到显著提高,也因此逐渐代替直流调速系统而被应用于包括家用电器、电梯曳引、汽车机床等各类工业应用场合。而同时,尤其是高性能的工业控制场合,如电动汽车、数控机床主轴驱动以及电力机车等领域对高精度高效率高质量的要求促使电机驱动变频调速系统朝着高速化方向发展。在电动汽车领域,面对复杂的道路交通状况,电动汽车的驱动需要具有运行于多种工况下的能力。在中/高速区域能够实现平稳高速行驶,以满足人们对于舒适安全性的需求。在数控机床用主轴驱动系统中,作为核心部件,主轴驱动控制性能的优劣直接决定了数控机床的整体水平。采用直驱方式能够达到高效率、高精度和高可靠性的要求,这也就使得驱动电机逐渐向着高速化方向发展。同样的,在大功率传动场合如电力机车领域,列车运行的高速化对电机车牵引传动系统的要求除了在高转速下的稳定运行,还包括始终能够输出最大转矩以克服列车高速运行过程中的巨大阻力。With the wide application and continuous development of vector control in the field of AC motor speed control systems, AC motors have been significantly improved in terms of driving capability, dynamic response, speed control range and accuracy, etc., and therefore gradually replaced DC speed control systems. It is used in various industrial applications including household appliances, elevator traction, and automotive machine tools. At the same time, especially in high-performance industrial control occasions, such as electric vehicles, CNC machine tool spindle drives, and electric locomotives, the high-precision, high-efficiency, and high-quality requirements have prompted the motor-driven variable frequency speed control system to develop toward high speed. In the field of electric vehicles, in the face of complex road traffic conditions, the drive of electric vehicles needs to have the ability to operate under various working conditions. In the medium/high-speed area, smooth high-speed driving can be achieved to meet people's needs for comfort and safety. In the spindle drive system for CNC machine tools, as the core component, the quality of the spindle drive control performance directly determines the overall level of the CNC machine tool. The direct drive method can meet the requirements of high efficiency, high precision and high reliability, which makes the drive motor gradually develop towards high speed. Similarly, in the field of high-power transmission, such as the field of electric locomotives, the requirements of the high-speed train operation for the traction drive system of electric locomotives not only require stable operation at high speeds, but also always output the maximum torque to overcome the high-speed operation of the train. huge resistance.

高性能控制场合对电机调速提出的要求包括:调速范围宽(尤其是高速区),高速稳定运行能力,最大转矩输出能力,充分利用有限的电压电流资源,高动态响应等。The requirements for motor speed regulation in high-performance control occasions include: wide speed regulation range (especially high-speed area), high-speed stable operation capability, maximum torque output capability, full use of limited voltage and current resources, high dynamic response, etc.

但是传统交流电机高速化驱动存在以下两个问题:However, the high-speed drive of traditional AC motors has the following two problems:

1、传统地采用SVPWM调制方法只能使三相相电压输出幅值达到直流母线电压的

Figure BDA0001893398970000011
而SVPWM模块能够输出的最大基波电压幅值为直流母线电压的2/π,这就说明了传统方法中存在对直流母线电压资源的浪费。1. The traditional SVPWM modulation method can only make the output amplitude of the three-phase phase voltage reach the same value as the DC bus voltage.
Figure BDA0001893398970000011
The maximum fundamental voltage amplitude that the SVPWM module can output is 2/π of the DC bus voltage, which shows that there is a waste of DC bus voltage resources in the traditional method.

2、若采用过调制技术对逆变器输出电压进行极限拓展,电流畸变不可避免,不得不使用滤波器进行滤波,而传统滤波器会导致电流带宽减小,对整个系统的控制性能产生影响。2. If the overmodulation technology is used to expand the output voltage of the inverter to the limit, the current distortion is inevitable, and the filter has to be used for filtering. The traditional filter will reduce the current bandwidth and affect the control performance of the entire system.

发明内容SUMMARY OF THE INVENTION

本发明为了解决电机定子电压的非线性拓展造成谐波增大,并造成系统性能变差的问题,同时针对传统滤波器会减小电流环带宽的问题,提出了一种基于优化六拍运行的感应电机弱磁控制方法。In order to solve the problem that the non-linear expansion of the stator voltage of the motor causes the harmonics to increase and the system performance deteriorates, and at the same time for the traditional filter to reduce the current loop bandwidth, a new method based on optimized six-beat operation is proposed. Induction motor field weakening control method.

本发明所述的一种基于优化六拍运行的感应电机弱磁控制方法,使用于感应电机系统处于高速弱磁运行状态下,所采取的技术方案如下:A field weakening control method for an induction motor based on optimized six-beat operation according to the present invention is used when the induction motor system is in a high-speed field weakening operation state, and the adopted technical solutions are as follows:

一种基于优化六拍运行的感应电机弱磁控制方法,所述控制方法包括:A field weakening control method for an induction motor based on optimized six-beat operation, the control method comprising:

第一步、将弱磁控制器中的电压给定值设定为2Udc/π以保证系统可达到六拍运行模式,并基于广义二阶积分器搭建反馈电流谐振滤波器;The first step is to set the voltage given value in the field weakening controller to 2U dc /π to ensure that the system can reach the six-beat operation mode, and build a feedback current resonant filter based on a generalized second-order integrator;

第二步、利用所述反馈电流谐振滤波器对带有六次谐波的反馈电流进行滤波处理并与电流给定值做差后利用复矢量电流调节器产生指令电压;The second step is to use the feedback current resonant filter to filter the feedback current with the sixth harmonic and to use the complex vector current regulator to generate the command voltage after making a difference with the current given value;

步骤三:步骤二所述复矢量电流调节器产生的指令电压一方面进入SVPWM调制模块产生逆变器触发脉冲,另一方面根据电压约束方程作为弱磁一区控制器的反馈信号完成弱磁控制;Step 3: On the one hand, the command voltage generated by the complex vector current regulator described in Step 2 enters the SVPWM modulation module to generate the inverter trigger pulse; ;

步骤四、根据步骤三中所述弱磁控制过程中,弱磁控制器产生的励磁电流分量给定isd,ref,再根据电流约束产生转矩电流分量给定isq,ref;其中,isd,ref和isq,ref分别表示定子d轴和q轴给定电流;Step 4, according to the field weakening control process described in step 3, the excitation current component generated by the field weakening controller is given i sd, ref , and then the torque current component is given i sq, ref according to the current constraint; wherein, i sd, ref and i sq, ref represent the given current of stator d-axis and q-axis respectively;

步骤五:随着转速继续升高,为满足最大转差率限制,将复矢量电流调节器输出的d轴指令电压作为弱磁二区控制器反馈信号,并且dq轴指令电压之间满足

Figure BDA0001893398970000021
u* sd、u* sq分别为dq轴指令电压,直至到达给定转速。Step 5: As the speed continues to increase, in order to meet the maximum slip rate limit, the d-axis command voltage output by the complex vector current regulator is used as the feedback signal of the field weakening second-zone controller, and the dq-axis command voltages satisfy
Figure BDA0001893398970000021
u * sd and u * sq are respectively the command voltage of dq axis until reaching the given speed.

进一步地,步骤一所述基于广义二阶积分器搭建反馈电流谐振滤波器的过程包括:Further, the process of building a feedback current resonant filter based on a generalized second-order integrator described in step 1 includes:

将谐振频率ωh设置为ω* c与6·ωe之间的较大值,即:如ω* c与6·ωe之间的较大值为ω* c,则将谐振频率ωh设置为ω* c,如ω* c与6·ωe之间的较大值为6·ωe,则将谐振频率ωh设置为6·ωe,其中,ω* c为电流环带宽的设计值,ωe为瞬时同步角频率,并且在基速区内6·ωe* c,滤波器谐振频率ωh=ω* c,电机定子电压为标准正弦波,反馈定子电流中无六次谐波,谐振滤波器不起作用。The resonant frequency ω h is set to the larger value between ω * c and 6·ω e , that is: if the larger value between ω * c and 6·ω e is ω * c , then the resonance frequency ω h is set to Set to ω * c , if the larger value between ω * c and 6 · ω e is 6 · ω e , then set the resonant frequency ω h to 6 · ω e , where ω * c is the current loop bandwidth The design value, ω e is the instantaneous synchronization angular frequency, and in the base speed region 6·ω e* c , the filter resonant frequency ω h* c , the motor stator voltage is a standard sine wave, and there is no feedback in the stator current. Sixth harmonic, resonant filters do not work.

进一步地,步骤二所述产生指令电压的过程为:Further, the process of generating the command voltage described in step 2 is:

所述控制方法的详细过程包括:随转速升高,当感应电机进入弱磁区时,开始对感应电机进行弱磁控制;根据此时逆变器在弱磁区输出六拍电压并造成电机定子电流中产生大量的六次谐波时,弱磁区内六次谐波频率频率6·ωe* c的特性,设定谐振滤波器的谐振频率ωh=6·ωe;将所述六次谐波通过谐振滤波器进行削弱处理获得只有基波电流分量的电流信号,将所述基波电流分量输入至电流闭环控制器中,与电流给定值作差后进入复矢量电流调节器产生指令电压。The detailed process of the control method includes: as the rotation speed increases, when the induction motor enters the field weakening region, the field weakening control is started to the induction motor; When a large number of sixth harmonics are generated, the characteristic of the sixth harmonic frequency frequency 6·ω e* c in the weak magnetic region, the resonant frequency of the resonant filter is set to ω h =6·ω e ; The harmonics are weakened by the resonant filter to obtain the current signal with only the fundamental current component, and the fundamental current component is input into the current closed-loop controller, and then enters the complex vector current regulator after making a difference with the current given value. Voltage.

其中,谐振滤波器谐振频率选取方式使得该滤波器在全速域范围内不影响电流环带宽;同时在谐振滤波器作用下电流调节器输出指令电压u* sdq在同步旋转坐标系内不存在6次谐波,在两相静止坐标内不存在5次、7次谐波,保证六拍运行下提升弱磁区输出转矩的同时控制模块(电流调节器与电压调节器)不受谐波干扰。Among them, the resonant frequency of the resonant filter is selected so that the filter does not affect the current loop bandwidth in the full-speed domain; at the same time, under the action of the resonant filter, the output command voltage u * sdq of the current regulator does not exist 6 times in the synchronous rotating coordinate system Harmonics, there are no 5th and 7th harmonics in the two-phase static coordinates, which ensures that the control module (current regulator and voltage regulator) is not disturbed by harmonics while improving the output torque of the field weakening area under six-beat operation.

本发明有益效果:Beneficial effects of the present invention:

本发明提出的一种基于优化六拍运行的感应电机弱磁控制方法能在感应电机处于弱磁区六拍运行的同时,实现控制器中无谐波运行并保持电流带宽不受影响,实现了感应电机弱磁区稳定的六拍运行,提升了弱磁区最大转矩并优化了动静态性能。本发明提出的一种基于优化六拍运行的感应电机弱磁控制方法通过对弱磁控制器与电流闭环进行有效的改进设计,实现了在控制模块不受影响的情况下提升最大转矩。并且根据电流环的特点,设计出了不影响电流环带宽的反馈电流滤波器,保证电流环动态性能。A field weakening control method for an induction motor based on optimized six-beat operation proposed by the invention can realize no harmonic operation in the controller and keep the current bandwidth unaffected while the induction motor is in the six-beat operation in the field weakening region, thereby realizing the induction motor. The stable six-beat operation in the field weakening area of the motor improves the maximum torque in the field weakening area and optimizes the dynamic and static performance. An induction motor field weakening control method based on optimized six-beat operation proposed by the present invention realizes the increase of the maximum torque without the control module being affected by effectively improving the design of the field weakening controller and the current closed loop. And according to the characteristics of the current loop, a feedback current filter that does not affect the bandwidth of the current loop is designed to ensure the dynamic performance of the current loop.

附图说明Description of drawings

图1:基于电压闭环的传统感应电机弱磁控制系统框图;Figure 1: Block diagram of traditional induction motor field weakening control system based on voltage closed loop;

图2:感应电机最大转矩输出电压电流轨迹;Figure 2: Induction motor maximum torque output voltage and current trajectory;

图3:过调制一区定子电压矢量轨迹;Figure 3: Overmodulation zone one stator voltage vector trace;

图4:过调制一区定子相电压空间矢量轨迹与时域波形对应图;Figure 4: The corresponding diagram of the space vector trajectory of the stator phase voltage in the overmodulation area and the time domain waveform;

图5:过调制二区定子电压矢量轨迹;Figure 5: Overmodulation two-zone stator voltage vector trace;

图6:过调制二区定子相电压空间矢量轨迹与时域波形对应图;Figure 6: The corresponding diagram of the space vector trace of the stator phase voltage in the second region of overmodulation and the time domain waveform;

图7:基于优化六拍运行的感应电机弱磁控制系统框图;Figure 7: Block diagram of induction motor field weakening control system based on optimized six-beat operation;

图8:基于广义二阶积分器(SOGI)的反馈电流谐振滤波器框图;Figure 8: Block diagram of a feedback current resonant filter based on a generalized second-order integrator (SOGI);

图9:反馈电流谐振滤波器bode图;Figure 9: Bode diagram of feedback current resonant filter;

图10:所发明方法与传统方法在三倍基速阶跃加速过程的对比实验结果,其中,(a)为传统内切圆方法实验结果图,(b)为本发明所述方法的实验结果图;Figure 10: Comparative experimental results between the invented method and the traditional method in the step acceleration process of three times the base speed, wherein (a) is the experimental result of the traditional inscribed circle method, (b) is the experimental result of the method of the present invention picture;

图11:所发明方法在1、2、3、4倍基速下输出转矩情况;Figure 11: The output torque of the invented method at 1, 2, 3, and 4 times the base speed;

图12:所发明方法在1、2、3、4倍基速下的稳态转矩频谱,其中,(a)为1倍基速下稳态转矩频谱图,(b)为2倍基速下稳态转矩频谱图,(c)为3倍基速下稳态转矩频谱图,(d)为4倍基速下稳态转矩频谱图;Figure 12: The steady-state torque spectrum of the invented method at 1, 2, 3, and 4 times the base speed, where (a) is the steady-state torque spectrum at 1 times the base speed, (b) is the 2 times the base speed Spectrum of steady-state torque at high speed, (c) is the frequency spectrum of steady-state torque at 3 times the base speed, (d) is the frequency spectrum of steady-state torque at 4 times the base speed;

图13:传统方法与所发明方法在0%、20%、40%、60%负载下转速、转矩电流、相电流情况及相电流THD情况,其中,(a)为传统内切圆方法的情况波形图,(b)为本发明所述方法的情况波形图,(c)为不同负载下本发明方法中相电流波形总畸变率;Figure 13: Condition of rotational speed, torque current, phase current and phase current THD at 0%, 20%, 40%, 60% load of traditional method and invented method, where (a) is the traditional inscribed circle method Situation waveform diagram, (b) is the situation waveform diagram of the method of the present invention, (c) is the total distortion rate of the phase current waveform in the method of the present invention under different loads;

图14:滤波前后的反馈电流及其频谱,其中,(a)为无谐波滤波器反馈电流及其频谱,(b)为经过本发明所述谐波滤波器后的反馈电流及其频谱;Figure 14: The feedback current and its frequency spectrum before and after filtering, wherein, (a) is the feedback current and its frequency spectrum without harmonic filter, (b) is the feedback current and its frequency spectrum after passing through the harmonic filter of the present invention;

图15:有无反馈电流谐振滤波器情况下的电流调节器输出电压轨迹、两相静止坐标系下电压时域波形,其中,(a)为无反馈电流滤波器的电流调节器输出电压轨迹、两相静止坐标系下电压时域波形图,(b)为有反馈滤波器的电流调节器输出电压轨迹、两相静止坐标系下电压时域波形图。Figure 15: Current regulator output voltage trace with and without feedback current resonant filter, voltage time-domain waveform in two-phase static coordinate system, where (a) is the current regulator output voltage trace without feedback current filter, The voltage time-domain waveform diagram in the two-phase static coordinate system, (b) is the output voltage trace of the current regulator with the feedback filter, and the voltage time-domain waveform diagram in the two-phase static coordinate system.

具体实施方式Detailed ways

下面将结合本专利实施例中的附图对本发明实施例中的方案进行清楚、完整地描述,但本发明不受实施例的限制,以下所描述的实施例仅是本发明中一部分实施例,并不是全部实施例。基于本发明中的实施例,本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实例,都属于本发明保护的范围。The solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention, but the present invention is not limited by the embodiments, and the embodiments described below are only a part of the embodiments of the present invention, Not all examples. Based on the embodiments of the present invention, all other examples obtained by those of ordinary skill in the art without creative efforts shall fall within the protection scope of the present invention.

实施例1:Example 1:

下面将结合本专利实施例中的附图对本发明实施例中的方案进行清楚、完整地描述,显然,所描述的实施例仅是本发明中一部分实施例,并不是全部实施例。基于本发明中的实施例,本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实例,都属于本发明保护的范围。The solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other examples obtained by those of ordinary skill in the art without creative efforts shall fall within the protection scope of the present invention.

本发明基于传统的基于电压闭环的弱磁控制策略,下面首先结合图1对电压闭环弱磁控制策略进行阐述:The present invention is based on the traditional voltage closed-loop based field weakening control strategy. The following first describes the voltage closed-loop field weakening control strategy with reference to FIG. 1 :

在电压闭环弱磁控制策略中感应电机的运行范围可分为三个区间:恒转矩区、恒功率区(弱磁一区)、恒电压区(弱磁二区)。在电动机转速低于额定转速的区间,称为恒转矩区。在恒转矩区,定子电流为额定电流ismax,电压与转速保持固定比例增长,感应电机运行的反电动势小于额定电压值,输出额定转矩并保持恒定。当电压升高至额定电压usmax时,感应电机达到额定转速,进入恒功率区。由于最大电压usmax和最大电流的限制ismax,在此区间内,能够输出额定功率。在恒电压区,由于稳定运行最大转差率的限制,电机不能保持最大电流,同时受最大电压usmax的限制,输出转矩和功率随转速升高急剧减小。In the voltage closed-loop field weakening control strategy, the operating range of the induction motor can be divided into three zones: constant torque zone, constant power zone (field weakening zone 1), and constant voltage zone (field weakening zone 2). In the interval where the motor speed is lower than the rated speed, it is called the constant torque zone. In the constant torque region, the stator current is the rated current ismax , the voltage and the rotational speed keep a constant increase, the back electromotive force of the induction motor is less than the rated voltage value, and the rated torque is output and kept constant. When the voltage rises to the rated voltage u smax , the induction motor reaches the rated speed and enters the constant power region. Due to the limitation of the maximum voltage usmax and the maximum current ismax , within this interval, the rated power can be output. In the constant voltage region, due to the limitation of the maximum slip rate in stable operation, the motor cannot maintain the maximum current, and at the same time, limited by the maximum voltage u smax , the output torque and power decrease sharply with the increase of the speed.

感应电机按转子磁链定向的数学模型如下:The mathematical model of the induction motor oriented according to the rotor flux linkage is as follows:

Figure BDA0001893398970000041
Figure BDA0001893398970000041

式中:usd,usq为d-q轴系定子电压分量;isd,isq为d-q轴系定子电流分量;ωe为电角度同步转速;Rs为定子电阻;Ls,Lr为定转子自感;Lm为互感;σ为漏感因子,

Figure BDA0001893398970000042
ψr为转子磁链。In the formula: u sd , u sq are the dq shaft stator voltage components; i sd , i sq are the dq shaft stator current components; ω e is the electrical angle synchronous speed; R s is the stator resistance; L s , L r are the constant Rotor self-inductance; L m is mutual inductance; σ is leakage inductance factor,
Figure BDA0001893398970000042
ψ r is the rotor flux linkage.

在感应电机运行过程中,主要受到最大电压和最大电流约束。这些约束是考虑电机和逆变器自身限制及运行环境综合决定的,如逆变器允许的最大输出电压,电机允许的最大电流及不同工况下的工作时间、散热条件等。因此,受限下的最大转矩控制可以描述为:During the operation of the induction motor, it is mainly constrained by the maximum voltage and maximum current. These constraints are determined by considering the limitations of the motor and the inverter itself and the operating environment, such as the maximum allowable output voltage of the inverter, the maximum allowable current of the motor, and the working time and heat dissipation conditions under different working conditions. Therefore, the limited maximum torque control can be described as:

最大值:

Figure BDA0001893398970000051
Max:
Figure BDA0001893398970000051

约束条件:

Figure BDA0001893398970000052
Restrictions:
Figure BDA0001893398970000052

如果忽略式(1)中的定子电阻及暂态部分,将其与式(3)结合,得到表示为电压形式下的弱磁区约束条件为:If the stator resistance and transient part in equation (1) are ignored and combined with equation (3), the constraint conditions of the weak magnetic region expressed in the form of voltage are:

Figure BDA0001893398970000053
Figure BDA0001893398970000053

认为usmax和ismax均设为恒定值,那么随着感应电机转速的升高,理想的约束轨迹曲线如图2所示,其中ABO表示电流轨迹,OCD表示电压轨迹。这样整个运行区间可以划分为三个区域:恒转矩区对应为点A与线段OC,恒功率区(弱磁一区)对应为曲线段AB与CD,恒电压区(弱磁二区)对应为线段BO与点D。那么感应电机弱磁区最大转矩控制就是保证随着转速的升高,电流矢量轨迹可以沿着ABO运行而电压矢量轨迹可以沿着OCD段运行并最终停留在D点。Considering that both u smax and is smax are set to constant values, then as the speed of the induction motor increases, the ideal constrained trajectory curve is shown in Figure 2, where ABO represents the current trajectory and OCD represents the voltage trajectory. In this way, the entire operating range can be divided into three areas: the constant torque area corresponds to point A and line segment OC, the constant power area (field weakening area 1) corresponds to curve segments AB and CD, and the constant voltage area (magnetic field weakening area 2) corresponds to is the line segment BO and the point D. Then the maximum torque control in the field weakening region of the induction motor is to ensure that as the speed increases, the current vector trajectory can run along the ABO and the voltage vector trajectory can run along the OCD segment and finally stay at point D.

图1给出了基于间接转子磁链定向的电压闭环弱磁控制框图。在通常使用的双闭环结构基础上,额外使用两个PI电压闭环用于迫使电压矢量按照图2所示的理想最大转矩轨迹运行。其中,控制器I用于弱化d轴电流给定,故也称为弱磁控制器,给定设为

Figure BDA0001893398970000054
即空间矢量脉宽调制最大输出电压六边形的内切圆半径大小,反馈值为电流环输出的电压矢量幅值,限幅值设为0。控制器II用于恒电压区弱化q轴电流给定,给定设为
Figure BDA0001893398970000055
(对应图2中点D),反馈值为d轴电压幅值,限幅值为0。Figure 1 shows the block diagram of voltage closed-loop field weakening control based on indirect rotor flux linkage orientation. On the basis of the commonly used double closed loop structure, two additional PI voltage closed loops are used to force the voltage vector to operate according to the ideal maximum torque trajectory shown in Figure 2. Among them, the controller I is used to weaken the d-axis current given, so it is also called the field weakening controller, and the given value is set as
Figure BDA0001893398970000054
That is, the radius of the inscribed circle of the hexagon of the maximum output voltage of the space vector pulse width modulation, the feedback value is the voltage vector amplitude output by the current loop, and the limit value is set to 0. The controller II is used to weaken the q-axis current given in the constant voltage area, and the given is set to
Figure BDA0001893398970000055
(Corresponding to point D in Figure 2), the feedback value is the d-axis voltage amplitude, and the limit value is 0.

电机启动后,励磁电流维持在给定值保证合适的磁场大小,转矩电流通过

Figure BDA0001893398970000056
限定,从而保证最大电流状态。此阶段电机只受到最大电流约束,电压控制器不起作用,输出为0,转速升速很快,dq轴电压|usd|usq快速上升,反电动势迅速逼近逆变器最大输出电压。由转矩方程式(2)可知,此阶段可以保证恒定的转矩输出能力,故称为恒转矩区。After the motor starts, the excitation current is maintained at the given value to ensure a suitable magnetic field size, and the torque current passes through
Figure BDA0001893398970000056
limit, thereby guaranteeing the maximum current state. At this stage, the motor is only constrained by the maximum current, the voltage controller does not work, the output is 0, the speed increases rapidly, the dq axis voltage |u sd |u sq rises rapidly, and the back EMF quickly approaches the maximum output voltage of the inverter. From the torque equation (2), it can be known that this stage can ensure constant torque output capability, so it is called the constant torque zone.

当反电动势上升至弱磁控制器(控制器I)开始作用,意味着电机进入恒功率区。id跟随id,ref的减小而减小,使d轴磁链始终保持理想弱磁状态,对应于输出电压始终维持在约束电压大小。此时转矩电流iq

Figure BDA0001893398970000061
作用而增大,依然保持最大电流状态。根据电压方程知转速ωr和转矩电流iq同时增大使得|usd|迅速上升,uq则被迫开始下降。整个恒功率区电机同时受到最大电压和最大电流双重约束,弱磁控制器一直作用。When the back EMF rises to the point where the field weakening controller (controller I) starts to function, it means that the motor enters the constant power region. id decreases with the decrease of id,ref , so that the d -axis flux linkage always maintains the ideal field weakening state, which corresponds to the output voltage always maintaining the size of the constraint voltage. At this time, the torque current i q is affected by
Figure BDA0001893398970000061
The effect is increased, and the maximum current state is still maintained. According to the voltage equation, it is known that the simultaneous increase of the rotational speed ω r and the torque current i q makes |u sd | rise rapidly, and u q is forced to start to decrease. The motor in the entire constant power area is subject to the dual constraints of the maximum voltage and the maximum current at the same time, and the field weakening controller is always on.

当|usd|上升至

Figure BDA0001893398970000062
电压控制器II开始作用,意味着电机进入恒电压区,q轴电流的减小使得ud维持在
Figure BDA0001893398970000063
处,而uq也将自然调整到
Figure BDA0001893398970000064
处,在此阶段,最大电流状态无法满足,电机只受最大电压约束,两个电压控制器一直作用。when |u sd | rises to
Figure BDA0001893398970000062
The voltage controller II starts to function, which means that the motor enters the constant voltage region, and the reduction of the q-axis current keeps ud at
Figure BDA0001893398970000063
, and u q will naturally adjust to
Figure BDA0001893398970000064
At this stage, the maximum current state cannot be satisfied, the motor is only constrained by the maximum voltage, and the two voltage controllers are always on.

其次本发明应用于六拍运行模式,由于六拍运行为过调制运行的最终状态,因此下面对过调制算法进行阐述:Secondly, the present invention is applied to the six-beat operation mode. Since the six-beat operation is the final state of the overmodulation operation, the overmodulation algorithm is described below:

定义调制度系数M为:The modulation factor M is defined as:

Figure BDA0001893398970000065
Figure BDA0001893398970000065

其中

Figure BDA0001893398970000066
为电流调节器输出给定电压矢量模长,2Udc/π为逆变器能够输出的最大基波电压幅值。in
Figure BDA0001893398970000066
Given the voltage vector modulus length for the current regulator output, 2U dc /π is the maximum fundamental voltage amplitude that the inverter can output.

本发明中过调制算法采用定子电压基波幅值等效原则,随调制度系数的增大分为过调制一区与过调制二区。下面首先介绍过调制一区,如图3中虚线所示,当u* sdq超过正六边形后,超出部分的电压矢量轨迹会被限制在正六边形边界,由此必将引起定子电压基波幅值的减小,需要增加靠近六边形顶点处的电压矢量的幅值来弥补这部分损失,如图3中实线所示。In the present invention, the overmodulation algorithm adopts the principle of equivalent amplitude of the fundamental wave of the stator voltage, and is divided into the first overmodulation region and the second overmodulation region with the increase of the modulation factor. The first overmodulation area is introduced below. As shown by the dotted line in Figure 3, when u * sdq exceeds the regular hexagon, the voltage vector trajectory of the excess part will be limited to the regular hexagon boundary, which will inevitably cause the stator voltage fundamental wave The reduction of the amplitude requires an increase in the amplitude of the voltage vector near the vertex of the hexagon to make up for this loss, as shown by the solid line in Figure 3.

在图三中定义过调制一区调制角αr,则根据图4可得四分之一周期内(0<θ<π/2)定子相电压瞬时值u与αr存在如下关系:Define the modulation angle α r in the first area of overmodulation in Figure 3, then according to Figure 4, it can be obtained that the instantaneous value u of the stator phase voltage in a quarter cycle (0<θ<π/2) has the following relationship with α r :

Figure BDA0001893398970000071
Figure BDA0001893398970000071

式中θ为电流调节器输出电压矢量的相位角。where θ is the phase angle of the current regulator output voltage vector.

由公式(7)可得u的基波分量U1与αr有如下关系:From formula (7), the fundamental wave component U 1 of u and α r can be obtained as follows:

Figure BDA0001893398970000072
Figure BDA0001893398970000072

则根据基波幅值等效原则可得到αr与调制度M的关系如下:Then according to the principle of equivalence of fundamental wave amplitude, the relationship between α r and modulation degree M can be obtained as follows:

Figure BDA0001893398970000073
Figure BDA0001893398970000073

随着

Figure BDA0001893398970000074
的进一步增大,αr由π/6不断减小直至0,则通过该方法无法进一步增大定子电压基波幅值,只能通过增加六边形顶点处电压矢量的停留时间来进一步增加定子电压的基波幅值,进入过调制二区。along with
Figure BDA0001893398970000074
The further increase of α r from π/6 to 0, the stator voltage fundamental wave amplitude cannot be further increased by this method, and the stator voltage can only be further increased by increasing the dwell time of the voltage vector at the hexagonal vertex The fundamental amplitude of the voltage enters the second area of overmodulation.

如图5所示定义二区调制角αh As shown in Fig. 5, the second region modulation angle α h is defined

以扇区一内电压矢量为例说明过调制二区算法。如图5所示,当给定电压矢量

Figure BDA0001893398970000075
由A旋转至B时,实际电压矢量停留在A点,当
Figure BDA0001893398970000076
由B旋转至C,实际电压矢量以更快的转速转至D点,当
Figure BDA0001893398970000077
由C旋转至D,实际电压矢量停留在D点,如此便在改变了给定电压矢量相位的条件下增加了电压幅值。调制后的电压矢量相位角θ0与给定电压矢量相位角θ的关系如下:Taking the voltage vector in sector one as an example, the overmodulation two-zone algorithm is described. As shown in Figure 5, when a voltage vector is given
Figure BDA0001893398970000075
When rotating from A to B, the actual voltage vector stays at point A, when
Figure BDA0001893398970000076
Rotating from B to C, the actual voltage vector rotates to point D at a faster speed, when
Figure BDA0001893398970000077
Rotating from C to D, the actual voltage vector stays at point D, thus increasing the voltage amplitude while changing the phase of the given voltage vector. The relationship between the modulated voltage vector phase angle θ 0 and the given voltage vector phase angle θ is as follows:

Figure BDA0001893398970000078
Figure BDA0001893398970000078

根据图6同样可以得到过调制二区相电压瞬时值u2与αh的关系如下:According to Fig. 6, the relationship between the instantaneous value of phase voltage u 2 in the second region of overmodulation and α h can also be obtained as follows:

Figure BDA0001893398970000081
Figure BDA0001893398970000081

可以得到αh与调制度M得关系如下:The relationship between α h and the modulation degree M can be obtained as follows:

Figure BDA0001893398970000082
Figure BDA0001893398970000082

当αh=π/6时,定子电压矢量在六边形六个顶点依次跳变,称该状态下逆变器处于六拍模式,输出电压的基波幅值也达到了最大值,这也是本发明中弱磁区内逆变器的运行状态。When α h = π/6, the stator voltage vector jumps in turn at the six vertices of the hexagon, and it is said that the inverter is in the six-beat mode in this state, and the fundamental amplitude of the output voltage also reaches the maximum value, which is also The operating state of the inverter in the field weakening area in the present invention.

在以上对电压闭环理论与过调制算法的介绍基础上,下面对本发明进行详细介绍:On the basis of the above introduction to the voltage closed-loop theory and the overmodulation algorithm, the present invention is introduced in detail below:

步骤一:结合图7,将参考电压给定设置为2Udc/π,参考电压与反馈电压做差后进入PI调节器,通过PI调节和限幅作用得到定子电流励磁分量给定值isd,ref;同时采用测速码盘得到转子转速ωr,给定转速与转子转速做差后经过PI得到定子电流转矩分量给定值isq,ref,转矩分量限幅值由电流约束条件决定。isq,ref和isd,ref与反馈电流isd-fdb和isq-fdb做差,经过复矢量电流调节器分别得到u* sq和u* sd。isd-fdb、isq-fdb由检测到的电机三相电流ia、ib,对其进行坐标变换和滤波得到。Step 1: Combined with Figure 7, set the reference voltage given to 2U dc /π, enter the PI regulator after the difference between the reference voltage and the feedback voltage, and obtain the stator current excitation component given value i sd through PI adjustment and amplitude limiting, ref ; at the same time, the rotor speed ω r is obtained by using the speed-measuring code disc. After the difference between the given speed and the rotor speed, the given value of the stator current torque component i sq, ref is obtained through PI. The torque component limit value is determined by the current constraint. i sq, ref and i sd, ref and feedback currents i sd-fdb and i sq-fdb make a difference, and obtain u * sq and u * sd respectively through the complex vector current regulator. i sd-fdb and i sq-fdb are obtained by performing coordinate transformation and filtering on the detected three-phase currents ia and ib of the motor.

步骤二:根据电流调节器输出的给定电压矢量usd,usq与转子磁链角,采用SVPWM调制策略输出逆变器开关管触发脉冲。其中,磁链角由isd-fdb、isq-fdb和转子转速ωr计算得来。感应电机启动后,首先会运行在恒转矩区,此时转速ωr较小,SVPWM调制处于线性区,产生磁场为均匀旋转磁场,未带来电流畸变。Step 2: According to the given voltage vector u sd , u sq and the rotor flux linkage angle output by the current regulator, the SVPWM modulation strategy is used to output the trigger pulse of the switch tube of the inverter. Among them, the flux linkage angle is calculated by isd-fdb , isq-fdb and rotor speed ωr . After the induction motor is started, it will first run in the constant torque region. At this time, the rotational speed ω r is small, and the SVPWM modulation is in the linear region, and the generated magnetic field is a uniform rotating magnetic field without current distortion.

步骤三:随着转速ωr升高,同步角频率不断升高,同时定子电压幅值不断上升,当定子电压

Figure BDA0001893398970000083
等于给定电压2Udc/π时,感应电机从恒转矩区进入恒功率区(弱磁一区)。如图7所示,由于整个弱磁区内反馈电压等于参考的2Udc/π,此时调制度等于1,根据前文对过调制算法的说明,调制度为1时过调制二区调制角αh=π/3,电流调节器输出的给定电压经过SVPWM过调制后变为六阶梯波,该过调制过程保持电压的基波不变,但根据调制角改变了电压矢量的相位,使得逆变器运行在六拍模式。相比于传统内切圆电压闭环的弱磁方法,逆变器输出基波电压得到极限提升,并提升了相同转速下的定子电流基波,减弱了弱磁深度,进而提升了弱磁区的电机最大输出转矩。Step 3: As the rotational speed ω r increases, the synchronous angular frequency continues to increase, and at the same time, the amplitude of the stator voltage continues to increase.
Figure BDA0001893398970000083
When equal to a given voltage of 2U dc /π, the induction motor enters the constant power region (field weakening region) from the constant torque region. As shown in Fig. 7, since the feedback voltage in the entire weak field area is equal to the reference 2U dc /π, the modulation degree is equal to 1 at this time. According to the description of the overmodulation algorithm above, when the modulation degree is 1, the modulation angle α h of the second overmodulation area is overmodulated =π/3, the given voltage output by the current regulator becomes a six-step wave after SVPWM overmodulation. This overmodulation process keeps the fundamental wave of the voltage unchanged, but changes the phase of the voltage vector according to the modulation angle, so that the inverter The device operates in six-beat mode. Compared with the traditional field weakening method of inscribed circle voltage closed-loop, the inverter output fundamental voltage is greatly improved, and the stator current fundamental wave at the same speed is improved, weakening the field weakening depth, and improving the motor in the field weakening area. maximum output torque.

步骤四:根据之前对过调制算法的说明,六拍运行是通过调整相位对基波电压进行提升的,因此六拍运行将造成严重的相电压畸变,进一步导致转矩波动和电流谐波。针对该问题本发明设计了带阻的谐振滤波器以减轻反馈电流谐波对电流调节器的影响。由于电流环实际带宽等于电流环带宽设计值ω* c与谐振滤波器谐振频率ωh之间的较小值。同时将滤波器谐振频率设置为6ωe(六次谐波电流频率)与电流环带宽设计值(ω* c)之间的较大值,如图8所示,当电机运行于基速区时,滤波器谐振频率为ωh=ω* c>6ωe,实际电流环带宽等于ω* c,滤波器不影响电流环带宽;进入弱磁区后,滤波器谐振频率为6ωe=ωh* c,因此实际电流环带宽等于ω* c,滤波器仍不会影响电流环带宽,电流环实际带宽值始终等于带宽设计值ω* c,从而实现了谐振滤波器在滤除谐波的同时不会影响实际电流环带宽的效果。图8、图9给出了所设计的基于广义二阶积分器的谐振滤波器框图与传递函数波特图。可已看出本发明中的转速自适应的反馈电流谐振滤波器不需要电机参数,有很高的鲁棒性,同时根据波特图可知较强的带阻特性也减小了滤波器产生的相位延迟。Step 4: According to the previous description of the overmodulation algorithm, the six-beat operation improves the fundamental voltage by adjusting the phase. Therefore, the six-beat operation will cause severe phase voltage distortion, which will further lead to torque fluctuations and current harmonics. Aiming at this problem, the present invention designs a band-stop resonant filter to reduce the influence of the feedback current harmonics on the current regulator. Since the actual bandwidth of the current loop is equal to the smaller value between the design value of the current loop bandwidth ω * c and the resonant frequency ω h of the resonant filter. At the same time, set the filter resonant frequency to a larger value between 6ω e (sixth harmonic current frequency) and the design value of the current loop bandwidth (ω * c ), as shown in Figure 8, when the motor runs in the base speed region , the filter resonant frequency is ω h* c >6ω e , the actual current loop bandwidth is equal to ω * c , the filter does not affect the current loop bandwidth; after entering the field weakening region, the filter resonance frequency is 6ω eh* c , so the actual current loop bandwidth is equal to ω * c , the filter still does not affect the current loop bandwidth, the actual bandwidth value of the current loop is always equal to the bandwidth design value ω * c , so that the resonant filter can filter out harmonics at the same time. Does not affect the effect of the actual current loop bandwidth. Figure 8 and Figure 9 show the block diagram and transfer function Bode plot of the designed resonant filter based on a generalized second-order integrator. It can be seen that the speed-adaptive feedback current resonant filter in the present invention does not require motor parameters and has high robustness. phase delay.

步骤五:随着转速ωr继续升高,由于最大转差率限制,感应电机不能保持最大电流,进入恒电压区(弱磁二区)。isd,ref减小,isq,ref也减小,系统此时只受到电压与转差约束,并且dq轴电压之间满足

Figure BDA0001893398970000091
电机将在该运行区运行至给定转速。由于在弱磁区电机始终处于六拍运行模式,整个弱磁区基波电压与输出转矩都得到了提升。Step 5: As the rotational speed ω r continues to increase, due to the limitation of the maximum slip, the induction motor cannot maintain the maximum current and enters the constant voltage region (the second region of field weakening). i sd,ref decreases, i sq,ref also decreases, the system is only constrained by voltage and slip at this time, and the dq axis voltage satisfies
Figure BDA0001893398970000091
The motor will run to the given speed in this operating zone. Since the motor is always in the six-beat operation mode in the field weakening area, the fundamental voltage and output torque of the entire field weakening area have been improved.

实验效果:根据图10,六拍运行下三倍基速阶跃升速时间由内切圆电压下的0.796s缩小至六拍的0.695s,说明转矩输出变大,验证了本发明的有效性。对比转矩与A相定子电流波形可知在转矩提升的同时转矩与定子电流的波动变大。说明过调制产生的电压畸变造成了相电流畸变。图11为利用本发明算法在100%,200%,300%,400%额定转速下输出转矩情况,图12为相应转速下输出转矩频谱。根据图12中转矩频谱分析可知转矩脉动只在弱磁区出现,并且六拍运行造成的转矩脉动主要为6倍基频的转矩脉动。Experimental effect: According to Figure 10, the step-up time of triple base speed under six-beat operation is reduced from 0.796s under the inscribed circle voltage to 0.695s under six-beat operation, indicating that the torque output becomes larger, which verifies the effectiveness of the present invention. sex. Comparing the torque and A-phase stator current waveforms, it can be seen that the torque and stator current fluctuations become larger when the torque is increased. It shows that the voltage distortion caused by overmodulation causes the phase current distortion. Figure 11 shows the output torque at 100%, 200%, 300%, and 400% rated speed using the algorithm of the present invention, and Figure 12 shows the output torque spectrum at the corresponding speed. According to the torque spectrum analysis in Figure 12, it can be seen that the torque ripple only occurs in the field weakening region, and the torque ripple caused by the six-beat operation is mainly the torque ripple of 6 times the fundamental frequency.

根据图13,负载转矩由空载增至20%、40%、60%的额定负载时,转矩电流与定子电流随负载的增大而增大。当转矩增至20%与40%时,传统内切圆电压与提出的六拍电压下系统均可以跟随给定转速,但当负载增至60%的额定转矩时,内切圆电压输出下转速开始无法跟随给定,转速下降直至系统进入一个新的稳态。相比之下,六拍运行下转速仍能跟随给定,说明六拍运行下系统带载能力更强,验证了本发明的有效性。According to Figure 13, when the load torque increases from no-load to 20%, 40%, and 60% of the rated load, the torque current and stator current increase with the increase of the load. When the torque increases to 20% and 40%, the system can follow the given speed under both the traditional inscribed circle voltage and the proposed six-beat voltage, but when the load increases to 60% of the rated torque, the inscribed circle voltage output The lower speed begins to fail to follow the reference, and the speed drops until the system enters a new steady state. In contrast, the rotational speed can still follow the given value under the six-beat operation, indicating that the system has a stronger load capacity under the six-beat operation, which verifies the effectiveness of the present invention.

结合图14,对反馈电流滤波前后进行谐波分析,可以看出谐波电流主要是6次谐波(同步转速为150Hz对应900Hz),并且谐振滤波器可以有效滤除六次谐波,验证了本发明中的谐振滤波器的有效性。根据图15所示的电流调节器输出电压矢量轨迹可知,在没有反馈电流滤波器时候,电流调节器受同步旋转坐标系下反馈电流6次谐波影响而输出包含6次谐波的给定电压,变换至两相静止坐标系中后转换为5次,7次谐波,使得电压矢量轨迹变为六边形。该六边形电压造成过调算法中调制度参数的不稳定,最终造成六拍运行无法稳定,如图15中无滤波器的六拍输出电压时域波形。相比之下,在加入反馈电流滤波器后,由于几乎没有电流谐波进入电流调节器,调节器输出给定电压恢复至圆形,使得调制度稳定,进而过调制过程变得稳定,最终输出稳定的六拍电压,进一步验证了本发明中的谐振滤波器的有效性。Combined with Figure 14, the harmonic analysis before and after filtering the feedback current shows that the harmonic current is mainly the 6th harmonic (the synchronous speed is 150Hz corresponding to 900Hz), and the resonant filter can effectively filter out the 6th harmonic, which verifies Effectiveness of the resonant filter in the present invention. According to the output voltage vector trace of the current regulator shown in Figure 15, when there is no feedback current filter, the current regulator is affected by the 6th harmonic of the feedback current in the synchronous rotating coordinate system and outputs a given voltage including the 6th harmonic , transformed into the two-phase static coordinate system and then converted into 5th and 7th harmonics, so that the voltage vector trajectory becomes a hexagon. The hexagonal voltage causes the instability of the modulation parameter in the overshoot algorithm, and finally causes the six-beat operation to be unstable, as shown in the time-domain waveform of the six-beat output voltage without a filter in Figure 15. In contrast, after adding the feedback current filter, since almost no current harmonics enter the current regulator, the regulator output given voltage returns to a circle, which makes the modulation degree stable, and then the overmodulation process becomes stable, and the final output The stable six-beat voltage further verifies the effectiveness of the resonant filter in the present invention.

虽然本发明已以较佳的实施例公开如上,但其并非用以限定本发明,任何熟悉此技术的人,在不脱离本发明的精神和范围内,都可以做各种改动和修饰,因此本发明的保护范围应该以权利要求书所界定的为准。Although the present invention has been disclosed above with preferred embodiments, it is not intended to limit the present invention. Anyone who is familiar with this technology can make various changes and modifications without departing from the spirit and scope of the present invention. Therefore, The protection scope of the present invention should be defined by the claims.

Claims (3)

1. A weak magnetic control method for an induction motor based on optimized six-beat operation is characterized by comprising the following steps:
first of allStep, setting the given voltage value in the flux-weakening controller to be 2UdcThe feedback current resonant filter is built based on the generalized second-order integrator;
secondly, filtering the feedback current with the sixth harmonic by using the feedback current resonant filter, and generating a command voltage by using a complex vector current regulator after the difference is made between the feedback current and a current set value;
step three: secondly, the instruction voltage generated by the complex vector current regulator enters an SVPWM module to generate an inverter trigger pulse on one hand, and on the other hand, the instruction voltage is used as a feedback signal of the flux weakening first-zone controller to complete flux weakening control according to a voltage constraint equation;
step four, according to the weak magnetic control process in step three, the exciting current component produced by the weak magnetic controller gives isd,refThen, a torque current component given i is generated according to the current constraintsq,ref(ii) a Wherein isd,refAnd isq,refRespectively representing the d-axis and q-axis given currents of the stator;
step five: d-axis command voltage output by the complex vector current regulator is used as a flux weakening two-zone controller feedback signal as the rotating speed continues to rise, and dq-axis command voltage is satisfied
Figure FDA0001893398960000011
u* sd、u* sqThe dq-axis command voltages are respectively applied until a given rotation speed is reached.
2. The weak magnetic control method of the induction motor based on the optimized six-beat operation is characterized in that the step one of building a feedback current resonant filter based on the generalized second-order integrator comprises the following steps of:
will resonate at a frequency omegahSet to ω* cAnd 6. omegaeThe larger value in between, namely: e.g. omega* cAnd 6. omegaeThe larger value therebetween is ω* cThen the resonant frequency ωhSet to ω* cE.g. omega* cAnd 6. omegaeThe larger value between is 6. omegaeThen the resonant frequency ωhIs set to be 6 omegaeWherein, ω is* cFor the design value of the current loop bandwidth, ωeIs an instantaneous synchronous angular frequency and is 6 · ω in the fundamental regione* cFilter resonant frequency omegah=ω* cThe voltage of the motor stator is a standard sine wave, no sixth harmonic exists in the feedback stator current, and the resonant filter does not work.
3. The weak magnetic control method of the induction motor based on the optimized six-beat operation is characterized in that the process of generating the command voltage in the second step is as follows:
the detailed process of the control method comprises the following steps: when the induction motor enters a weak magnetic area along with the increase of the rotating speed, the weak magnetic control of the induction motor is started; according to the condition that the inverter outputs six-beat voltage in the weak magnetic region and generates a large amount of sixth harmonic in the stator current of the motor, the six-order current harmonic frequency in the weak magnetic region is 6 omegae* cSetting the resonant frequency omega of the resonant filterh=6·ωe(ii) a And attenuating the sixth harmonic through a resonant filter to obtain a current signal only with a fundamental current component, inputting the fundamental current component into a current closed-loop controller, and entering a complex vector current regulator to generate an instruction voltage after making a difference with a current given value.
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