CN111181463B - Built-in permanent magnet synchronous motor controller and control method thereof - Google Patents
Built-in permanent magnet synchronous motor controller and control method thereof Download PDFInfo
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- CN111181463B CN111181463B CN202010010461.2A CN202010010461A CN111181463B CN 111181463 B CN111181463 B CN 111181463B CN 202010010461 A CN202010010461 A CN 202010010461A CN 111181463 B CN111181463 B CN 111181463B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
- H02P21/0017—Model reference adaptation, e.g. MRAS or MRAC, useful for control or parameter estimation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/28—Arrangements for controlling current
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/34—Modelling or simulation for control purposes
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Abstract
The invention provides a controller of an internal permanent magnet synchronous motor and a control method thereof, wherein three-phase stator current signals, electric angular speed and electric angle of the internal permanent magnet synchronous motor are firstly collected, then a stator current corresponding to a given rotating speed is obtained by using an active disturbance rejection controller and an extended state observer based on a maximum torque current ratio control principle, then an optimal control voltage is obtained through a prediction model of the internal permanent magnet synchronous motor, and the method can obtain a corresponding duty ratio for each bridge arm of an inverter, and finally the rotating speed control of the permanent magnet synchronous motor is realized by applying the duty ratio on each bridge arm to the inverter. The invention can well estimate and compensate disturbance by using active disturbance rejection control, and the adopted current loop control method not only can rapidly and accurately track given current, but also can enable the inverter to generate fixed switching frequency, has smaller current fluctuation, and can meet the high-performance control requirement of the built-in permanent magnet synchronous motor.
Description
Technical Field
The invention relates to the technical field of permanent magnet synchronous motor control, in particular to a built-in permanent magnet synchronous motor controller combining Active Disturbance Rejection Control (ADRC) and Finite Control Set Model Predictive Control (FCSMPC) and a control method thereof.
Background
At present, a clean, efficient and safe electric automobile becomes an important direction for the development of the twenty-first-year automobile due to the problems of serious pollution and energy shortage caused by the large emission of automobile tail gas and the consumption of petroleum. Whichever electric vehicle is, its power performance and cruising ability are determined to a great extent by the driving motor. Because of the advantages of high efficiency, high power density, high reliability and the like, the permanent magnet synchronous motor has become a main stream driving motor of the current electric automobile, in particular to an interior permanent magnet synchronous motor (IPSM), and the unique rotor structure and electrical characteristics of the permanent magnet synchronous motor enable the motor to have large starting and braking torque, have good weak magnetic performance, can realize wide-range smooth speed regulation, and have light weight and small volume, thereby being an ideal driving motor of the electric automobile.
The running condition of the electric automobile is complex, so that high requirements are put on the speed regulation range, dynamic response, robustness and the like of the driving motor. The conventional PI control method cannot meet the control performance requirement. In order to obtain better performance, many methods have been proposed, such as back-step control, adaptive control, robust control, sliding mode control, fuzzy control, etc., which all improve the control performance of the motor from different aspects.
In recent years, the computing power of the digital controller is greatly improved due to the rapid development of the microprocessor, so that a model predictive control algorithm can be realized in motor control. In order for an internal permanent magnet synchronous motor to have a fast torque response and high performance action, a high performance current controller is required. The predictive current control exhibits better performance than conventional vector control and torque control, and can provide high dynamic performance and low current harmonics to ensure the quality of torque and speed control. The traditional model predictive current control is designed based on a limited control set, can directly generate a switching signal acting on an inverter, and introduces variable switching frequency although a modulator is omitted, so that current fluctuation is large in a steady state. The control mode still needs to be matched with a rotating speed ring to generate a reference rotating speed, and the conventional rotating speed ring adopts PI control, so that the rapidity and the robustness of the rotating speed ring are improved.
Therefore, in order to obtain better control performance, improvements in the control methods of the speed loop and the current loop are required. The addition of Active Disturbance Rejection Control (ADRC) to the speed loop and PWM predictive control to the current loop is a good improvement idea.
Disclosure of Invention
The invention mainly aims to provide an ADRC and FCSMPC combined built-in permanent magnet synchronous motor controller and a control method thereof, wherein the ADRC-based speed loop control method can accelerate response speed and play an anti-interference role on load change, and the PWM-based prediction control-based limited control set model prediction method can rapidly track given current, has small current fluctuation in a steady state and generates fixed switching frequency.
In order to achieve the above object, the present invention provides a built-in permanent magnet synchronous motor controller, comprising: the system comprises an ADRC controller, a maximum torque current ratio controller, a limited control set model prediction controller based on PWM prediction control, a rotating speed and angle calculation unit, a Clark conversion unit, a Park conversion unit and a voltage source inverter;
the rotating speed and angle calculating unit is used for obtaining the electric angular speed and the electric angle of the built-in permanent magnet synchronous motor and calculating the electric angular speed and the electric angle of the built-in permanent magnet synchronous motor by using the built-in permanent magnet synchronous motor;
the Clark conversion unit is used for performing Clark conversion on three-phase stator current signals of the built-in permanent magnet synchronous motor to obtain stator current under an alpha beta coordinate system;
the Park conversion unit is used for carrying out Park conversion on the stator current under the alpha beta coordinate system to obtain the stator current under the synchronous rotation dq coordinate system;
the ADRC controller is used for determining a reference stator current through a second-order expansion state observer and a first-order nonlinear control law;
the maximum torque current ratio controller is used for ensuring that the generated d and q axis reference voltages have maximum torque;
the limited control set model prediction controller based on PWM prediction control is used for generating PWM control signals with fixed switching frequency and ensuring that the generated current fluctuation is small;
the voltage source inverter is used for controlling three-phase stator current according to the PWM control signal so as to control the rotating speed of the built-in permanent magnet synchronous motor.
The invention also provides a control method of the built-in permanent magnet synchronous motor controller, which comprises the following steps:
s10, performing Clark conversion and Park conversion on three-phase stator current signals of the built-in permanent magnet synchronous motor to obtain stator currents under synchronous rotation coordinates;
s20, establishing a prediction model, taking stator current under synchronous rotation coordinates, the electric angular speed and a given rotating speed of the built-in permanent magnet synchronous motor as input of a speed loop controller based on an active disturbance rejection controller, and obtaining reference current of a current loop by adopting a maximum torque current control method;
s30, generating PWM signals which enable each bridge arm of the inverter to have different duty ratios by adopting a limited control set model predictive control method based on PWM predictive control;
s40, the PWM signals are acted on the inverter, so that three-phase stator currents are controlled through the inverter, and the built-in permanent magnet synchronous motor can control the rotating speed.
Preferably, before step S10, the method further includes:
s00, collecting three-phase definition current signals, electrical angular velocity and electrical angle of the built-in permanent magnet synchronous motor;
the step S00 specifically comprises the following steps:
sampling three-phase stator current signal i with sensor during operation of built-in permanent magnet synchronous motor for electric automobile a 、i b 、i c The method comprises the steps of carrying out a first treatment on the surface of the And obtaining the electric angular velocity omega and the electric angle theta of the built-in permanent magnet synchronous motor by adopting a position-sensor-free detection technology.
Preferably, step S20 specifically includes:
s201, selecting a state variable and establishing a prediction model;
s202, designing a 2-order expansion state observer according to a rotational speed differential equation in a built-in permanent magnet synchronous motor model to estimate rotational speed and load disturbance;
s203, obtaining a reference stator current by using a 1-order nonlinear state error feedback rate according to a given rotating speed, an estimated rotating speed and an estimated disturbance;
s204, determining reference values of d and q axis currents according to a maximum torque current ratio method.
Preferably, step S201 is specifically:
with d, q-axis current i d 、i q And the electric angular velocity omega is a state variable, and a mathematical model of the built-in permanent magnet synchronous motor under a d and q synchronous rotation coordinate system is established:
wherein L is d Represents the direct axis inductance, L q Represents the quadrature axis inductance, R s Represents stator resistance, ψ f Representing rotor flux linkage, p n Represents the pole pair number, J represents the moment of inertia, B represents the viscous friction factor, u d 、u q The control voltages on the d and q axes are shown, respectively.
Preferably, step S202 is specifically:
writing the third formula of formula (1) into the following form
In the method, in the process of the invention,
u is a control amount;
the second-order extended state observer corresponding to the formula (2) is
Moderately selected parameters allow a good estimation of the state w and the load disturbance f, i.e. z 1 Is an estimate of w, z 2 Is an estimate of f.
Preferably, step S203 is specifically:
the 1-order nonlinear state error feedback control law is selected as follows:
u o (t)＝βfal(ω ref -ω,g 2 ,d 2 ) (21)
to eliminate the interference, the actual control amount u is taken as:
the u obtained here is taken as the stator voltage i s Reference value i of (2) sRef ；
The step S204 specifically includes:
according to the obtained i sRef Calculating d, q-axis current reference values according to equation (6):
preferably, step S30 specifically includes:
s301, obtaining expected d and q axis control voltages according to reference d and q axis currents and a prediction model;
s302, converting the voltage expected to be obtained into an inverter switching signal duty ratio expected to be obtained;
s303, obtaining the duty ratio of each bridge arm of the inverter in one adoption period through an inverter switch duty ratio selection algorithm.
Preferably, step S301 specifically includes:
the first order Euler dispersion of the current portion in equation (1) is available:
for the inverter switching variable, the following relationship exists between it and dq axis current:
in the method, in the process of the invention,
u a ,u b ,u c the value of (2) can only be 1 or 0, and represents the on-off state of the corresponding bridge arm switch of the inverter, EIs the voltage of a direct current bus;
let ρ be λ (k) Is the duty cycle of the lambda-th leg in each period (lambda epsilon a, b, c),
defined as the average voltage of the d, q axes of this period, the following relationship holds:
defining a duty cycle matrix asρ(k)＝[ρ a (k),ρ b (k),ρ c (k)] T Average voltage of d and q axes in one period
Can be expressed by the following formula
Knowing the d, q-axis reference current, the required d, q-axis voltages to track this current over a period can be expressed as:
preferably, step S302 is specifically:
substituting formula (10) into formula (7) includes:
X(k+1)＝F(k)·X(k)+G·M(k)·D·ρ(k)+H(k) (29)
formula (11) is substituted into formula (12) and includes:
D·ρ(k)＝M -1 (k)·G -1 ·(X # -F(k)·X(k)-H(k)) (30)
formula (8) is substituted into formula (13) and developed:
three switching signals ρ of equation (14) a 、ρ b 、ρ c To obtain a certain solution, additional conditions need to be added, otherwise there are numerous solutions, where the sum of the two minimum and maximum values is chosen to be 1, namely (14) to the form:
equation (15) corresponds to a hidden condition,
ρ b +ρ c ＝1,ρ b ＞ρ a ＞ρ c orρ c ＞ρ a ＞ρ b (33)
the equation (15) can be solved, and the required duty ratio matrix can be obtained as long as the solution obtained satisfies the equation (16), otherwise the solution is expressed as ρ a +ρ b =1 or ρ a +ρ c Reelect equation (15) until the set constraint is correctly satisfied, =1;
step S303 specifically includes:
in order to reduce the amount of on-line calculation, after the result is calculated in equation (15), the final PWM signal is obtained by performing calculation using the following equation:
novel duty cycle matrix [ ρ ]' a ,ρ′ b ,ρ′ c ] T The PWM signal can be directly applied to an inverter to realize the rotation speed control.
The invention provides an ADRC and FCSMPC combined built-in permanent magnet synchronous motor controller and a control method thereof, and compared with the prior art, the invention has the following beneficial effects:
(1) The ground active disturbance rejection controller comprises an extended state observer and a nonlinear control rate, so that disturbance can be well estimated, response speed is increased, and disturbance rejection is realized on load change;
(2) Maximum torque current ratio control is adopted, so that the efficiency of the motor for outputting torque is maximized;
(3) The current loop adopts a limited control set model predictive control method based on PWM predictive control, can generate a switching signal with fixed frequency, can rapidly track given current, and has small current fluctuation in a steady state.
Drawings
In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings that are required in the embodiments or the description of the prior art will be briefly described, and it is obvious that the drawings in the following description are only some embodiments of the present invention, and other drawings may be obtained according to the structures shown in these drawings without inventive effort for a person skilled in the art.
FIG. 1 is a block diagram of a control architecture of the present invention;
FIG. 2 is a graph showing the comparison of the rotational speed response of the built-in permanent magnet synchronous motor under the predictive control method of the traditional finite control set model and the traditional cascade PI control method;
FIG. 3 shows the d-axis current i of the permanent magnet synchronous motor under the predictive control of the present model predictive control method and the conventional finite control set model predictive control d A rotating speed response comparison chart;
FIG. 4 shows the q-axis current i of the permanent magnet synchronous motor under the predictive control of the present model predictive control method and the conventional finite control set model predictive control q A rotating speed response comparison chart;
the achievement of the objects, functional features and advantages of the present invention will be further described with reference to the accompanying drawings, in conjunction with the embodiments.
Detailed Description
The following description of the embodiments of the present invention will be made clearly and fully with reference to the accompanying drawings, in which it is evident that the embodiments described are only some, but not all embodiments of the invention. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.
It should be noted that, if directional indications (such as up, down, left, right, front, and rear … …) are included in the embodiments of the present invention, the directional indications are merely used to explain the relative positional relationship, movement conditions, etc. between the components in a specific posture (as shown in the drawings), and if the specific posture is changed, the directional indications are correspondingly changed.
In addition, if there is a description of "first", "second", etc. in the embodiments of the present invention, the description of "first", "second", etc. is for descriptive purposes only and is not to be construed as indicating or implying a relative importance or implicitly indicating the number of technical features indicated. Thus, a feature defining "a first" or "a second" may explicitly or implicitly include at least one such feature. In addition, the technical solutions of the embodiments may be combined with each other, but it is necessary to base that the technical solutions can be realized by those skilled in the art, and when the technical solutions are contradictory or cannot be realized, the combination of the technical solutions should be considered to be absent and not within the scope of protection claimed in the present invention.
The term "Clark transformation" in the present invention means the transformation from a three-phase to a two-phase stationary coordinate system, belonging to a generic term; "Park transformation" means the transformation from a two-phase stationary coordinate system to a rotating coordinate system, and belongs to the generic term; "SVPWM" means space vector pulse width modulation, which is a generic term; "PWM" means pulse width modulation, which is a generic term.
Example 1
The invention provides a built-in permanent magnet synchronous motor controller, which comprises: the system comprises an ADRC controller, a maximum torque current ratio controller, a limited control set model prediction controller based on PWM prediction control, a rotating speed and angle calculation unit, a Clark conversion unit, a Park conversion unit and a voltage source inverter;
the rotating speed and angle calculating unit is used for obtaining the electric angular speed and the electric angle of the built-in permanent magnet synchronous motor and calculating the electric angular speed and the electric angle of the built-in permanent magnet synchronous motor by using the built-in permanent magnet synchronous motor;
the Clark conversion unit is used for performing Clark conversion on three-phase stator current signals of the built-in permanent magnet synchronous motor to obtain stator current under an alpha beta coordinate system;
the Park conversion unit is used for carrying out Park conversion on the stator current under the alpha beta coordinate system to obtain the stator current under the synchronous rotation dq coordinate system;
the ADRC controller is used for determining a reference stator current through a second-order expansion state observer and a first-order nonlinear control law;
the maximum torque current ratio controller is used for ensuring that the generated d and q axis reference voltages have maximum torque;
the limited control set model prediction controller based on PWM prediction control is used for generating PWM control signals with fixed switching frequency and ensuring that the generated current fluctuation is small;
the voltage source inverter is used for controlling three-phase stator current according to the PWM control signal so as to control the rotating speed of the built-in permanent magnet synchronous motor.
The controller comprises a speed loop and a current loop, wherein the speed loop adopts an active disturbance rejection control method, and the current loop adopts a limited control centralized control method and directly generates a switching signal of an inverter.
Example 2
The invention provides a control method of a built-in permanent magnet synchronous motor controller combining ADRC and FCSMPC; in a preferred embodiment of the present invention, a specific description is made of a controller portion of an internal permanent magnet synchronous motor, as shown in fig. 1, including the following steps:
s00, collecting three-phase definition current signal i of built-in permanent magnet synchronous motor a 、i b 、i c An electrical angular velocity ω and an electrical angle θ.
In the embodiment of the invention, when the built-in permanent magnet synchronous motor for the electric automobile runs, a sensor is used for sampling a three-phase stator current signal i a 、i b 、i c The method comprises the steps of carrying out a first treatment on the surface of the Electric angular velocity of built-in permanent magnet synchronous motor obtained by adopting position-sensor-free detection technologyDegree ω and electrical angle θ;
s10, performing Clark conversion and Park conversion on three-phase stator current signals of the built-in permanent magnet synchronous motor to obtain stator currents under synchronous rotation coordinates;
s20, establishing a prediction model, taking stator current under synchronous rotation coordinates, the electric angular speed and a given rotating speed of the built-in permanent magnet synchronous motor as input of a speed loop controller based on an active disturbance rejection controller, and obtaining reference current of a current loop by adopting a maximum torque current control (MTPA) method;
in a preferred embodiment of the present invention, S20 is specifically:
s201, selecting a state variable and establishing a prediction model;
the method comprises the following steps:
with d, q-axis current i d 、i q And the electric angular velocity omega is a state variable, and a mathematical model of the built-in permanent magnet synchronous motor under a d and q synchronous rotation coordinate system is established:
wherein L is d Represents the direct axis inductance, L q Represents the quadrature axis inductance, R s Represents stator resistance, ψ f Representing rotor flux linkage, p n Represents the pole pair number, J represents the moment of inertia, B represents the viscous friction factor, u d 、u q Control voltages of d and q axes are respectively represented;
s202, designing a 2-order expansion state observer according to a rotational speed differential equation in a built-in permanent magnet synchronous motor model to estimate rotational speed and load disturbance;
the method comprises the following steps:
writing the third formula of formula (1) into the following form
In the method, in the process of the invention,
u is the control amount.
The second-order extended state observer corresponding to the formula (2) is
Moderately selected parameters allow a good estimation of the state w and the load disturbance f, i.e. z 1 Is an estimate of w, z 2 Is an estimate of f.
S203, obtaining a reference stator current by using a 1-order nonlinear state error feedback rate according to a given rotating speed, an estimated rotating speed and an estimated disturbance;
the method comprises the following steps:
the 1-order nonlinear state error feedback control law is selected as follows:
u o (t)＝βfal(ω ref -ω,g 2 ,d 2 ) (38)
to eliminate the interference, the actual control amount u is taken as:
the u obtained here can be regarded as the stator voltage i s Reference value i of (2) sRef 。
Preferably, the reference values of the d and q-axis currents are determined according to the maximum torque current ratio method as described in S204.
The method comprises the following steps:
according to the obtained i sRef Calculating d, q-axis current reference values according to equation (6):
s30, generating PWM signals which enable each bridge arm of the inverter to have different duty ratios by adopting a limited control set model predictive control method based on PWM predictive control;
in a preferred embodiment of the present invention, S30 is specifically S301, where desired d-axis and q-axis control voltages are obtained according to reference d-axis and q-axis currents and a prediction model;
the method comprises the following steps:
the first order Euler dispersion of the current portion in equation (1) is available:
for the inverter switching variable, the following relationship exists between it and dq axis current:
in the method, in the process of the invention,
u a ,u b ,u c the value of (2) can only be 1 or 0, the value represents the on-off state of the bridge arm switch corresponding to the inverter, and E is the voltage of the direct current bus.
Let ρ be λ (k) Is the duty cycle of the lambda-th leg in each period (lambda epsilon a, b, c),
defined as the average voltage of the d, q axes of this period, the following relationship holds:
defining a duty cycle matrix asρ(k)＝[ρ a (k),ρ b (k),ρ c (k)] T Average voltage of d and q axes in one period
Can be expressed by the following formula
Knowing the d, q-axis reference current, the required d, q-axis voltages to track this current over a period can be expressed as:
s302, converting the voltage expected to be obtained into an inverter switching signal duty ratio expected to be obtained;
the method comprises the following steps:
substituting formula (10) into formula (7) includes:
X(k+1)＝F(k)·X(k)+G·M(k)·D·ρ(k)+H(k) (46)
formula (11) is substituted into formula (12) and includes:
D·ρ(k)＝M -1 (k)·G -1 ·(X # -F(k)·X(k)-H(k)) (47)
formula (8) is substituted into formula (13) and developed:
three switching signals ρ of equation (14) a 、ρ b 、ρ c To obtain a certain solution, additional conditions need to be added, otherwise there are numerous solutions, where the sum of the two minimum and maximum values is chosen to be 1, namely (14) to the form:
equation (15) corresponds to a hidden condition,
ρ b +ρ c ＝1,ρ b ＞ρ a ＞ρ c orρ c ＞ρ a ＞ρ b (50)
the equation (15) can be solved, as long as the solution is obtainedIf equation (16) is satisfied, the required duty cycle matrix can be found, otherwise, ρ is a +ρ b =1 or ρ a +ρ c And =1 re-choose equation (15) until the set constraint is correctly met.
S303, obtaining the duty ratio of each bridge arm of the inverter in one adoption period through an inverter switch duty ratio selection algorithm;
the method comprises the following steps:
in order to reduce the amount of on-line calculation, after the result is calculated in equation (15), the final PWM signal is obtained by performing calculation using the following equation:
novel duty cycle matrix [ ρ ]' a ,ρ′ b ,ρ′ c ] T The PWM signal can be directly applied to an inverter to realize the rotation speed control.
In a preferred embodiment of the invention, in order to verify the effectiveness of the ADRC and FCSMPC combined internal permanent magnet synchronous motor control method provided by the invention on the internal permanent magnet synchronous motor, the internal permanent magnet synchronous motor under the control method of the invention is compared and analyzed with the traditional PI regulator control method and the traditional rotating speed and current response curve under the prediction control of a limited control set model. A simulation model as shown in fig. 1 is built in the simulation. The simulation conditions are as follows: the system simulation time was 0.6s, the initial given rotational speed was 1000r/min, and at 0.4s it was reduced to 500r/min. The initial given load torque is 0n.m, and at 0.3s a load torque of 50n.m is added and remains unchanged.
FIG. 2 shows the present invention (labeled ADRC+FCS in the figure) and a conventional cascaded PI control method (labeled PI in the figure, based on i) d Control principle of =0), a comparison of the rotational speed response of the permanent magnet synchronous motor under a conventional finite control set model predictive control method (labeled DPC in the figure, based on the same maximum torque-to-current ratio control as herein), it can be seen that the maximum torque-to-current ratio control is adopted to generate a larger torque, so that the rise time is shorter,when load disturbance changes by adopting ADRC, the change rate of the rotating speed is smaller, and compared with DPC using the same rotating speed loop control method, the PWM prediction control-based finite control set model prediction control method can track the given rotating speed more quickly without overshoot. Demonstrating the superiority of the invention;
FIG. 3 shows the d-axis current i of an internal permanent magnet synchronous motor under the present invention (labeled ADRC+FCS in the figure) and conventional finite control set model predictive control (labeled DPC in the figure, based on the same maximum torque to current ratio control as herein) d And (5) a rotating speed response comparison chart. FIG. 4 shows the q-axis current i of an internal permanent magnet synchronous motor under the present invention (labeled ADRC+FCS in the figure) and conventional finite control set model predictive control (labeled DPC in the figure, based on the same maximum torque to current ratio control as herein) q And (5) a rotating speed response comparison chart. Compared with the traditional finite control set model predictive control method, the finite control set model predictive control method based on PWM predictive control can track a given rotating speed more quickly and accurately, and obviously, current fluctuation in a steady state is greatly reduced; the feasibility of the method is proved again, and the method is further explained to realize the control of the high-performance built-in permanent magnet synchronous motor.
The foregoing description is only of the preferred embodiments of the present invention and is not intended to limit the scope of the invention, and all equivalent structural changes made by the description of the present invention and the accompanying drawings or direct/indirect application in other related technical fields are included in the scope of the invention.
Claims (6)
1. A built-in permanent magnet synchronous motor controller, comprising: the system comprises an ADRC controller, a maximum torque current ratio controller, a limited control set model prediction controller based on PWM prediction control, a rotating speed and angle calculation unit, a Clark conversion unit, a Park conversion unit and a voltage source inverter;
the rotating speed and angle calculating unit is used for obtaining the electric angular speed and the electric angle of the built-in permanent magnet synchronous motor and calculating the electric angular speed and the electric angle of the built-in permanent magnet synchronous motor by using the built-in permanent magnet synchronous motor;
the Clark conversion unit is used for performing Clark conversion on three-phase stator current signals of the built-in permanent magnet synchronous motor to obtain stator current under an alpha beta coordinate system;
the Park conversion unit is used for carrying out Park conversion on the stator current under the alpha beta coordinate system to obtain the stator current under the synchronous rotation dq coordinate system;
the ADRC controller is used for determining a reference stator current through a second-order expansion state observer and a first-order nonlinear control law;
the maximum torque current ratio controller is used for ensuring that the generated d and q axis reference voltages have maximum torque;
the limited control set model prediction controller based on PWM prediction control is used for generating PWM control signals with fixed switching frequency and ensuring that the generated current fluctuation is small;
with d, q-axis current i d 、i q And the electric angular velocity omega is a state variable, and a mathematical model of the built-in permanent magnet synchronous motor under a d and q synchronous rotation coordinate system is established:
wherein L is d Represents the direct axis inductance, L q Represents the quadrature axis inductance, R s Represents stator resistance, ψ f Representing rotor flux linkage, p n Represents the pole pair number, J represents the moment of inertia, B represents the viscous friction factor, u d 、u q Control voltages of d and q axes are respectively represented;
the finite control set model predictive controller based on PWM predictive control is used for generating PWM control signals with fixed switching frequency and ensuring small generated current fluctuation, and specifically comprises the following steps:
s301, obtaining expected d and q axis control voltages according to reference d and q axis currents and a prediction model;
s302, converting the voltage expected to be obtained into an inverter switching signal duty ratio expected to be obtained;
s303, obtaining the duty ratio of each bridge arm of the inverter in one adoption period through an inverter switch duty ratio selection algorithm;
the step S301 specifically includes:
the first order Euler dispersion of the current portion in equation (1) is available:
for the inverter switching variable, the following relationship exists between it and dq axis current:
in the method, in the process of the invention,
u a ,u b ,u c the value of (2) can only be 1 or 0, the value represents the on-off state of a bridge arm switch corresponding to the inverter, and E is the voltage of a direct current bus;
let ρ be λ (k) Is the duty cycle of the lambda-th leg in each period, where lambda e a, b, c,
defined as the average voltage of the d, q axes of this period, the following relationship holds:
defining a duty cycle matrix asρ(k)＝[ρ a (k),ρ b (k),ρ c (k)] T Average voltage of d and q axes in one period
Can be expressed by the following formula
Knowing the d, q-axis reference current, the required d, q-axis voltages to track this current over a period can be expressed as:
the step S302 specifically includes:
substituting formula (10) into formula (7) includes:
X(k+1)＝F(k)·X(k)+G·M(k)·D·ρ(k)+H(k) (12)
formula (11) is substituted into formula (12) and includes:
D·ρ(k)＝M -1 (k)·G -1 ·(X # -F(k)·X(k)-H(k)) (13)
formula (8) is substituted into formula (13) and developed:
three switching signals ρ of equation (14) a 、ρ b 、ρ c To obtain a certain solution, additional conditions need to be added, otherwise there are numerous solutions, where the sum of the two minimum and maximum values is chosen to be 1, namely (14) to the form:
equation (15) corresponds to a hidden condition,
ρ b +ρ c ＝1,ρ b ＞ρ a ＞ρ c orρ c ＞ρ a ＞ρ b (16)
the equation (15) can be solved, and the required duty ratio matrix can be obtained as long as the solution obtained satisfies the equation (16), otherwise the solution is expressed as ρ a +ρ b =1 or ρ a +ρ c Reelect equation (15) until the set constraint is correctly satisfied, =1;
step S303 specifically includes:
in order to reduce the amount of on-line calculation, after the result is calculated in equation (15), the final PWM signal is obtained by performing calculation using the following equation:
novel duty cycle matrix [ ρ ]' a ,ρ' b ,ρ' c ] T The PWM signal can be directly applied to an inverter to realize the rotation speed control;
the voltage source inverter is used for controlling three-phase stator current according to the PWM control signal so as to control the rotating speed of the built-in permanent magnet synchronous motor.
2. The control method of the built-in permanent magnet synchronous motor controller is characterized by comprising the following steps of:
s10, performing Clark conversion and Park conversion on three-phase stator current signals of the built-in permanent magnet synchronous motor to obtain stator currents under synchronous rotation coordinates;
s20, establishing a prediction model, taking stator current under synchronous rotation coordinates, the electric angular speed and a given rotating speed of the built-in permanent magnet synchronous motor as input of a speed loop controller based on an active disturbance rejection controller, and obtaining reference current of a current loop by adopting a maximum torque current control method;
s30, generating PWM signals which enable each bridge arm of the inverter to have different duty ratios by adopting a limited control set model predictive control method based on PWM predictive control;
with d, q-axis current i d 、i q And the electric angular velocity omega is a state variable, and a mathematical model of the built-in permanent magnet synchronous motor under a d and q synchronous rotation coordinate system is established:
wherein L is d Representing the direct axis inductance, i q Represents the quadrature axis inductance, R s Represents stator resistance, ψ f Representing rotor flux linkage, p n Represents the pole pair number, J represents the moment of inertia, B represents the viscous friction factor, u d 、u q Control voltages of d and q axes are respectively represented;
the step S30 specifically includes:
s301, obtaining expected d and q axis control voltages according to reference d and q axis currents and a prediction model;
s302, converting the voltage expected to be obtained into an inverter switching signal duty ratio expected to be obtained;
s303, obtaining the duty ratio of each bridge arm of the inverter in one adoption period through an inverter switch duty ratio selection algorithm;
the step S301 specifically includes:
the first order Euler dispersion of the current portion in equation (1) is available:
for the inverter switching variable, the following relationship exists between it and dq axis current:
in the method, in the process of the invention,
u a ,u b ,u c the value of (2) can only be 1 or 0, the value represents the on-off state of a bridge arm switch corresponding to the inverter, and E is the voltage of a direct current bus;
let ρ be λ (k) Is the duty cycle of the lambda-th leg in each period, where lambda e a, b, c,
defined as the average voltage of the d, q axes of this period, the following relationship holds:
defining a duty cycle matrix asρ(k)＝[ρ a (k),ρ b (k),ρ c (k)] T Average voltage of d and q axes in one period
Can be expressed by the following formula
Knowing the d, q-axis reference current, the required d, q-axis voltages to track this current over a period can be expressed as:
the step S302 specifically includes:
substituting formula (10) into formula (7) includes:
X(k+1)＝F(k)·X(k)+G·M(k)·D·ρ(k)+H(k) (12)
formula (11) is substituted into formula (12) and includes:
D·ρ(k)＝M -1 (k)·G -1 ·(X # -F(k)·X(k)-H(k)) (13)
formula (8) is substituted into formula (13) and developed:
three switching signals ρ of equation (14) a 、ρ b 、ρ c To obtain a certain solution, additional conditions need to be added, otherwise there are numerous solutions, where the sum of the two minimum and maximum values is chosen to be 1, namely (14) to the form:
equation (15) corresponds to a hidden condition,
ρ b +ρ c ＝1,ρ b ＞ρ a ＞ρ c orρ c ＞ρ a ＞ρ b (16)
the equation (15) can be solved, and the required duty ratio matrix can be obtained as long as the solution obtained satisfies the equation (16), otherwise the solution is expressed as ρ a +ρ b =1 or ρ a +ρ c Reelect equation (15) until the set constraint is correctly satisfied, =1;
step S303 specifically includes:
in order to reduce the amount of on-line calculation, after the result is calculated in equation (15), the final PWM signal is obtained by performing calculation using the following equation:
novel duty cycle matrix [ ρ ]' a ,ρ' b ,ρ' c ] T The PWM signal can be directly applied to an inverter to realize the rotation speed control;
s40, the PWM signals are acted on the inverter, so that three-phase stator currents are controlled through the inverter, and the built-in permanent magnet synchronous motor can control the rotating speed.
3. The method for controlling a built-in permanent magnet synchronous motor controller according to claim 2, further comprising, prior to step S10:
s00, collecting three-phase definition current signals, electrical angular velocity and electrical angle of the built-in permanent magnet synchronous motor;
the step S00 specifically comprises the following steps:
sampling three-phase stator current signal i with sensor during operation of built-in permanent magnet synchronous motor for electric automobile a 、i b 、i c The method comprises the steps of carrying out a first treatment on the surface of the And obtaining the electric angular velocity omega and the electric angle theta of the built-in permanent magnet synchronous motor by adopting a position-sensor-free detection technology.
4. The method for controlling a built-in permanent magnet synchronous motor controller according to claim 2, wherein step S20 specifically includes:
s201, selecting a state variable and establishing a prediction model;
s202, designing a 2-order expansion state observer according to a rotational speed differential equation in a built-in permanent magnet synchronous motor model to estimate rotational speed and load disturbance;
s203, obtaining a reference stator current by using a 1-order nonlinear state error feedback rate according to a given rotating speed, an estimated rotating speed and an estimated disturbance;
s204, determining reference values of d and q axis currents according to a maximum torque current ratio method.
5. The method for controlling a built-in permanent magnet synchronous motor controller according to claim 2, wherein step S202 is specifically:
writing the third formula of formula (1) into the following form
In the method, in the process of the invention,
u is a control amount;
the second-order extended state observer corresponding to the formula (2) is
Moderately selected parameters allow a good estimation of the state w and the load disturbance f, i.e. z 1 Is an estimate of w, z 2 Is an estimate of f.
6. The method of claim 5, wherein step S203 is specifically:
the 1-order nonlinear state error feedback control law is selected as follows:
u o (t)＝βfal(ω ref -ω,g 2 ,d 2 ) (4)
to eliminate the interference, the actual control amount u is taken as:
the u obtained here is taken as the stator voltage i s Reference value i of (2) sRef ；
The step S204 specifically includes:
according to the obtained i sRef Calculating d, q-axis current reference values according to equation (6):
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| JP2013118746A (en) * | 2011-12-02 | 2013-06-13 | Denso Corp | Rotating machine control device |
| CN103312255A (en) * | 2013-06-18 | 2013-09-18 | 山东大学（威海） | Method and device for controlling speed of permanent-magnet synchronous motor |
| CN109639192A (en) * | 2018-12-12 | 2019-04-16 | 西安交通大学 | A kind of durface mounted permanent magnet synchronous motor time-sharing control system and Time-sharing control method |
| CN110022105A (en) * | 2019-04-25 | 2019-07-16 | 西安理工大学 | Permanent magnet synchronous motor predictive-current control method and system based on FOSMC |
| CN110165962A (en) * | 2019-05-15 | 2019-08-23 | 浙江工业大学 | A kind of direct-drive permanent magnetism synchronous wind generating system and its full Auto-disturbance-rejection Control |
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| JP2013118746A (en) * | 2011-12-02 | 2013-06-13 | Denso Corp | Rotating machine control device |
| CN103312255A (en) * | 2013-06-18 | 2013-09-18 | 山东大学（威海） | Method and device for controlling speed of permanent-magnet synchronous motor |
| CN109639192A (en) * | 2018-12-12 | 2019-04-16 | 西安交通大学 | A kind of durface mounted permanent magnet synchronous motor time-sharing control system and Time-sharing control method |
| CN110022105A (en) * | 2019-04-25 | 2019-07-16 | 西安理工大学 | Permanent magnet synchronous motor predictive-current control method and system based on FOSMC |
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