CN111030537A - Two-phase direct current offset vernier motor controller and control method - Google Patents

Two-phase direct current offset vernier motor controller and control method Download PDF

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CN111030537A
CN111030537A CN201911243357.1A CN201911243357A CN111030537A CN 111030537 A CN111030537 A CN 111030537A CN 201911243357 A CN201911243357 A CN 201911243357A CN 111030537 A CN111030537 A CN 111030537A
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phase
axis current
value
motor
current
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贾少锋
李斌珂
梁得亮
闫宽宽
刘进军
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Xian Jiaotong University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/08—Reluctance motors
    • H02P25/092—Converters specially adapted for controlling reluctance motors

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Abstract

The invention discloses a two-phase direct current offset sinusoidal current motor controller and a control method thereof. After topology optimization, the driving resources can be reduced to half, and meanwhile, the control signals required to be output by the control chip are reduced to half.

Description

Two-phase direct current offset vernier motor controller and control method
Technical Field
The invention belongs to an alternating current motor drive control device, and particularly relates to a two-phase direct current bias vernier motor controller and a control method, which can be used for drive control of a novel two-phase direct current bias vernier magnetic resistance and a vernier permanent magnet motor.
Background
The permanent magnet motor has the excellent characteristics of low speed and large torque, but the cost of the permanent magnet motor is high due to the higher price of the permanent magnet material, so that the application of the permanent magnet motor in industrial occasions is severely limited. Although the traditional switched reluctance motor has simple structure and low cost, the specific operation mode of the traditional switched reluctance motor determines that the noise and the vibration of the motor are large and the torque ripple is also large. These drawbacks affect the application of switched reluctance machines. In order to combine the advantages of both motors, three-phase dc-biased vernier reluctance motors have been proposed by researchers in recent years. However, the three-phase dc offset vernier motor requires too many power switches, which results in high cost of the driving and controlling units. The two-phase motor is shown in fig. 1, and includes a stator 01, a rotor 02, a winding 03, and other common structural members such as a rotating shaft, a casing, an end cover, and a position sensor. The motor can adopt a single-layer fractional-slot non-overlapping concentrated winding, the winding comprises A, B two phases, and each phase of the winding consists of two small fractional phases. The connection mode of the two small split phases is shown in figure 2. The winding current per phase is shown in fig. 3(a), 3(b) and 3 (c). When the DC bias sine current motor adopts the maximum torque current ratio control method, the amplitude of the DC excitation component needs to be kept equal to the amplitude of the AC component.
For such a dc offset motor in which each phase current includes two components, i.e., ac and dc, the existing controller generally adopts a single-phase full-bridge converter to control each phase current. The connection mode of one-phase winding of DC offset motor controller using single-phase full-bridge converter is shown in FIG. 4, and four turn-off power devices S11、S12、S13、S14The converter adopted by the controller occupies too many power electronic devices and resources of the controller.
Disclosure of Invention
The invention aims to provide a two-phase direct current bias vernier motor controller and a control method, which are used for solving the problems that a converter adopted by the existing controller occupies the number of power electronic devices and the controller resources are excessive.
In order to achieve the purpose, the invention adopts the following technical scheme:
the invention provides a two-phase direct current offset sinusoidal current motor controller which comprises a first subtracter, a second subtracter, a third subtracter, a fourth subtracter, a speed regulator, a d-axis current regulator, a q-axis current regulator, a 0-axis current regulator, a pulse width modulator, a current transformer, a motor position sensor, phase A and phase B current sensors and a current converter.
Given value n of motor speedrFeeding the measured value n of the motor speed into a first subtracter, nrThe difference value of-n is sent to a speed regulator, and the speed regulator outputs a q-axis current set value i* qBy adjusting iqSo that n isrThe difference of-n is always zero, i.e. n always follows nrThe d-axis current given value is 0, and the 0-axis current given value is equal to the q-axis current given value, so that the d-axis current given value i can be obtained* dQ-axis current given value i* q0 given value of axial Current i* 0；
The motor speed measurement value is a rotor position signal theta measured by a motor position sensorrIs obtained by treatment;
d-axis current set value i* dAnd d-axis current feedback value idIs fed into the second subtractor i* d-idIs fed into a d-axis current regulator which outputs a d-axis voltage set value V* dBy adjusting V* dSo that i* d-idIs always zero, i.e. for i* dCarrying out no-difference tracking;
given value of q-axis current i* qAnd q-axis current feedback value iqIs sent to the stationThe third subtractor, i* q-iqIs fed into a q-axis current regulator which outputs a q-axis voltage set value V* qBy adjusting V* qSo that the difference is always zero, i.e. for i* qCarrying out no-difference tracking;
0 axis current set value i* 0And 0 axis current feedback value i0Is fed to the fourth subtractor i* 0-i0Is fed into a 0-axis current regulator, which outputs a 0-axis voltage set value V* 0By adjusting V* 0So that i* 0-i0Is always zero, i.e. for i* 0Carrying out no-difference tracking;
V* d、V* q、V* 0inputting the pulse width modulator to perform space vector modulation, and respectively generating an A-phase PWM signal and a B-phase PWM signal;
the converter is formed by connecting a direct current power supply, 4 MOS field effect transistors and 5 freewheeling diodes, wherein drain electrodes of the first MOS field effect transistor, the second MOS field effect transistor, the third MOS field effect transistor and the fourth MOS field effect transistor are connected with the anode of the direct current power supply; the source electrode of the first MOS field effect transistor is connected with the cathode of the first fly-wheel diode and connected with one end of the phase A of the two-phase motor; the source electrode of the third MOS field effect transistor is connected with the cathode of the third fly-wheel diode and is connected with the other end of the phase A of the two-phase motor; the source electrode of the second MOS field effect transistor is connected with the cathode of the second fly-wheel diode and connected with one end of the phase B of the two-phase motor; the source electrode of the fourth MOS field effect transistor is connected with the negative electrode of the fourth fly-wheel diode and is connected with the other end of the B phase of the two-phase motor; the anodes of the first, second, third, fourth and fifth freewheeling diodes are connected with the cathode of the direct current power supply; the midpoint of the phase winding of the motor A, B is connected with the anode of the fifth freewheeling diode; the phase-A PWM signal is used as a control signal of a first MOS field effect transistor grid and a third MOS field effect transistor grid of the inverter, and the phase-B PWM signal is used as a control signal of a second MOS field effect transistor grid and a fourth MOS field effect transistor grid of the inverter;
the output voltage of the converter acts on a direct current bias sine current motor phase winding to control A, B phase current of the direct current bias sine current motor winding and generate a sine current signal with direct current bias corresponding to an input PWM signal; finally, the current of the d, q and 0 axes is tracked without difference;
the phase A and phase B current sensors respectively measure and obtain phase A two split-phase currents iA+、iA-Two phase-separated currents i of phase BB+、iB-Motor rotor position signal θ detected by a motor position sensorrAnd iA+、iA、iB+、iB-Inputting the current into the current converter, performing static-rotation coordinate conversion by the current converter to obtain d, q and 0 axis current signals which are respectively used as d axis current feedback values idQ-axis current iq0 axis current feedback value i0；
The speed regulator outputs a given value i of q-axis current according to the following processq；
The speed regulator judges the input thereof, if nrThe difference of-n is zero, keeping i at this timeqIs constant and output;
if n isrIf the difference between-n is positive, the q-axis current given value i is continuously increasedqUntil the feedback value n of the motor rotating speedrIs increased so that nrN is zero, holding time iqIs constant to maintain nrThe difference of n is always zero;
if n isrThe difference between-n is negative, the given value i of the q-axis current is continuously reducedqUntil the feedback value n of the motor rotating speedrIs reduced so that nrN is zero, holding time iqIs constant to maintain nrThe difference of n is always zero;
the d-axis current regulator outputs a d-axis voltage set value V according to the following process* d；
The d-axis current regulator judges the input thereof if i* d-idIf the difference value of (c) is zero, V is maintained at this time* dIs constant and output;
if i* d-idIf the difference value of (d) is positive, the d-axis voltage given value V is continuously increased* dUp to d-axis current feedback value idIs increased so that i* d-idIs zero, keeping V at this time* dIs constant to maintain i* d-idThe difference of (a) is always zero;
if i* d-idIf the difference is negative, the d-axis voltage given value V is continuously reduced* dUp to d-axis current feedback value idIs reduced so that i* d-idIs zero, keeping V at this time* dIs constant to maintain i* d-idThe difference of (a) is always zero;
the q-axis current regulator outputs a q-axis voltage given value V* q0-axis current regulator outputting 0-axis voltage set value V* 0The procedure of (2) is the same as the above-described procedure, and only the corresponding parameters need to be changed.
The pulse width modulator generates A, B phase PWM signals, including the following processes:
(A) the pulse width modulator is used for generating a motor rotor position signal thetarAnd V* d、V* qRotating and static changing are carried out to obtain V* a、V* b；
Figure BDA0002306856000000041
Wherein V* aIs a given value of A AC voltage in a static coordinate system, V* bThe given value of the B alternating current voltage under a static coordinate system, and p is the pole pair number of the motor;
(B) modulating the generated A AC voltage duty cycle signal Ta＝V* a/VdcDuty cycle signal T of B-phase AC voltageb＝V* b/Vdc(ii) a For V* 0Outputting a DC voltage duty ratio signal T0＝V* 0/VdcIn which V isdcIs the converter DC bus power supply voltage;
(C) finally, the duty ratio T of the A-phase PWM signal is obtainedA：TA＝Ta+T0(ii) a Duty ratio T of B-phase PWM signalB：TB＝Tb+T0。
The current converter is used for measuring the position signal theta of the motor rotorrAnd iA+、iA-、iB+、iB-Carrying out stationary-rotating coordinate transformation according to the following formula to obtain a d-axis current feedback value idQ-axis current feedback value iq0 value of feedback of the axis current i0Is the current amplitude of phase A or phase B;
Figure BDA0002306856000000051
Figure BDA0002306856000000052
wherein RME (i)A+) Is the root mean square value of the A + phase-splitting current.
The invention can realize the optimization control strategy of the novel two-phase direct current bias vernier reluctance motor, and according to the electromagnetic torque formula of the direct current bias sine motor:
Figure BDA0002306856000000053
wherein T iseIs the electromagnetic torque of a DC bias sine current motor, p is the pole pair number of the motor, LdcIs the equivalent inductance of the 0 shaft of the motor, i0For motor 0 shaft current, iqFor the q-axis current of the motor, defining the effective value of the motor phase current as:
Figure BDA0002306856000000054
i is id、iq、i0The resultant vector current in three-dimensional space.
Compared with the prior art, the invention has the following beneficial technical effects:
the converter optimizes the topological structure of the traditional single-phase full-bridge converter, one turn-off power device of each half bridge is replaced by a fly-wheel diode according to the single-phase current characteristic of a load motor, the adopted fly-wheel diode needs to have the fast recovery characteristic, and when the turn-off device performs turn-off action, the fly-wheel diode can rapidly enter a conducting state to form a single-phase current control loop of the motor; because each bridge arm of the motor unidirectional current control loop only comprises a turn-off power device and a freewheeling diode, the danger of bridge arm direct connection is physically avoided, the problem of bridge arm direct connection caused by the traditional full-bridge structure converter is thoroughly solved, and the reliability and the stability of the system are enhanced;
compared with the traditional structure, the quantity of the turn-off devices and the quantity of the diodes in the power electronic device forming the converter are both reduced by half, the gate driving circuit for driving the turn-off devices is correspondingly reduced by half, the pulse width modulation output requirement for generating the digital signal processing of the PWM driving signal is synchronously reduced, the required total devices are fewer, the volume and the weight of the system are reduced, and the power density of the system is improved.
In conclusion, the invention has smaller volume and weight, avoids the bridge arm direct connection risk of the traditional structure in principle, improves the system reliability, effectively reduces the using quantity of power electronic devices, particularly the using quantity of power devices on the premise of ensuring the performance of the controller, can greatly reduce the cost of the controller, and is suitable for the drive control of the novel two-phase direct current bias vernier reluctance motor.
Drawings
FIG. 1 is a schematic cross-sectional view of a two-phase DC-biased vernier reluctance motor;
FIG. 2 is a wiring diagram of the A-phase winding of the two-phase DC bias vernier reluctance motor;
FIG. 3(a) is a diagram illustrating a DC component of a phase current of a DC-biased vernier reluctance motor;
FIG. 3(b) is a schematic diagram of the AC component of one phase current of the DC-biased vernier reluctance motor;
FIG. 3(c) is a diagram of one-phase current of the DC-biased vernier reluctance motor;
fig. 4 is a schematic diagram of a connection mode (single-phase full-bridge inversion topology) of a phase winding of a conventional dc-biased motor controller;
FIG. 5 is a schematic structural view of the present invention;
fig. 6 is a schematic diagram of a current transformer according to the present invention.
Wherein, 1, a first subtracter; 2. a second subtractor; 3. a third subtractor; 4. a fourth subtractor; 5. a speed regulator; 6d, d-axis current regulators; 6q, q-axis current regulators; 60. a 0-axis current regulator; 7. a pulse width modulator; 8. a current transformer; 9. a motor; 10. a motor position sensor; 11. phase A and phase B current sensors; 12. a current transformer.
Detailed Description
The invention is further described with reference to the accompanying drawings in which:
the invention makes the DC component in the DC bias circuit slightly larger than the AC component amplitude, the output torque is slightly reduced at the moment, but the current direction of the motor is ensured to be constant all the time when the motor runs, and only the current magnitude is changed, so the device S in figure 412And S13The four devices output command voltage in a pulse width modulation mode only by playing a role of conducting follow current. Due to the power device S12And S13The winding current is not controlled, and the switching action is not required, so that the control device can be simplified into a freewheeling diode.
Through the analysis, the number of the turn-off power devices of the motor controller is reduced from 4 to 2 per phase, only one freewheeling diode is added per phase, the number of the turn-off power devices required by the whole system is reduced from 12 to 4, and the cost of the system is greatly reduced because the price of the turn-off power devices is higher than that of the freewheeling diodes.
The problem is explained by the fact that in the field of motor control, the natural coordinate system ABC (namely the three-phase stationary coordinate system) is often transformed into the two-phase stationary coordinate system (α - β coordinate system) and the synchronous rotating coordinate system dq0 (rotating coordinate system), the ABC axes are mutually different in 120 electrical degrees in space, and the d, q and 0 axes are mutually perpendicular in the three-dimensional dq0 space and are surrounded by the 0 axis in p omegarRotational speed ofSynchronous rotation, where p is the number of pole pairs of the motor, ωrIs the mechanical angular velocity of the motor. In the present invention, since there are only two-phase windings, the conversion between the two-phase stationary coordinate system and the rotating coordinate system can be directly performed.
In a traditional topological converter, each phase winding needs to be configured with two paths of PWM signals, and two groups of driving units and two paths of control signals are correspondingly needed. After topology optimization, the driving resources can be reduced to half, and meanwhile, the control signals required to be output by the control chip are reduced to half.
As shown in fig. 5, the present invention includes a first subtractor 1, a second subtractor 2, a third subtractor 3, a fourth subtractor 4, a speed regulator 5, a d-axis current regulator 6d, a q-axis current regulator 6q, a 0-axis current regulator 60, a pulse width modulator 7, a current transformer 8, a motor 9, a motor position sensor 10, an a-phase and B-phase current sensor 11, a current transformer 12;
given value n of motor speedrAnd the measured value n of the motor speed is sent to a first subtracter 1, nrThe difference value of-n is fed into a speed regulator 5, and the speed regulator 5 outputs a q-axis current set value i* qBy adjusting iqSo that n isrThe difference of-n is always zero, i.e. n always follows nrWhen the direct current bias sine current motor is in the working state of the maximum torque current ratio, the given value of the d-axis current is 0, and the given value of the q-axis current is equal to that of the 0-axis current, so that the given value i of the d-axis current can be obtained* dQ-axis current given value i* q0 given value of axial Current i* 0；
The motor speed measurement is a rotor position signal theta measured by a motor position sensor 10rIs obtained by treatment, i.e. from
Figure BDA0002306856000000081
Obtaining;
d-axis current set value i* dAnd d-axis current feedback value idIs fed to said second subtractor 2, i* d-idDifference of (2)The value is fed to a d-axis current regulator 6d, said d-axis current regulator 6d outputting a d-axis voltage set value V* dBy adjusting V* dSo that i* d-idIs always zero, i.e. for i* dCarrying out no-difference tracking;
given value of q-axis current i* qAnd q-axis current feedback value iqIs fed to said third subtractor 3, i* q-iqIs fed into a q-axis current regulator 6q, said q-axis current regulator 6q outputting a q-axis voltage set value V* qBy adjusting V* qSo that i* q-iqIs always zero, i.e. for i* qCarrying out no-difference tracking;
0 axis current set value i* 0And 0 axis current feedback value i0Is fed to said fourth subtractor 4, i* 0-i0The difference of (a) is fed to a 0-axis current regulator 60, and the 0-axis current regulator 60 outputs a 0-axis voltage set value V* 0By adjusting V* 0So that i* 0-i0Is always zero, i.e. for i* 0Carrying out no-difference tracking;
V* d、V* q、V* 0inputting the signal into the pulse width modulator 7, performing space vector modulation, and generating an A-phase PWM signal and a B-phase PWM signal respectively;
as shown in fig. 6, the converter 8 is formed by connecting a dc power supply, 4 mosfets and 5 freewheeling diodes, wherein the drains of the first mosfet S1, the second mosfet S2, the third mosfet S3 and the fourth mosfet S4 are connected to the positive electrode of the dc power supply; the source electrode of the first MOS field effect transistor S1 is connected with the negative electrode of the first fly-wheel diode D1 and is connected with one end A of the phase A of the two-phase motor; the source electrode of the third MOS field effect transistor S3 is connected with the negative electrode of the third fly-wheel diode D3 and is connected with the other end X of the phase A of the two-phase motor; the source of the second MOS FET S2 is connected to the negative pole of the second freewheeling diode D2 and connected to the two-phase motorThe other end of phase B is B; the source electrode of the fourth MOS field effect transistor S4 is connected with the negative electrode of the third fly-wheel diode D4 and is connected with the other end Y of the B phase of the two-phase motor; the anodes of the first freewheeling diode D1, the second freewheeling diode D2, the third freewheeling diode D3 and the fourth freewheeling diode D4 are connected with the cathode of the direct-current power supply; the negative electrode of the fifth fly-wheel diode D5 is connected with the negative electrode of the direct current power supply, and the positive electrode of the fifth fly-wheel diode D5 is connected with the midpoint A of the phase winding of the two-phase motor A, BN、BNAre connected. The A-phase PWM signal is used as a control signal of the grid electrodes of a first MOS field effect transistor S1 and a third MOS field effect transistor S3 of the inverter, and the B-phase PWM signal is used as a control signal of the grid electrodes of a second MOS field effect transistor S2 and a fourth MOS field effect transistor S4 of the inverter;
as shown in fig. 5, the output voltage of the converter 8 acts on the dc-biased sinusoidal current motor phase winding to control the A, B phase current of the winding of the dc-biased vernier reluctance motor 9, and generates a sinusoidal current signal with dc bias corresponding to the input PWM signal; finally, the current of the d, q and 0 axes is tracked without difference;
the A-phase and B-phase current sensors 11 respectively measure and obtain two split-phase current signals i of the A-phaseA+、iA-B phase two split-phase current signal iB+、iB-Motor rotor position signal θ detected by motor position sensor 10rAnd iA+、iA-、iB+、iB-Inputting the current into the current converter, performing stationary-rotating coordinate transformation by the current converter 12 to obtain d, q, and 0 axis current signals as d axis current feedback values idQ-axis current iq0 axis current feedback value i0；
The speed regulator 5 outputs a q-axis current set value i according to the following processq；
The speed regulator 5 determines its input if nrThe difference of-n is zero, keeping i at this timeqIs constant and output;
if n isrIf the difference between-n is positive, the q-axis current given value i is continuously increasedqUntil the feedback value n of the motor rotating speedrIs increased so that nr-n is zero,keep at this time iqIs constant to maintain nrThe difference of n is always zero;
if n isrThe difference between-n is negative, the given value i of the q-axis current is continuously reducedqUntil the feedback value n of the motor rotating speedrIs reduced so that nrN is zero, holding time iqIs constant to maintain nrThe difference of n is always zero;
the d-axis current regulator outputs a d-axis voltage set value V according to the following process* d；
The d-axis current regulator 6d judges the input thereof, if i* d-idIf the difference value of (c) is zero, V is maintained at this time* dIs constant and output;
if i* d-idIf the difference value of (d) is positive, the d-axis voltage given value V is continuously increased* dUp to d-axis current feedback value idIs increased so that i* d-idIs zero, keeping V at this time* dIs constant to maintain i* d-idThe difference of (a) is always zero;
if i* d-idIf the difference is negative, the d-axis voltage given value V is continuously reduced* dUp to d-axis current feedback value idIs reduced so that i* d-idIs zero, keeping V at this time* dIs constant to maintain i* d-idThe difference of (a) is always zero;
the q-axis current regulator 6q outputs a q-axis voltage given value V* q0-axis current regulator 60 outputs 0-axis voltage set value V* 0The procedure of (2) is the same as the above-described procedure, and only the corresponding parameters need to be changed.
The generation of the A, B-phase PWM signal by the pulse width modulator 7 comprises the following processes:
(A) the pulse width modulator 7 is used for the motor rotor position signal thetarAnd V* d、V* q、V* d0The rotation and the static state of the rotating shaft are changed,to obtain V* a、V* b；
Figure BDA0002306856000000101
Wherein
Figure BDA0002306856000000102
Is a given value of A AC voltage in a static coordinate system, V* bThe given value of the B alternating current voltage under a static coordinate system, and p is the pole pair number of the motor;
(B) modulating the generated A AC voltage duty cycle signal Ta＝V* a/VdcDuty cycle signal T of B-phase AC voltageb＝V* b/Vdc(ii) a For V* 0Outputting a DC voltage duty ratio signal T0＝V* 0/VdcIn which V isdcIs the converter DC bus power supply voltage;
(C) finally, the duty ratio T of the A-phase PWM signal is obtainedA：TA＝Ta+T0(ii) a Duty ratio T of B-phase PWM signalB：TB＝Tb+T0。
The current converter 12 is used for the motor rotor position signal thetarAnd iA+、iA-、iB+、iB-Carrying out stationary-rotating coordinate transformation according to the following formula to obtain a d-axis current feedback value idQ axial flow feedback value iq0 value of feedback of the axis current i0Is the amplitude of phase A or phase B current;
Figure BDA0002306856000000111
Figure BDA0002306856000000112
the invention is suitable for motor driving occasions sensitive to cost control, including but not limited to consumer-grade household appliance driving motors such as washing machines, air conditioners, dust collectors and refrigerators, lawn mowers, low-speed electric vehicle driving motors and the like. The requirements for motor performance and control accuracy for these applications are often not particularly high, but are extremely cost sensitive in view of commercial production. The direct current bias vernier motor driving system with low cost and high reliability is a good choice.

Claims (10)

1. A two-phase direct current offset vernier motor controller is characterized by comprising a first subtracter (1), a second subtracter (2), a third subtracter (3), a fourth subtracter (4), a speed regulator (5), a d-axis current regulator (6d), a q-axis current regulator (6q), a 0-axis current regulator (60), a pulse width modulator (7), a current transformer (8), a motor (9), a motor position sensor (10), an A-phase and B-phase current sensor (11) and a current converter (12);
the first subtracter (1) is used for receiving a given value n of the rotating speed of the motorrAnd a measured value n of the motor speed is obtained, and the difference value n is obtainedr-n is fed into a speed regulator (5), the output end of the speed regulator (5) is respectively connected to a d-axis current regulator (6d), a q-axis current regulator (6q) and a 0-axis current regulator (60) through a second subtractor (2), a third subtractor (3) and a fourth subtractor (4), the output ends of the d-axis current regulator (6d), the q-axis current regulator (6q) and the 0-axis current regulator (60) are respectively connected to a pulse width modulator (7), the output end of the pulse width modulator (7) is connected to a current transformer (8), the output end of the current transformer (8) is connected to a motor (9), a motor position sensor (10) is connected to the motor (9), an a-phase and a-phase current sensor (11) is further connected between the current transformer (8) and the motor (9), and the output ends of the a-phase and B-phase current sensors (11) are respectively connected to the second subtractor (2) and the, A third subtracter (3) and a fourth subtracter (4), and the output end of the motor position sensor (10) is respectively connected to the first subtracter (1), the pulse width modulator (7) and the current converter (12).
2. The two-phase DC-biased vernier motor controller according to claim 1, wherein said current transformer (8) is composed of a DC power supply, a first MOS FET S1, a second MOS FET S2, a third MOS FET S3, a fourth MOS FET S4, a first freewheeling diode D1, a second freewheeling diode D2, a third freewheeling diode D3, a fourth freewheeling diode D4 and a fifth freewheeling diode D5;
the drains of the first MOS field effect transistor S1, the second MOS field effect transistor S2, the third MOS field effect transistor S3 and the fourth MOS field effect transistor S4 are connected with the anode of the direct current power supply; the source electrode of the first MOS field effect transistor S1 is connected with the negative electrode of the first fly-wheel diode D1 and connected with the A end of the A-phase winding of the motor (9); the source electrode of the second MOS field effect transistor S2 is connected with the negative electrode of the second fly-wheel diode D2 and connected with the B end of the B-phase winding of the motor (9); the source electrode of the third MOS field effect transistor S3 is connected with the negative electrode of the third fly-wheel diode D3 and is connected with the X end of the A-phase winding of the motor (9); the source electrode of the fourth MOS field effect transistor S4 is connected with the negative electrode of a fourth freewheeling diode D4 and connected with the Y end of the B-phase winding of the motor (9); the cathodes of the first freewheeling diode D1, the second freewheeling diode D2, the third freewheeling diode D3, the fourth freewheeling diode D4 and the fifth freewheeling diode D5 are connected with the cathode of the direct-current power supply; the positive pole of the fifth fly-wheel diode D5 and the midpoint A of A, B phase winding of the motor (9)N、BNConnecting; the pulse width modulator (7) generates an A-phase PWM signal as a control signal of the gates of the first and third MOSFETs S1 and S3, and the pulse width modulator (7) generates a B-phase PWM signal as a control signal of the gates of the second and fourth MOSFETs S2 and S4.
3. A control method of two-phase DC offset vernier motor, which adopts the two-phase DC offset vernier motor controller as claimed in claim 1, characterized in that the given value n of the motor rotation speedrAnd the measured value n of the motor rotating speed is sent to a first subtracter (1), nrThe difference value of-n is sent to a speed regulator (5), and the speed regulator (5) outputs a q-axis current set value i* qBy adjusting i* qSo that n isrThe difference of-n is always zero, i.e. n always follows nrThe d-axis current given value is 0, the 0-axis current given value is equal to the q-axis current given value, and the d-axis current given value i is obtained* dQ-axis current given value i* qAnd 0 axis current set value i* 0；
d-axis current set value i* dAnd d-axis current feedback value idIs fed into the second subtractor (2), i* d-idIs fed into a d-axis current regulator (6d), said d-axis current regulator (6d) outputting a d-axis voltage set value V* dBy adjusting V* dSo that i* d-idIs always zero, i.e. for i* dCarrying out no-difference tracking;
given value of q-axis current i* qAnd q-axis current feedback value iqIs fed into the third subtractor (3), i* q-iqIs fed into a q-axis current regulator (6q), said q-axis current regulator (6q) outputting a q-axis voltage setpoint value V* qBy adjusting V* qSo that i* q-iqIs always zero, i.e. for i* qCarrying out no-difference tracking;
0 axis current set value i* 0And 0 axis current feedback value i0Is fed into the fourth subtractor (4), i* 0-i0Is fed into a 0-axis current regulator (60), and the 0-axis current regulator (60) outputs a 0-axis voltage set value V* 0By adjusting V* 0So that i* 0-i0Is always zero, i.e. i* 0Carrying out error-free tracking;
V* d、V* q、V* 0inputting the signal into the pulse width modulator (7) for space vector modulation to generate an A-phase PWM signal and a B-phase PWM signal respectively;
the output voltage of the converter (8) acts on a winding of the motor (9), controls A, B phase current of the winding of the motor (9), and generates a sinusoidal current signal with direct current bias corresponding to an input A phase PWM signal and an input B phase PWM signal; finally, the no-difference tracking of the d, q and 0 axis currents is realized.
4. The method as claimed in claim 3, wherein the d-axis current feedback value i is obtained by a two-phase DC offset vernier motordQ-axis current feedback value iqAnd 0 axis current feedback value i0The acquisition mode is as follows:
a phase current sensor (11) and a B phase current sensor (11) are utilized to respectively measure and obtain two split-phase currents i of the A phaseA+、iA-Two phase-separated currents i of phase BB+、iB-Motor rotor position signal theta detected by motor position sensor (10)rWill thetarAnd iA+、iA-、iB+、iB-An input current converter (12) for performing stationary-rotary coordinate conversion by the current converter (12) in accordance with the following equation to obtain d, q, and 0 axis current signals as d axis current feedback values idQ-axis current feedback value iqAnd 0 axis current feedback value i0；
Figure FDA0002306855990000031
Figure FDA0002306855990000032
Where p is the number of pole pairs of the motor, RMS (i)A+) Is the root mean square value of the A + phase-splitting current.
5. A method for controlling a two-phase dc offset vernier motor according to claim 3, wherein the measured value of the motor speed n is the rotor position signal θ measured by the motor position sensor (10)rThe method comprises the following steps:
Figure FDA0002306855990000033
6. the method as claimed in claim 3, wherein the two-phase DC offset vernier motor control method is characterized in thatThe speed regulator (5) outputs a q-axis current set value i according to the following processq: the speed regulator (5) determines its input if nrThe difference of-n is zero, keeping i at this timeqIs constant and output; if n isrIf the difference between-n is positive, the q-axis current given value i is continuously increasedqUntil the feedback value n of the motor rotating speedrIs increased so that nrN zero, holding at this time iqIs a constant value; if n isrThe difference between-n is negative, the given value i of the q-axis current is continuously reducedqUntil the feedback value n of the motor rotating speedrIs reduced so that nrN is zero, holding time iqIs a constant value.
7. A two-phase dc offset vernier motor control method according to claim 3, characterized in that said d-axis current regulator (6d) outputs a d-axis voltage set value V according to the following procedure* d: the d-axis current regulator (6d) judges the input thereof, if i* d-idIf the difference value of (c) is zero, V is maintained at this time* dIs constant and output; if i* d-idIf the difference value of (d) is positive, the d-axis voltage given value V is continuously increased* dUp to d-axis current feedback value idIs increased so that i* d-idIs zero, keeping V at this time* dIs a constant value; if i* d-idIf the difference is negative, the d-axis voltage given value V is continuously reduced* dUp to d-axis current feedback value idIs reduced so that i* d-idIs zero, keeping V at this time* dIs a constant value.
8. A two-phase dc offset vernier motor control method according to claim 3, characterized in that said q-axis current regulator (6q) outputs a q-axis voltage set value V according to the following procedure* q: the q-axis current regulator (6q) determines the input thereof, if i* q-iqIf the difference value of (A) is zero, then it is guaranteedAt this moment V* qIs constant and output; if i* q-iqIf the difference value of (1) is positive, the given value V of the q-axis voltage is continuously increased* qUp to the q-axis current feedback value iqIs increased so that i* q-iqIs zero, keeping V at this time* qIs a constant value; if i* q-iqIf the difference is negative, the given value V of the q-axis voltage is continuously reduced* qUp to the q-axis current feedback value iqIs reduced so that i* q-iqIs zero, keeping V at this time* qIs a constant value.
9. A two-phase dc offset vernier motor control method according to claim 3, wherein said 0-axis current regulator (60) outputs a 0-axis voltage set value V according to the following procedure* 0: the 0-axis current regulator (60) determines the input thereof if i* 0-i0If the difference value of (c) is zero, V is maintained at this time* 0Is constant and output; if i* 0-i0If the difference value of (1) is positive, the given value V of the shaft voltage of 0 is continuously increased* 0Up to 0 axis current feedback value i0Is increased so that i* 0-i0Is zero, keeping V at this time* 0Is a constant value; if i* 0-i0If the difference is negative, the given value V of the shaft voltage of 0 is continuously reduced* 0Up to 0 axis current feedback value i0Is reduced so that i* 0-i0Is zero, keeping V at this time* 0Is a constant value.
10. A two-phase dc-biased vernier motor control method according to claim 3, wherein the pulse width modulator (7) generating a-phase PWM signal and a B-phase PWM signal comprises the following process:
(A) pulse width modulator (7) for motor rotor position signal thetarAnd
Figure FDA0002306855990000051
rotating and static changing are carried out to obtain
Figure FDA0002306855990000052
Figure FDA0002306855990000053
Wherein
Figure FDA0002306855990000054
Is a given value of A AC voltage in a static coordinate system, V* bThe given value of the B alternating current voltage under a static coordinate system, and p is the pole pair number of the motor;
(B) modulating the generated A AC voltage duty cycle signal Ta＝V* a/VdcDuty cycle signal T of B-phase AC voltageb＝V* b/Vdc(ii) a For V* 0Outputting a DC voltage duty ratio signal T0＝V* 0/VdcIn which V isdcIs the converter DC bus power supply voltage;
(C) finally, the duty ratio T of the A-phase PWM signal is obtainedA：TA＝Ta+T0(ii) a Duty ratio T of B-phase PWM signalB：TB＝Tb+T0。
CN201911243357.1A 2019-12-06 2019-12-06 Two-phase direct current offset vernier motor controller and control method Pending CN111030537A (en)

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