CN110995098B - Inversion control method of permanent magnet synchronous motor - Google Patents
Inversion control method of permanent magnet synchronous motor Download PDFInfo
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- CN110995098B CN110995098B CN201911357307.6A CN201911357307A CN110995098B CN 110995098 B CN110995098 B CN 110995098B CN 201911357307 A CN201911357307 A CN 201911357307A CN 110995098 B CN110995098 B CN 110995098B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
Abstract
The invention provides an inversion control method of a permanent magnet synchronous motor, which comprises the following steps: s1, establishing a motor electrical characteristic equation in a state space form and a mechanical characteristic equation in the state space form; s2, establishing a discrete state transfer equation of the electrical characteristics and the mechanical characteristics of the motor; s3, predicting future k + n through the motor rotation state variable and current of the current k period 1 The periodic motor rotation state is obtained to obtain the future k + n of the motor 1 A periodic rotation state deviation predicted value; s4, according to the predicted value of the deviation of the rotation state, inverting to obtain k + 1-k + n 1 -1 period control current instruction sequence, compensating the control current instruction sequence according to the rotation state deviation caused by the load, and obtaining a control voltage instruction U of k period compensation load through inversion 1，r (k) '; s5, predicting k + n according to the voltage and current of the current k period 2 The current error value of the periodic motor is reversely deduced according to the predicted current error value to obtain a control voltage instruction of k period, and U is added 1，r (k) ' as the actual control voltage input for the motor.
Description
Technical Field
The invention relates to the field of electric automation control, in particular to an inversion control method of a permanent magnet synchronous motor.
Background
The permanent magnet synchronous motor has the characteristics of time variation, nonlinearity and strong coupling, and is widely applied to high-precision position servo systems, such as rotary tables, numerical control machines, robot joints, control moment gyros, missile tail fin steering engines and other servo systems which require high precision, high stability and high dynamic response, due to the advantages of high power density, low loss, small torque fluctuation, convenience in maintenance and the like. Under various tasks and load working conditions, the permanent magnet synchronous motor is required to have the control quality of timely response, small control overshoot, short control stabilization time, small control static difference, strong torque interference and strong load robustness.
With the development of computer, power electronics and sensor technologies, at present, variables such as voltage, current, corner, rotating speed and the like can be measured at high speed and high precision, and a high-performance chip can quickly calculate a complex algorithm and control a power supply and an inverter to work at high frequency, so that a plurality of advanced and complex control algorithms based on models can be applied to the drive control of the permanent magnet synchronous motor.
At present, methods of PID control, fuzzy control, sliding mode variable structure control, self-adaptive control, neural network control and the like of a permanent magnet synchronous motor have the problems of poor control precision/stability, low robustness on interference load and motor body parameters, large influence of artificial subjective factors of controller design and the like in different degrees, and the like, and the problems of good dynamic performance and steady-state performance and the like are difficult to ensure due to design from the stability perspective.
Disclosure of Invention
The invention aims to provide an inversion control method of a permanent magnet synchronous motor, which is used for predicting the k + n th according to the rotation characteristic, the electrical characteristic and the mechanical characteristic of the permanent magnet synchronous motor from the aspect of system controllability 1 The motor rotation state after the period is reversely deduced according to the predicted motor rotation state to obtain a compensated control current instruction of the k period 2 Periodic current value (where n 2 ∈[1,n 1 -1]) According to predicted k + n 2 Periodic current value and said compensated control current for k-periodAnd the actual control input voltage value of the k-period motor is obtained by inverting the difference value, so that stronger robustness to external torque disturbance is realized.
In order to achieve the above object, the present invention provides an inversion control method for a permanent magnet synchronous motor, comprising the steps of:
s1, establishing a motor electrical characteristic equation in a state space form and a motor mechanical characteristic equation in a state space form under a d-q coordinate system of a continuous time system;
s2, respectively establishing a discrete state transfer equation of the electrical characteristic and the mechanical characteristic of the motor;
s3, according to a discrete state transition equation of the mechanical characteristics of the motor, the motor rotation state variable X (k) = [ theta ] of the current k period m (k),ω m (k)]K period current i q (k) Predicting future k + n 1 Periodic motor rotation state variable
And with the desired k + n 1 Periodic reference rotational state variable X r (k+n 1 ) The comparison results in the future k + n of the motor 1 A periodic rotation state deviation predicted value; wherein, theta m (k)、ω m (k) Rotor angular position and rotor mechanical angular velocity, respectively, of k cycles; n is a radical of an alkyl radical 1 ≥2；
S4, according to the rotation state deviation predicted value, inverting to obtain k + 1-k + n 1 -a control current command sequence of 1 cycle; then, calculating a current compensation value of a k +1 period according to the motor rotation state deviation caused by load and disturbance; on the basis, a control voltage instruction of the k-period compensation load is further calculated; the control voltage command of the k-period compensation load is also the compensation voltage value of the k period;
s5, predicting k + n according to the motor voltage and the motor current of the current k period 2 Current error value of periodic motor
Wherein n is 2 ∈[1,n 1 -1](ii) a According to
The actual control voltage instruction of the k period is obtained through reverse deduction, and the actual control voltage instruction of the k period compensation load is added to be used as the actual control voltage input of the motor in the k period; k is updated to k +1, and the process proceeds to S3.
The step S1 includes:
s11, under a d-q coordinate system, using a current U 2 ＝[i d i q ] T Is a state variable, voltage U 1 ＝[v d v q ] T As a control input, a state space form of a motor electrical characteristic equation shown in formula (1) is obtained:
in the formula (1)
Wherein L is d 、L q Respectively d-axis and q-axis inductances, R is a stator winding resistance, omega m The mechanical angular speed of the rotor is shown, lambda is the magnetic flux amplitude of the permanent magnet of the rotor at the side of the stator, and p is the pole pair number; i.e. i d ，i q Current of the permanent magnet synchronous motor on a d axis and a q axis respectively, v d ，v q The voltages of the permanent magnet synchronous motor on the d axis and the q axis are obtained through Clark conversion and Park conversion of the measured three-phase current and voltage.
S12, establishing a mechanical property equation under a continuous time system under a d-q coordinate system, wherein the equation is shown in the formulas (2) and (3):
T e ＝1.5p[λi q +(L d -L q )i d i q ] (2)
wherein, T e Is electricityMagnetic torque, J is rotor moment of inertia, B is viscous friction coefficient, T l Is the load torque;
s13, establishing a rotation characteristic equation under the continuous time system, wherein the equation is shown in a formula (4):
wherein, theta m Is the rotor angular position;
s14, adopting i d The vector control strategy of =0 obtains a state space form of a motor mechanical characteristic equation shown in formula (5) according to formulas (2), (3) and (4):
wherein X = [ theta ] m ω m ] T Is a state variable, U 2 ＝[i q ]For current control input, Q = [ T ] l ]In order to be the load,
the step S2 includes:
s21, discretizing the formula (1) by utilizing bilinear transformation to obtain a discrete state transfer equation of the electrical characteristics of the motor shown as the formula (6):
U 2 (k+1)＝A d1 (k)U 2 (k)+B d1 U 1 (k)+F(k) (6)
(6) Is an equation representing the current generated by the voltage; wherein k represents the current cycle, and k +1 represents the next cyclePeriod i d (k)、i q (k) The currents omega of the k period permanent magnet synchronous motor on the d axis and the q axis respectively m (k) For k-period rotor mechanical angular velocity, T s Is a sampling period;
s22, carrying out state transition solution on the formula (5), and obtaining a discrete state transition equation of the mechanical characteristics of the motor shown in the formula (7) by utilizing pull transformation or power series approximate solution:
X(k+1)＝A d2 X(k)+B d2 U 2 (k)+H d Q(k) (7)
a load Q = [ T ] with tau as a control period and Q (k) as k period l ]The value of (c).
The step S3 specifically includes:
s31, calculating k + n according to the formula (7) without considering the load Q (k) of the k period 1 Periodic motor rotation state variable
In the formula (8), the reaction mixture is,
S32, in k +1 to k + n 1 1 cycle without control input and without considering load effect, let k + n 1 Periodic motor rotation reference state is X r (k+n 1 ) And calculating to obtain the motor at k + n 1 Periodic rotation state to obtain k + n 1 Periodic rotational state deviation prediction
The step S4 specifically includes:
s41, obtaining a control current instruction sequence set shown in a formula (10) according to the formulas (8) and (9) and a generalized inverse control theory,
in the formula (10), M (k + n) 1 Row full rank of-1,k + 1), M (k + n) 1 -1,k+1) - For its generalized inverse, M (k + n) 1 -1,k+1) - ＝M(k+n 1 -1,k+1) T (M(k+n 1 -1,k+1)M(k+n 1 -1,k+1) T ) -1 ，I 2 Is (n) 1 -1) dimensional unit matrix, V being (n) 1 -1) a dimension arbitrary column vector;
s42, obtaining a control current command sequence by carrying out the unique least square and minimum norm solution in the solution of the formula (10):
s43, calculating a predicted value of the deviation of the rotation state of the motor in consideration of the load action in the control current command sequence expressed by equation (11), as expressed by equation (12):
wherein, the first and the second end of the pipe are connected with each other,
s44, designing a current compensation value of a k +1 period according to the formula (12);
wherein, K c ＝diag([k c1 k c2 ])，k c1 k c2 A compensation factor is positive; Δ X (j) is a deviation value of a rotation state actually measured in a j period;
s45, obtaining the control voltage command of the k-period compensation load according to equations (13) and (6) as follows:
the step S5 specifically includes:
s51, rotor mechanical angular speed omega according to current k period m (k) And equation (6) predicting motor at k + n 2 Periodic current value
n 2 ∈[1,n 1 -1]；
Wherein the content of the first and second substances,
i is a 2-dimensional unit array,
s52, calculating k + n 2 Error value of periodically predicted current value
Wherein, U 2,r (k+n 2 ) A control current command sequence U obtained for the current reference value by equation (11) 2,r (k+n 1 -1, k + 1);
s53, according to
And a generalized inverse control theory, reversely deducing to obtain a control voltage command sequence of k period as shown in a formula (16),
P(k+n 2 -1,k) - is P (k + n) 2 -1, k) generalized inverse;
u obtained by equation (16) 1,r (k+n 2 U in-1,k) 1,r (k) U obtained by adding formula (14) 1，r (k) ' as the actual control voltage input for the k-cycle motor; k is updated to k +1, and the process proceeds to S3.
Compared with the prior art, the inversion control method of the permanent magnet synchronous motor has the beneficial effects that:
(1) The tracking precision is high, the response speed is high, and the robustness to external moment disturbance is strong.
The invention inverses the control quantity of the motor based on the established discrete state transfer equation of the electrical characteristic and the mechanical characteristic of the motor, the motor rotation state measured in real time and the planned motor rotation state. The parameter deviation and disturbance of the permanent magnet synchronous motor are finally reflected in the rotation state deviation of the motor. The invention directly solves the load compensation control input of the motor according to the predicted deviation value of the motor rotation state, and has stronger robustness. The invention relates to an accurate quantitative control method for adjusting control parameters (namely control current instructions) input by control in real time according to predicted motor rotation state deviation, which has the characteristics of self-adaption and accurate quantitative control.
(2) Wide application range
The reference tracks of the rotation angle and the rotation speed of the motor are planned according to different tasks, and high-precision position servo control, such as fixed point control and track tracking control, can be realized. The high-precision servo steering engine can be applied to high-dynamic servo steering engines of missile steering engines, numerical control machines, robot joints, rotary tables and other high-precision servo systems.
(3) The method is simple and effective and is convenient to adjust.
The control parameters (namely control current instructions) of the invention are determined by the control period and the measurable three-phase current of the motor, the angular position of the rotor and the angular speed of the rotor, are self-adaptive along with the time variation of the state of the motor, are convenient to apply to a digital control system, are suitable for the same type of motors with different parameters such as resistance, inductance, rotor rotational inertia, friction damping and the like, and are beneficial to designing the controllers of the permanent magnet synchronous motors in batches according to the method of the invention.
Drawings
In order to more clearly illustrate the technical solution of the present invention, the drawings used in the description will be briefly introduced, and it is obvious that the drawings in the following description are an embodiment of the present invention, and other drawings can be obtained by those skilled in the art without creative efforts according to the drawings:
fig. 1 is a flowchart of an inversion control method of a permanent magnet synchronous motor according to the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be obtained by a person skilled in the art without making any creative effort based on the embodiments in the present invention, belong to the protection scope of the present invention.
The invention aims to provide an inversion control method of a permanent magnet synchronous motor, which predicts the k + n th according to the rotation characteristic, the electrical characteristic and the mechanical characteristic of the permanent magnet synchronous motor from the viewpoint of system controllability 1 The motor rotation state after the period is reversely deduced according to the predicted motor rotation state to obtain a compensated control current instruction of the k period 2 Periodic current value (where n 2 ∈[1,n 1 -1]) According to predicted k + n 2 And inverting the difference value between the periodic current value and the compensated control current instruction of the k period to obtain the actual control input voltage value of the k period motor, thereby realizing stronger robustness to external torque disturbance.
The inversion control method of the permanent magnet synchronous motor comprises the following steps:
s1, establishing a motor electrical characteristic equation in a state space form and a motor mechanical characteristic equation in a state space form under a d-q coordinate system of a continuous time system;
the step S1 specifically includes:
s11, under a d-q coordinate system, using a current U 2 ＝[i d i q ] T As state variables, voltage U 1 ＝[v d v q ] T As a control input, a state space form of a motor electrical characteristic equation shown in formula (1) is obtained:
in the formula (1)
Wherein L is d 、L q Respectively d-axis and q-axis inductances, R is a stator winding resistance, omega m In order to obtain the mechanical angular velocity of the rotor, lambda is the magnetic flux amplitude of the rotor permanent magnet at the stator side, and p is the pole pair number; i.e. i d ，i q Current of the permanent magnet synchronous motor on a d axis and a q axis respectively, v d ，v q The voltages of the permanent magnet synchronous motor on the d axis and the q axis are obtained through Clark conversion and Park conversion of the measured three-phase current and voltage.
S12, establishing a mechanical property equation under a continuous time system under a d-q coordinate system, wherein the equations are shown in formulas (2) and (3):
T e ＝1.5p[λi q +(L d -L q )i d i q ] (2)
wherein, T e Is electromagnetic torque, J is rotor moment of inertia, B is viscous friction coefficient, T l Is the load torque;
s13, establishing a rotation characteristic equation under a continuous time system, wherein the equation is shown as the formula (4):
wherein, theta m Is the rotor angular position;
s14, adopting i d And (5) according to the vector control strategy of =0, obtaining a state space form of a motor mechanical characteristic equation shown in formula (5) according to the formulas (2), (3) and (4):
wherein X = [ theta ] m ω m ] T Is a state variable, U 2 ＝[i q ]For control input, Q = [ T = l ]Is a load of the electric vehicle,
equation (5) is an equation representing the rotation angle and the rotation speed generated by the current.
S2, respectively establishing a discrete state transfer equation of the electrical characteristic and the mechanical characteristic of the motor;
the step S2 includes:
s21, discretizing the formula (1) by utilizing bilinear transformation to obtain a discrete state transfer equation of the electrical characteristics of the motor shown in the formula (6):
U 2 (k+1)＝A d1 (k)U 2 (k)+B d1 U 1 (k)+F(k) (6)
where k represents the current cycle, k +1 represents the next cycle, i d (k)、i q (k) The currents of the k-period permanent magnet synchronous motor on the d axis and the q axis are respectively omega m (k) Mechanical angular speed of rotor of k period, T s Is a sampling period;
s22, carrying out state transition solution on the formula (5), and obtaining a discrete state transition equation of the mechanical characteristics of the motor shown in the formula (7) by utilizing pull transformation or power series approximate solution:
X(k+1)＝A d2 X(k)+B d2 U 2 (k)+H d Q(k) (7)
a load Q = [ T ] with tau as a control period and Q (k) as k period l ]The value of (c).
S3, according to a discrete state transition equation of the mechanical characteristics of the motor, the motor rotation state variable X (k) = [ theta ] of the current k period m (k),ω m (k)]K period of current i q (k) Predicting future k + n 1 Periodic motor rotation state variable
And with the desired k + n 1 Periodic reference rotational state variable X r (k+n 1 ) The comparison results in the future k + n of the motor 1 Predicting deviation values of the periodic rotation states; wherein, theta m (k)、ω m (k) Rotor angular position and rotor mechanical angular velocity, respectively, of k cycles; n is 1 ≥2；
The step S3 specifically includes:
s31, calculating k + n according to the formula (7) without considering the load Q (k) of the k period 1 Periodic motor rotation state variable
In the formula (8), the reaction mixture is,
S32, in k +1 to k + n 1 1 period without control input and without considering load effect, let k + n 1 Periodic motor rotation reference state is X r (k+n 1 ) And calculating to obtain the motor at k + n 1 Periodic rotation state, predicted deviation value
S4, according to the deviation value of the rotation state prediction, k + 1-k + n is obtained through inversion 1 -a control current command sequence of 1 cycle; next, calculating a current compensation value of a k +1 period according to the motor rotation state deviation caused by load and disturbance; on the basis, a control voltage instruction of the k-period compensation load is further calculated;
the step S4 specifically includes:
s41, obtaining a control current instruction sequence set shown in a formula (10) according to the formulas (8) and (9) and a generalized inverse control theory,
in the formula (10), M (k + n) 1 Row full rank of-1, k + 1), M (k + n) 1 -1,k+1) - M (k + n) being the generalized inverse thereof 1 -1,k+1) - ＝M(k+n 1 -1,k+1) T (M(k+n 1 -1,k+1)M(k+n 1 -1,k+1) T ) -1 ，I 2 Is (n) 1 -1) dimensional unit matrix, V being (n) 1 -1) a dimension arbitrary column vector. Since the equation (10) has different control current command sequences depending on the value of V, the equation (10) represents a control current command sequence set.
Equation (8) is what state of rotation the motor has, assuming that the control current command sequence is known, and the current is not known in fact, and has not yet occurred. And k + n in the future 1 The desired motor rotation state of the cycle is known as X r (k+n 1 ) Can be based on the predicted deviation value
And (4) inverting to obtain a control current command sequence, namely an equation (10).
S42, obtaining a control current command sequence by carrying out the unique least square and minimum norm solution in the solution of the formula (10):
equation (8) is a representation of a control current command sequence; (10) The formula is an expression form of the open solution of the control current command sequence, and different control current command sequences are obtained through any value of V and the formula (10); (11) The expression is a special solution with the minimum norm in the general solution of the expression (10), i.e., the expression (10) includes the expression (11).
S43, calculating a predicted deviation value of the rotation state of the motor under the control current command sequence shown in the formula (11) by considering the action of the load, wherein the predicted deviation value is shown in the formula (13):
wherein, the first and the second end of the pipe are connected with each other,
s44, according to the rotation state prediction deviation value obtained by the formula (12), designing the load compensation control input of the k period as follows,
wherein, K c ＝diag([k c1 k c2 ])，k c1 k c2 A compensation factor is positive; Δ X (j) is a deviation value of a rotation state actually measured in a j period;
because the actual deviation values of the rotation states in the k period, the k +1 period and the following periods can not be obtained, the deviation values actually measured in the rotation states in the k period and the preceding periods are accumulated and then compensated to obtain the compensation current required by the k +1 period,
s45, obtaining a control voltage command of the load needing to be compensated in the k period according to the formula (13) and the formula (6), wherein the control voltage command comprises the following steps:
s5, predicting k + n according to the motor voltage and the motor current of the current k period 2 Current error value of periodic motor
Wherein n is 2 ∈[1,n 1 -1](ii) a According to
The actual control voltage instruction of the k period is obtained through reverse deduction, and the actual control voltage instruction of the k period compensation load is added to be used as the actual control voltage input of the motor in the k period; k is updated to k +1, and the process proceeds to S3.
The step S5 specifically includes:
s51, according to the currentMotor voltage, current and equation (6) for k cycles, predicting motor at k + n 2 Periodic current value
n 2 ∈[1,n 1 -1]；
Wherein the content of the first and second substances,
i is a 2-dimensional unit array,
s52, calculating k + n 2 Error value of periodically predicted current value
Wherein, U 2,r (k+n 2 ) A current command sequence U obtained for the current reference value by equation (11) 2,r (k+n 1 -1,k + 1);
s53, according to
And a generalized inverse control theory, reversely deducing to obtain a control voltage command sequence of k period as shown in a formula (16),
P(k+n 2 -1,k) - is P (k + n) 2 -1, k) generalized inverse;
the sum of the formula (16) and the formula (14) is used as the actual control voltage input of the periodic motor; k is updated to k +1, and the process proceeds to S3.
While the invention has been described with reference to specific embodiments, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention. Therefore, the protection scope of the present invention shall be subject to the protection scope of the claims.
Claims (6)
1. An inversion control method of a permanent magnet synchronous motor is characterized by comprising the following steps:
s1, establishing a motor electrical characteristic equation in a state space form and a motor mechanical characteristic equation in a state space form under a d-q coordinate system of a continuous time system;
s2, respectively establishing a discrete state transfer equation of the electrical characteristic and the mechanical characteristic of the motor;
s3, according to a discrete state transition equation of the mechanical characteristics of the motor, the motor rotation state variable X (k) = [ theta ] of the current k period m (k),ω m (k)]K period current i q (k) Predicting future k + n 1 Periodic motor rotation state variable
And with the desired k + n 1 Periodic reference rotational state variable X r (k+n 1 ) The comparison result shows that the motor k + n in the future 1 A periodic rotation state deviation predicted value; wherein, theta m (k)、ω m (k) Rotor angular position and rotor mechanical angular velocity, respectively, of k cycles; n is a radical of an alkyl radical 1 ≥2；
S4, according to the predicted value of the rotation state deviation, inverting to obtain k + 1-k + n 1 -a control current command sequence of 1 cycle; next, calculating a current compensation value of a k +1 period according to the motor rotation state deviation caused by load and disturbance; on the basis, a control voltage instruction of the k-period compensation load is further calculated;
s5, predicting k + n according to the motor voltage and the motor current of the current k period 2 Current error value of periodic motor
Wherein n is 2 ∈[1,n 1 -1](ii) a According to
The actual control voltage instruction of the k period is obtained through reverse deduction, and the actual control voltage instruction of the k period compensation load is added to be used as the actual control voltage input of the motor in the k period; k is updated to k +1, and the process proceeds to S3.
2. The inversion control method of a permanent magnet synchronous motor according to claim 1, wherein the step S1 includes:
s11, under a d-q coordinate system, using a current U 2 ＝[i d i q ] T Is a state variable, voltage U 1 ＝[v d v q ] T As a control input, a state space form of a motor electrical characteristic equation shown in formula (1) is obtained:
in the formula (1)
Wherein L is d 、L q D-axis and q-axis inductances, R is stator winding resistance, omega m The mechanical angular speed of the rotor is shown, lambda is the magnetic flux amplitude of the permanent magnet of the rotor at the side of the stator, and p is the pole pair number; i.e. i d ，i q Current of the permanent magnet synchronous motor on a d axis and a q axis respectively, v d ，v q The voltages of the permanent magnet synchronous motor on the d axis and the q axis are respectively obtained by Clark conversion and Park conversion of the measured three-phase current and voltage;
s12, establishing a mechanical property equation under a continuous time system under a d-q coordinate system, wherein the equation is shown in the formulas (2) and (3):
T e ＝1.5p[λi q +(L d -L q )i d i q ] (2)
wherein, T e Is electromagnetic torque, J is rotor moment of inertia, B is viscous friction coefficient, T l Is the load torque;
s13, establishing a rotation characteristic equation under a continuous time system, wherein the equation is shown as the formula (4):
wherein, theta m Is the rotor angular position;
s14, adopting i d And (5) according to the vector control strategy of =0, obtaining a state space form of a motor mechanical characteristic equation shown in formula (5) according to the formulas (2), (3) and (4):
wherein X = [ theta ] m ω m ] T Is a state variable, U 2 ＝[i q ]For current control input, Q = [ T = l ]In order to be the load,
3. the inversion control method of a permanent magnet synchronous motor according to claim 2, wherein the step S2 includes:
s21, discretizing the formula (1) by utilizing bilinear transformation to obtain a discrete state transfer equation of the electrical characteristics of the motor shown in the formula (6):
U 2 (k+1)＝A d1 (k)U 2 (k)+B d1 U 1 (k)+F(k) (6)
where k represents the current cycle, k +1 represents the next cycle, i d (k)、i q (k) The currents of the k-period permanent magnet synchronous motor on the d axis and the q axis are respectively omega m (k) For k-period rotor mechanical angular velocity, T s Is a sampling period;
s22, carrying out state transition solution on the formula (5), and utilizing pull type transformation or power series approximation solution to obtain a discrete state transition equation of the mechanical characteristics of the motor shown in the formula (7):
X(k+1)＝A d2 X(k)+B d2 U 2 (k)+H d Q(k) (7)
the load Q = [ T ] with tau as control period and Q (k) as k period l ]The value of (c).
4. The inversion control method of the permanent magnet synchronous motor according to claim 3, wherein the step S3 specifically includes:
s31, calculating k + n according to the formula (7) without considering the load Q (k) of the k period 1 Periodic motor rotation state variable
In the formula (8), the reaction mixture is,
S32, in k +1 to k + n 1 1 cycle without control input and without considering load effect, let k + n 1 Periodic motor rotation reference state is X r (k+n 1 ) Calculating to obtain the motor at k + n 1 Periodic rotation state to obtain k + n 1 Periodic rotational state deviation prediction
5. The inversion control method of the permanent magnet synchronous motor according to claim 4, wherein the step S4 specifically includes:
s41, obtaining a control current instruction sequence set shown in a formula (10) according to the formulas (8) and (9) and a generalized inverse control theory,
in the formula (10), M (k + n) 1 Row full rank of-1, k + 1), M (k + n) 1 -1,k+1) - For its generalized inverse, M (k + n) 1 -1,k+1) - ＝M(k+n 1 -1,k+1) T (M(k+n 1 -1,k+1)M(k+n 1 -1,k+1) T ) -1 ，I 2 Is (n) 1 -1) dimensional unit matrix, V being (n) 1 -1) a dimension arbitrary column vector;
s42, obtaining a control current command sequence by carrying out the unique least square and minimum norm solution in the solution of the formula (10):
s43, under the control current command sequence shown in the formula (11), the predicted value of the deviation of the rotation state of the motor is calculated by considering the action of the load, and the predicted value is shown in the formula (12):
wherein the content of the first and second substances,
s44, designing a current compensation value of a k +1 period according to the formula (12);
wherein, K c ＝diag([k c1 k c2 ])，k c1 k c2 A positive compensation factor; Δ X (j) is a deviation value of a rotation state actually measured in a j period;
s45, obtaining a control voltage command of the k-cycle compensation load according to equations (13) and (6), wherein the control voltage command is:
6. the inversion control method of the permanent magnet synchronous motor according to claim 5, wherein the step S5 specifically includes:
s51, rotor mechanical angular speed omega according to current k period m (k) And equation (6) predicting motor at k + n 2 Periodic current value
n 2 ∈[1,n 1 -1]；
Wherein, the first and the second end of the pipe are connected with each other,
i is a 2-dimensional unit array,
s52, calculating k + n 2 Error value of periodically predicted current value
Wherein, U 2,r (k+n 2 ) A control current command sequence U obtained for the current reference value by equation (11) 2,r (k+n 1 -1, k + 1);
s53, according to
And a generalized inverse control theory, reversely deducing to obtain a control voltage command sequence of k period as shown in a formula (16),
P(k+n 2 -1,k) - is P (k + n) 2 -1, k) generalized inverse;
obtained by the following equation (16)Taking the obtained U 1,r (k+n 2 U in-1,k) 1,r (k) U obtained by adding equation (14) 1，r (k) ' as the actual control voltage input for the k-cycle motor; k is updated to k +1, and the process proceeds to S3.
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| JP2010178529A (en) * | 2009-01-30 | 2010-08-12 | Nippon Yusoki Co Ltd | Control device of induction electric motor |
| CN104485868A (en) * | 2014-12-23 | 2015-04-01 | 河北工业大学 | Predictive control method for current of surface-mounted permanent magnet synchronous motor |
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