CN110995094B - Permanent magnet synchronous motor position and speed estimation method under low carrier ratio control - Google Patents
Permanent magnet synchronous motor position and speed estimation method under low carrier ratio control Download PDFInfo
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- 238000012546 transfer Methods 0.000 claims description 3
- 238000013459 approach Methods 0.000 abstract description 2
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
- H02P21/0007—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control using sliding mode control
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
- H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
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Abstract
The invention discloses a method for estimating the position and the speed of a permanent magnet synchronous motor under the control of a low carrier ratio. The state equation unit is connected with the current comparator, the current comparator is connected with the sign function unit, the sign function unit is connected with band-pass filter I and coefficient unit respectively, the state equation unit is all connected with band-pass filter I and coefficient unit, band-pass filter II is all connected with band-pass filter I and coefficient unit, the phase-locked loop is connected to band-pass filter II. The method aims to reduce buffeting of the sliding mode observer caused by the fact that the permanent magnet synchronous motor controls the two sides of the lower sliding mode surface to approach speed asymmetry at a low carrier ratio, and improves position and speed estimation accuracy.
Description
Technical Field
The invention belongs to the technical field of motor control; in particular to a method for estimating the position and the speed of a permanent magnet synchronous motor under the control of a low carrier ratio.
Background
The position sensorless control technology of the permanent magnet synchronous motor is widely applied to occasions with low cost and high requirements on sensor environment, and the technology is generally divided into a low-speed section estimation method and a medium-high speed section estimation method. The sliding-mode observer is a common method for estimating the position and the speed of a middle-high speed section, extracts the angle and the speed information of a rotor from a motor back electromotive force or a flux linkage related to the angle of the rotor by observing the back electromotive force or the flux linkage of the motor in real time, and has the advantages of simple structure, good robustness and the like. However, a sliding mode surface switching function constructed by the sliding mode observer is usually a sign function, which causes a serious buffeting problem, the approaching speeds of two sides of the sliding mode surface are unequal, and buffeting can be further increased particularly under the control of a low carrier ratio.
Disclosure of Invention
Aiming at the defects of the prior art, the sliding mode observer is improved, and the purpose is to reduce buffeting of the sliding mode observer caused by the fact that the permanent magnet synchronous motor controls the two sides of a sliding mode surface to approach speed asymmetry at a low carrier ratio, and improve position and speed estimation accuracy.
The invention is realized by the following technical scheme:
a permanent magnet synchronous motor position and speed estimation method under low carrier ratio control is realized based on a variable weight coefficient sliding mode observer which comprises a state equation unit, a current comparator, a sign function unit, a coefficient unit, a phase-locked loop, a band-pass filter I and a band-pass filter II,
the state equation unit is connected with the current comparator, the current comparator is connected with the sign function unit, the sign function unit is connected with band-pass filter I and coefficient unit respectively, the state equation unit is all connected with band-pass filter I and coefficient unit, band-pass filter II is all connected with band-pass filter I and coefficient unit, the phase-locked loop is connected to band-pass filter II.
The method for estimating the position and the speed of the permanent magnet synchronous motor under the control of the low carrier ratio comprises the steps that the input of a current state observer is alpha beta shaft voltage and alpha beta shaft control voltage obtained through observation, the difference between the observed current value and the detected current value is obtained through a comparator and is input to a sign function unit to obtain a sliding mode switching variable, the sliding mode switching variable is subjected to partial observation back electromotive force through a band-pass filter I, the sliding mode switching variable and the partial observation back electromotive force are weighted to serve as control voltage of a state equation unit, the control voltage is subjected to a band-pass filter II to obtain a back electromotive force estimated value, and therefore the phase-locked loop is used for obtaining the estimated value of the position and the speed of the motor rotor.
Further, the sliding mode switching variable is weighted with a band-pass filter I to obtain a control voltage, and the sliding mode control law in the variable weight coefficient form adopts the following form:
wherein u iscα、ucβIs a control voltage; k is a radical of1、k2Is a weight coefficient; Δ iα、ΔiβThe alpha and beta axis current deviations, respectively,
alpha and beta axis current observations, i, respectivelyα、iβRespectively detecting alpha and beta axis currents; z is a radical ofα、zβSwitching variables for sliding mode; z is a radical ofFα、zFβIn order to partially observe the back-emf,
wherein the BPF is a band pass filter.
Further, the sliding mode switching variable
eα、eβIs the back electromotive force of the motor;
the transfer function of the bandpass filter is as follows:
in the above formula, kBPFA bandwidth adaptation coefficient for the BPF; omega0Is the passband center angular frequency;
in steady state, the fundamental wave amplitude before and after filtering is considered to be unchanged, and the control voltage is approximate to the actual counter electromotive force, including
Further, when the variable weight coefficient sliding mode observer carries out parameter design, the influence brought by the center frequency error and the rotating speed error of the BPF is considered, and k is2＝ksmo2ωψf，ksmo2For the adaptive coefficient, 0.3 is generally taken, and omega is the electrical angular velocity of the motor, psifIs a permanent magnet flux linkage of the motor.
The invention has the beneficial effects that:
the method is improved into the method that sliding mode switching variables and partial observation back electromotive force phase obtained after filtering are used for weighting, new control voltage is obtained and fed back to the current observer, and therefore a certain weakening effect is achieved on asymmetry of control input of state equations on two sides of a sliding mode surface, the problem of buffeting is solved, and performance of the observer is improved. Compared with the traditional sliding mode controller, the variable weight coefficient sliding mode observer can effectively solve the problem of current distortion observed under the control of a low carrier ratio, improve the observation precision of position and speed and improve the system performance.
Drawings
FIG. 1 is a block diagram of the system of the present invention;
FIG. 2 is a block diagram of a conventional sliding mode observer;
FIG. 3 is a block diagram of a phase locked loop;
FIG. 4 is a schematic diagram of state quantity change tracks on two sides of a sliding mode surface;
FIG. 5 is a structural diagram of a variable weight coefficient sliding mode observer according to the present invention;
FIG. 6 shows the alpha-phase back electromotive force z in the variable weight coefficient sliding mode observer according to the present inventionFαActual counter electromotive force eαAnd a control voltage ucαA schematic diagram;
FIG. 7 shows simulation results of actual stator current, observed stator current and sliding mode control function of a conventional sliding mode observer during low carrier ratio control;
fig. 8 is a measured result of the conventional sliding mode observer in the low carrier ratio control. Wherein, (a) is a current measured value and an observed value; (b) the observed value, the measured value and the observed value of the back electromotive force are obtained; (c) the speed measured value, the observed value and the position and speed observation error are obtained;
fig. 9 is an actual measurement result of the variable weight coefficient sliding mode observer of the present invention during low carrier ratio control. Wherein, (a) is a current measured value and an observed value; (b) the observed value, the measured value and the observed value of the back electromotive force are obtained; (c) the measured values of the velocity, the observed values and the errors of the position and velocity observation.
Detailed Description
The technical solutions in the embodiments of the present invention will be described clearly and completely with reference to the accompanying drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
Fig. 1 is composed of links such as a speed controller, a q-axis current controller, a d-axis current controller, Park (Park) inverse transformation, SVPWM (space vector pulse width modulation), a three-phase inverter, a permanent magnet synchronous motor, Clarke transformation, Park transformation, sliding mode position and speed estimation. The system is a speed and current double closed-loop structure, the outer ring is a rotating speed ring, and the inner ring is a dq-axis (a d axis is a direct axis and a q axis in a motor is a quadrature axis) current ring under vector decoupling. Sliding mode position and speed estimation for estimating rotor position of motor in real time
And velocity
Instead of a mechanical rotor position sensor. Wherein the estimated position
Park transformation and Park inverse transformation, speed for use in vector control systems
As a feedback quantity for the speed loop. The input quantity of sliding mode position and speed estimation is alpha beta axis voltage set value uαAnd uβα β axis current detection value iαAnd iβThe output being an estimate of rotor position
And velocity estimation
A permanent magnet synchronous motor position and speed estimation method under low carrier ratio control is realized based on a variable weight coefficient sliding mode observer which comprises a state equation unit, a current comparator, a sign function unit, a coefficient unit, a phase-locked loop, a band-pass filter I and a band-pass filter II,
the state equation unit is connected with the current comparator, the current comparator is connected with the sign function unit, the sign function unit is connected with band-pass filter I and coefficient unit respectively, the state equation unit is all connected with band-pass filter I and coefficient unit, band-pass filter II is all connected with band-pass filter I and coefficient unit, the phase-locked loop is connected to band-pass filter II.
The method further includes the steps that the input of the current state observer is alpha beta axis voltage and alpha beta axis control voltage obtained through observation, the difference between an observed current value and a detected current value is obtained through a comparator, the current value and the detected current value are input to a sign function unit to obtain a sliding mode switching variable, a part of observed back electromotive force is obtained through the sliding mode switching variable through a band-pass filter I, the sliding mode switching variable and the part of observed back electromotive force are weighted to be used as control voltage of a state equation unit, the control voltage is used for obtaining a back electromotive force estimated value through a band-pass filter II, and therefore estimated values of the position and the speed of a motor rotor are obtained through a phase-locked loop.
Further, the sliding mode switching variable is weighted with a band-pass filter I to obtain a control voltage, and the sliding mode control law in the variable weight coefficient form adopts the following form:
wherein u iscα、ucβIs a control voltage; k is a radical of1、k2Is a weight coefficient; Δ iα、ΔiβThe alpha and beta axis current deviations, respectively,
alpha and beta axis current observations, i, respectivelyα、iβRespectively detecting alpha and beta axis currents; z is a radical ofα、zβSwitching variables for sliding mode; z is a radical ofFα、zFβIn order to partially observe the back-emf,
wherein the BPF is a band pass filter.
Further, the sliding mode switching variable
eα、eβIs the back electromotive force of the motor;
the transfer function of the bandpass filter is as follows:
in the above formula, kBPFA bandwidth adaptation coefficient for the BPF; omega0Is the center of the pass bandAn angular frequency;
in steady state, the fundamental wave amplitude before and after filtering is considered to be unchanged, and the control voltage is approximate to the actual counter electromotive force, including
Further, when the variable weight coefficient sliding mode observer carries out parameter design, the influence brought by the center frequency error and the rotating speed error of the BPF is considered, and k is2＝ksmo2ωψf，ksmo2Taking 0.3 and omega as the electrical angular velocity of the motor psi for the adaptive coefficientfIs a permanent magnet flux linkage of the motor.
As shown in fig. 2, it is composed of a current state observer, a current comparator, a switching function, a low pass filter and a phase locked loop;
the current state observer is implemented in the following way:
wherein, R winding resistance; l is a winding inductance;
respectively are alpha and beta axis current observed values; u. ofα、uβRespectively, alpha and beta axis voltages; z is a radical ofα、zβIs the output of the switching function;
current comparator for observing current
And a detection value iα、iβThe difference, i.e. the current error, is found to be:
the switching function typically employs a sign function, namely:
The low-pass filter (14) typically employs a first-order low-pass filter, namely:
As shown in fig. 6, after the sliding-mode observer with variable weight coefficients is adopted, the input quantity e in the sliding-mode surface state equationx-ucxThe asymmetry of (x ═ α, β) is suppressed to a certain extent, so that the change slopes of the state quantities on both sides of the slip-form surface are approximately equal.
As shown in fig. 7, the variation speed of the observed current is different between the actual value and the actual value due to the asymmetric input amount of the sliding mode surface, so that the current buffeting is increased.
Comparing fig. 8 and fig. 9, it can be seen that, compared with the conventional sliding mode observer, the variable weight coefficient sliding mode observer of the present invention has the advantages that the buffeting of the observed values of the current and the back electromotive force is reduced, the phase shift of the observed values of the position is obviously reduced, the estimation errors of the rotating speed and the position are both reduced, the fluctuation of the rotating speed is reduced, and the estimation accuracy of the position and the speed of the system is improved.
Claims (5)
1. A permanent magnet synchronous motor position and speed estimation method under low carrier ratio control is characterized in that the estimation method is realized based on a variable weight coefficient sliding mode observer, the variable weight coefficient sliding mode observer comprises a state equation unit, a current comparator, a sign function unit, a coefficient unit, a phase-locked loop, a band-pass filter I and a band-pass filter II,
the state equation unit is connected with the current comparator, the current comparator is connected with the sign function unit, the sign function unit is connected with band-pass filter I and coefficient unit respectively, the state equation unit is all connected with band-pass filter I and coefficient unit, band-pass filter II is all connected with band-pass filter I and coefficient unit, the phase-locked loop is connected to band-pass filter II.
2. The method according to claim 1, wherein the current state observer inputs α β axis voltage and α β axis control voltage obtained through observation, an observed current value and a detected current value are subtracted through a comparator and input to a sign function unit to obtain a sliding mode switching variable, the sliding mode switching variable is subjected to a band pass filter i to obtain a part of observed back electromotive force, the sliding mode switching variable and the part of observed back electromotive force are weighted to be used as control voltage of a state equation unit, and the control voltage is subjected to a band pass filter ii to obtain a back electromotive force estimated value, so that a phase-locked loop is used to obtain an estimated value of a position and a speed of a motor rotor.
3. The method for estimating the position and the speed of the permanent magnet synchronous motor under the control of the low carrier ratio according to claim 2, wherein the sliding mode switching variable is weighted with a band-pass filter I to obtain a control voltage, and the sliding mode control law in the form of the variable weight coefficient is in the following form:
wherein u iscα、ucβIs a control voltage; k is a radical of1、k2Is a weight coefficient; Δ iα、ΔiβThe alpha and beta axis current deviations, respectively,
alpha and beta axis current observations, i, respectivelyα、iβRespectively detecting alpha and beta axis currents; z is a radical ofα、zβSwitching variables for sliding mode; z is a radical ofFα、zFβIn order to partially observe the back-emf,
wherein the BPF is a band pass filter.
4. The method according to claim 3, wherein the sliding mode switching variable is variable
eα、eβIs the back electromotive force of the motor;
the transfer function of the bandpass filter is as follows:
in the above formula, kBPFA bandwidth adaptation coefficient for the BPF; omega0Is the passband center angular frequency;
in steady state, the fundamental wave amplitude before and after filtering is considered to be unchanged, and the control voltage is approximate to the actual counter electromotive force, including
5. The method for estimating the position and the speed of the permanent magnet synchronous motor under the control of the low carrier ratio according to claim 2, wherein the variable weight coefficient sliding mode observer considers the influence of the BPF center frequency error and the rotating speed error in parameter design, k2＝ksmo2ωψf，ksmo2Taking 0.3 and omega as the electrical angular velocity of the motor psi for the adaptive coefficientfIs a permanent magnet flux linkage of the motor.
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