CN109905158B - Design method for self-adaptive beam forming optimal weight of uniform power broadband signal - Google Patents
Design method for self-adaptive beam forming optimal weight of uniform power broadband signal Download PDFInfo
- Publication number
- CN109905158B CN109905158B CN201910190116.9A CN201910190116A CN109905158B CN 109905158 B CN109905158 B CN 109905158B CN 201910190116 A CN201910190116 A CN 201910190116A CN 109905158 B CN109905158 B CN 109905158B
- Authority
- CN
- China
- Prior art keywords
- array
- signal
- array element
- broadband
- interference
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
Images
Classifications
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
- Y02D30/00—Reducing energy consumption in communication networks
- Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks
Abstract
The invention discloses a design method of an optimal weight value of self-adaptive beam forming of a uniform power broadband signal, and relates to a design method of an optimal weight value of self-adaptive digital beam forming. The invention aims to solve the problems that the existing broadband signal can affect the beam pointing of beam forming when the bandwidth is large, and can affect the adaptive interference suppression pointing, so that the angle error is large. The specific process is as follows: step one, constructing an expression of a uniform power broadband signal in any array; step two, applying the expression of the uniform power broadband signal in any array in the step one to a uniform linear phased array to obtain a covariance matrix of a useful signal, an interference signal and a noise signal; and step three, substituting the covariance matrixes of the useful signals, the interference signals and the noise signals obtained in the step two into a linear constraint minimum variance criterion to obtain the optimal self-adaptive weight. The invention is used for the field of optimal weight design of broadband signal adaptive beamforming.
Description
Technical Field
The invention relates to a design method of an optimal weight value of self-adaptive digital beam forming.
Background
The Digital phased array radar perfectly combines a Digital technology and an antenna technology, replaces the traditional analog Beam Forming by Digital Beam Forming (DBF) in a transmitting and receiving mode, and has the advantages which cannot be compared with the traditional radar. Meanwhile, in order to solve the problems of high distance resolution, high ranging precision, improvement of the target resolution of the radar, improvement of radar anti-interference and the like, the signal transmitted by the radar is a signal with large instantaneous bandwidth. If the phased array radar adopts broadband signals, the function and the performance of the phased array radar are greatly improved. For example, broadband signals are adopted in the early warning machine, so that the anti-interference performance of the early warning machine can be greatly improved, and the detection capability of the early warning machine under strong interference is improved. Meanwhile, the broadband can provide more target information, and the combination of air detection and ground detection and the combination of target detection and multi-target tracking are realized.
The prior art is mainly suitable for narrow-band signals, and the bandwidth B ratio of the signals is considered in the narrow-band signalsCarrier frequency f0The difference of signal envelopes on each array element is ignored in practical application; in addition, the signal bandwidth is small, so that the beam forming pointing direction and the angle of the adaptive interference suppression pointing direction cannot be influenced. However, for broadband signals, the difference of the signal envelopes on each array element cannot be ignored, and when the bandwidth is large, the beam pointing and adaptive interference suppression pointing errors of a directional diagram are large. When the signal is a carrier frequency f0When the broadband signal is 1.6GHZ and the bandwidth B is 200MHZ, the frequency f is f0-B/2，f0，f0The beam pattern at + B/2 is shown in FIG. 1.
As can be seen from fig. 1, the main beam deviation is about 6 °, and the adaptive interference suppression differs by about 10 °, which causes a large error.
Disclosure of Invention
The invention aims to solve the problem that beam pointing and self-adaptive interference suppression pointing errors of a directional diagram are large when the bandwidth is large for broadband signals in the prior art, and provides a design method for forming optimal weights by using broadband signals with uniform power in a self-adaptive beam mode.
The specific process of the design method for the uniform power broadband signal self-adaptive beam forming optimal weight is as follows:
step one, constructing an expression of a uniform power broadband signal in any array;
step two, applying the expression of the uniform power broadband signal in any array in the step one to a uniform linear phased array to obtain a covariance matrix of a useful signal, an interference signal and a noise signal;
and step three, substituting the covariance matrixes of the useful signals, the interference signals and the noise signals obtained in the step two into a linear constraint minimum variance criterion to obtain the optimal self-adaptive weight.
The invention has the beneficial effects that:
the invention applies the self-adaptive digital beam forming technology at the receiving end of the digital phased array system, reduces the beam pointing error of a directional diagram and the self-adaptive interference suppression pointing error, and completes the self-adaptive digital beam forming of broadband signals.
The method can realize the self-adaptive digital beam forming of the broadband signal with uniform power at the receiving end of the digital phased array radar, and form the self-adaptive null in the interference direction, so that the beam pointing direction of a directional diagram and the self-adaptive interference suppression error are reduced. Fig. 2 is a beam pattern after the present invention is used, and it can be seen from the figure that all beams point to the incoming wave direction of the signal, and the null error of the interference direction can be ignored, so that the present invention has a better correction effect.
As shown in the first embodiment, the main beams of the existing narrowband method beam forming are respectively directed at 27 °, 30 °, 33 °, the maximum deviation is about 6 °, interference nulls are formed at 45 °, 50 °, 55 °, and the maximum deviation is about 10 °.
The main beam of the beam forming method of the invention is totally pointed at 30 degrees, interference null is formed at 49 degrees, 50 degrees and 51 degrees, broadband beams are effectively formed, the main beam is improved by about 6 degrees, and the interference null is improved by about 10 degrees.
Drawings
FIG. 1 is a beam pattern;
FIG. 2 is a modified beam pattern of the present invention;
FIG. 3 is a diagram of a uniform linear array of the present invention;
FIG. 4 is a diagram of a uniform linear receive phased array of the present invention;
FIG. 5 is a flow chart of the optimal adaptive weight w according to the present invention.
Detailed Description
The first embodiment is as follows: the specific process of the design method for forming the optimal weight by the uniform power broadband signal self-adaptive beam in the embodiment is as follows:
step one, constructing an expression of a uniform power broadband signal in any array;
step two, applying the expression of the uniform power broadband signal in any array in the step one to a uniform linear phased array to obtain a covariance matrix of a useful signal, an interference signal and a noise signal;
and step three, substituting the covariance matrixes of the useful signals, the interference signals and the noise signals obtained in the step two into a linear constraint minimum variance criterion to obtain the optimal self-adaptive weight.
The second embodiment is as follows: the first embodiment is different from the first embodiment in that an expression of a uniform power broadband signal in any array is constructed in the first step; the specific process is as follows:
assuming that a uniform power broadband signal source is s (t), the bandwidth of the uniform power broadband signal is B, and the center frequency of the uniform power broadband signal is f0And the power spectral density of s (t) is subject to a uniform distribution, i.e. the power spectral density function of the signal s (t) is ps(f)；
Let an array (this array can be any array, array of uniform power spectral density broadband signals) have N array elements, the first array element is located at the origin of the rectangular coordinate system, and the coordinate of any array element in the array can be expressed as zi＝(xi,yi)，
Wherein i is 1,2, …, N, xiIs the abscissa, y, of the array elementiIs the ordinate of the array element;
when the wideband signal source s (t) is in the far field, θ is the propagation direction angle, and the beam vector can be expressed as (cos θ, sin θ)T；
Wherein T is transposition;
the delay time tau of any array element in the array relative to the first array elementi＝(ziU)/c (dot product is x)icosθ+yisin θ), i is 1,2, …, N, so that the time difference of the uniform power broadband signal incident on each array element and the first array element is τ [ τ ═ t [ ]1,…,τi,…,τN]，i＝1,2,…,N；
Wherein c is the speed of light;
when no noise exists, the output signal of the ith array element is expressed as
yi(t)＝si(t-τi) i＝1,2,…,N (2)
In the formula si(t-τi) An expression of the broadband signal source incident to the ith array element;
at the moment, the uniform power broadband signal passes through the array and then the output vector is y (t) ═ y1(t),…,yi(t),…,yN(t)]Wherein y (t) is expressed as Kramer since s (t) is a uniform power broadband signal source
In the formula, y1(t) is the output of the first array element of the array, yi(t) is the output of the ith array element of the array, yN(t) is the output of the Nth array element of the array, N is a variable, m is a variable, j is an imaginary number, j is a variable2＝-1；
Writing formula (3) to baseband form then represents
Dividing the uniform power broadband signal into L segments of sub-bands on the bandwidth B, and weighting each sub-band, the formula (4) can be written as
Wherein n islThe frequency of the l segment sub-band; w is alIs the weight of the l-th segment of sub-band.
Other steps and parameters are the same as those in the first embodiment.
The third concrete implementation mode: in this embodiment, which differs from the first or second embodiment, the power spectral density function p of the signal s (t)s(f) The expression is as follows:
wherein t is a time variable, ps(f) Is a function of the power spectral density of signal s (t), f is a frequency variable, and p is the power spectral density of signal s (t).
Other steps and parameters are the same as those in the first or second embodiment.
The fourth concrete implementation mode: this embodiment is different from the first to third embodiments in that the frequency of the l-th sub-band
Other steps and parameters are the same as those in one of the first to third embodiments.
The fifth concrete implementation mode: the difference between this embodiment and the first to the fourth embodiments is that the weight w of the l-th segment of sub-bandl＝0.5-0.5cos(2πl/L-1)，l＝1,2,…,L。
Other steps and parameters are the same as in one of the first to fourth embodiments.
The sixth specific implementation mode: the second step is to apply the expression of the uniform power broadband signal in any array in the first step to a uniform linear phased array to obtain a covariance matrix of a useful signal, an interference signal and a noise signal; the specific process is as follows:
the covariance matrix of the broadband useful signal can therefore be expressed as
R1＝E[d(t-τ)d(t-τ)H] (13)
The covariance matrix of the wideband interfering signal may be expressed as
R2＝E[k(t-τ)k(t-τ)H] (14)
The covariance matrix of the noise signal n (t) can be expressed as
Rn＝E[n(t)n(t)H] (15)
Wherein E [ ] is a covariance matrix, and H is a conjugate transpose; d (t-tau) is an M x 1-dimensional vector formed by useful signals received by all array elements of the array, k (t-tau) is an M x 1-dimensional vector formed by interference signals received by all array elements of the array, and n (t) is white Gaussian additive noise at the array element level.
Other steps and parameters are the same as those in one of the first to fifth embodiments.
The seventh embodiment: the difference between this embodiment and the first to sixth embodiments is that the covariance matrix of the useful signal, the interference signal, and the noise signal is specifically solved by:
according to the invention, a uniform linear phased array is adopted, an M-array element uniform linear array structure is provided as shown in figure 3, each antenna array element is an omnidirectional array element and is positioned on an X axis of a rectangular coordinate system, a first array element positioned at an origin is a reference array element, and the distance between any two adjacent array elements is d, wherein d is lambda/2, and the array length is L is (M-1) d; λ is the signal wavelength;
for a uniform linear array, only one-dimensional scanning can be done at the pitch angle, and no scanning at the azimuth angle, so the pattern in the theta direction is of interest only.
Considering that the far field has a broadband useful signal d (t) and a broadband interference signal k (t), the useful signal and the interference signal are respectively in thetadAnd thetakThe angle is simultaneously incident to each array element, and the space delay distance between any two adjacent array elements on the linear array is delta d sin thetadThe corresponding delay time τ is d sin θdSo that the time difference of the useful signal and the interference signal incident on each array element and the reference array element is tau ═ tau1,…,τi,…,τM]I ═ 1,2, …, M; (the time difference between the desired signal and the interfering signal is the same)
d is the array element spacing;
at this time, the output of the array element is
x(t)＝[x1(t),…,xi(t),…,xM(t)]T (6)
Wherein
x(t)＝d(t-τ)+k(t-τ)+n(t) (7)
Wherein x is1(t) signals received for the first array element of the array, xi(t) is the signal received by the ith array element of the array, xM(t) is a signal received by the Mth array element of the array, d (t-tau) is an M multiplied by 1 dimensional vector formed by useful signals received by all array elements of the array, k (t-tau) is an M multiplied by 1 dimensional vector formed by interference signals received by all array elements of the array, n (t) is array element-level Gaussian additive white noise, and noise in each channel is independent of each otherAnd the signals on different array elements are mutually independent;
that is, y (t) represents an expression after any array receives the broadband signal, and this expression can represent any broadband signal (with uniform power) and any array, and is generalized, while x (t) gives the above generalized specific, specifically linear array, and the array receives the broadband signal, the broadband interference and the noise at the same time, in fact, y (t) above represents the general case, and x (t) represents the actual case;
namely, it is
d(t-τ)＝[d(t-τ1),…,d(t-τi),…,d(t-τN)]T (8)
k(t-τ)＝[k(t-τ1),…,k(t-τi),…,k(t-τN)]T (9)
n(t)＝[n1(t),…,ni(t),…,nN(t)]T (10)
Wherein d (t- τ)1) For the useful signal received at the first array element of the array, d (t- τ)i) The useful signal received for the ith array element of the array, d (t- τ)M) For the useful signal received by the Mth array element of the array, k (t- τ)1) For interfering signals received by the first array element of the array, k (t- τ)i) Interference signal, k (t- τ), received for the ith array element of the arrayN) Interference signal received for the Mth array element of the array, n1(t) noise signals received by the first array element of the array, ni(t) noise signals received by the ith array element of the array, nM(t) noise signals received by the mth array element of the array;
the d (t-tau) and k (t-tau) signals are broadband signals and can be expressed as
Wherein, wqThe weight of the Q-th sub-band is 1,2, …, Q is the number of sub-bands of the interference signal, f1Carrier frequency, n, of interference signal k (t)qIs the frequency of the q-th segment of the sub-band,
step one, expression of uniform power broadband signals in any array
Applicable to any situation, two steps
An M multiplied by 1 dimensional vector d (t-tau) formed by useful signals received by all array elements of the array is expressed as d (t-tau) ═ d (t-tau)1),…,d(t-τi),…,d(t-τN)]TAn M × 1-dimensional vector k (t- τ) formed by interference signals received by all array elements of the array is represented by k (t- τ) ═ k (t- τ)1),…,k(t-τi),…,k(t-τN)]TArray element level white gaussian noise n (t) is expressed as n (t) ═ n1(t),…,ni(t),…,nN(t)]T；
The covariance matrix of the broadband useful signal can therefore be expressed as
R1＝E[d(t-τ)d(t-τ)H] (13)
The covariance matrix of the wideband interfering signal may be expressed as
R2＝E[k(t-τ)k(t-τ)H] (14)
The covariance matrix of the noise signal n (t) can be expressed as
Rn＝E[n(t)n(t)H] (15)
Wherein E [ ] is a covariance matrix, and H is a conjugate transpose;
other steps and parameters are the same as those in one of the first to sixth embodiments.
The specific implementation mode eight: the difference between the present embodiment and one of the first to seventh embodiments is that in the third step, the covariance matrix of the useful signal, the interference signal, and the noise signal obtained in the second step is substituted into the linear constraint minimum variance criterion to obtain the optimal adaptive weight; the specific process is as follows:
design of self-adaptive weight
The receive array is shown in fig. 4, where phase shifters for each array element are used to control beam pointing. In order to adaptively suppress interference, signals need to be weighted, and LCMV (Linear Constrained Minimum Variance) criterion is adopted for designing an adaptive weight in the invention; that is, the signal-to-interference-and-noise ratio (the ratio of the signal to the interference plus noise power) of the system output is maximized, i.e., the optimal adaptive interference suppression performance is achieved.
Let the beam of the array point to theta0The optimal adaptive weight vector w is a column vector with M dimension, and the optimal adaptive weight vector w is obtained based on LCMV criterion, that is
Wherein μ is any constant; α (f, θ)0) A steering vector for the array;
order to
Wherein
Spherical wave received by the ith array element, f is working frequency, then
where K is the number of samples, will be α (f, θ)0) And
and (5) carrying in (16), namely obtaining the optimal adaptive weight w for the digital beam forming of the broadband signal with uniform power spectral density.
In the present invention, a flow chart for determining the optimal adaptive weight w is shown in fig. 5.
Other steps and parameters are the same as those in one of the first to seventh embodiments.
The following examples were used to demonstrate the beneficial effects of the present invention:
the first embodiment is as follows: the parameters used in the simulation were as follows: total array element number 32 of uniform linear array, useful signal center frequency f0The bandwidth is 200MHz at 1.6GHz, the array is uniformly weighted, the incoming wave direction is 30 degrees, and the interference signal with the bandwidth of 200MHz is added in the 50 degrees direction, and the number of data sampling points is assumed to be 100 in order to estimate the covariance matrix in the adaptive algorithm.
The main beams of the existing narrow-band method beam forming are respectively directed at 27 degrees, 30 degrees and 33 degrees, the maximum deviation is about 6 degrees, interference null is formed at 45 degrees, 50 degrees and 55 degrees, and the maximum deviation is about 10 degrees.
The main beam of the beam forming method of the invention is totally pointed at 30 degrees, interference null is formed at 49 degrees, 50 degrees and 51 degrees, broadband beams are effectively formed, the main beam is improved by about 6 degrees, and the interference null is improved by about 10 degrees.
The present invention is capable of other embodiments and its several details are capable of modifications in various obvious respects, all without departing from the spirit and scope of the present invention.
Claims (4)
1. The design method of the optimal weight of the uniform power broadband signal self-adaptive beam forming is characterized in that: the method comprises the following specific processes:
step one, constructing an expression of a uniform power broadband signal in any array;
step two, applying the expression of the uniform power broadband signal in any array in the step one to a uniform linear phased array to obtain a covariance matrix of a useful signal, an interference signal and a noise signal;
step three, substituting the covariance matrixes of the useful signals, the interference signals and the noise signals obtained in the step two into a linear constraint minimum variance criterion to obtain an optimal self-adaptive weight;
constructing an expression of the uniform power broadband signal in any array in the first step; the specific process is as follows:
assuming that a uniform power broadband signal source is s (t), the bandwidth of the uniform power broadband signal is B, and the center frequency of the uniform power broadband signal is f0And the power spectral density of s (t) is subject to a uniform distribution, i.e. the power spectral density function of the signal s (t) is ps(f)；
Setting N array elements in an array, wherein the first array element is positioned at the origin of a rectangular coordinate system, and the coordinate of any array element in the array is expressed as zi＝(xi,yi)，
Wherein i is 1,2, …, N, xiIs the abscissa, y, of the array elementiIs the ordinate of the array element;
when the broadband signal source s (t) is in the far field, θ is the propagation direction angle, and the beam vector is expressed as (cos θ, sin θ)T；
Wherein T is transposition;
the delay time tau of any array element in the array relative to the first array elementi＝(ziU)/c, i is 1,2, …, N, so that the time difference of the uniform power broadband signal incident on each array element from the first is τ ═ N1,…,τi,…,τN]，i＝1,2,…,N；
Wherein c is the speed of light;
when no noise exists, the output signal of the ith array element is expressed as
yi(t)＝si(t-τi) i＝1,2,…,N (2)
In the formula, si(t-τi) An expression of the broadband signal source incident to the ith array element;
at the moment, the uniform power broadband signal passes through the array and then the output vector is y (t) ═ y1(t),…,yi(t),…,yN(t)]Wherein y (t) is expressed as Kramer since s (t) is a uniform power broadband signal source
In the formula, y1(t) is the output of the first array element of the array, yi(t) is the output of the ith array element of the array, yN(t) is the output of the Nth array element of the array, N is a variable, m is a variable, j is an imaginary number, j is a variable2＝-1；
Writing formula (3) to baseband form then represents
Dividing the uniform power broadband signal into L sections of sub-bands on the bandwidth B, and respectively weighting each sub-band, then the formula (4) is written as
Wherein n islThe frequency of the l segment sub-band; w is alThe weight of the first segment of sub-band;
in the second step, the expression of the uniform power broadband signal in any array in the first step is applied to a uniform linear phased array to obtain a covariance matrix of a useful signal, an interference signal and a noise signal; the specific process is as follows:
the covariance matrix of the broadband useful signal is therefore expressed as
R1＝E[d(t-τ)d(t-τ)H] (13)
The covariance matrix of the broadband interference signal is expressed as
R2＝E[k(t-τ)k(t-τ)H] (14)
The covariance matrix of the noise signal n (t) is expressed as
Rn＝E[n(t)n(t)H] (15)
Wherein E [ ] is a covariance matrix, and H is a conjugate transpose; d (t-tau) is an M x 1-dimensional vector formed by useful signals received by all array elements of the array, k (t-tau) is an M x 1-dimensional vector formed by interference signals received by all array elements of the array, and n (t) is array element-level Gaussian additive white noise;
the specific solving process of the covariance matrix of the useful signal, the interference signal and the noise signal is as follows:
the antenna array is provided with an M-array element uniform linear array, each antenna array element is an omnidirectional array element and is positioned on an X axis of a rectangular coordinate system, the first array element positioned at an origin is a reference array element, and the distance between any two adjacent array elements is d, wherein d is lambda/2, and the array length is L is (M-1) d; λ is the signal wavelength;
the far field has a broadband useful signal d (t) and a broadband interference signal k (t), the useful signal and the interference signal are respectively in thetadAnd thetakThe angle is simultaneously incident to each array element, and the space delay distance between any two adjacent array elements on the linear array is delta d sin thetadThe corresponding delay time τ is d sin θdSo that the time difference of the useful signal and the interference signal incident on each array element and the reference array element is tau ═ tau1,…,τi,…,τM]，i＝1,2,…,M；
d is the array element spacing;
at this time, the output of the array element is
x(t)＝[x1(t),…,xi(t),…,xM(t)]T (6)
Wherein
x(t)＝d(t-τ)+k(t-τ)+n(t) (7)
Wherein x is1(t) signals received for the first array element of the array, xi(t) is the signal received by the ith array element of the array, xM(t) is a signal received by the Mth array element of the array, d (t-tau) is an M multiplied by 1 dimensional vector formed by useful signals received by all array elements of the array, k (t-tau) is an M multiplied by 1 dimensional vector formed by interference signals received by all array elements of the array, and n (t) is array element-level Gaussian additive white noise;
namely that
d(t-τ)＝[d(t-τ1),…,d(t-τi),…,d(t-τN)]T (8)
k(t-τ)＝[k(t-τ1),…,k(t-τi),…,k(t-τN)]T (9)
n(t)＝[n1(t),…,ni(t),…,nN(t)]T (10)
Wherein d (t- τ)1) For the useful signal received at the first array element of the array, d (t- τ)i) The useful signal received for the ith array element of the array, d (t- τ)M) The useful signal, k (t- τ), received for the Mth array element of the array1) For interfering signals received by the first array element of the array, k (t- τ)i) For interfering signals received by the ith array element of the array, k (t- τ)N) Interference signal received for the Mth array element of the array, n1(t) noise signals received by the first array element of the array, ni(t) is the noise signal received by the ith array element of the array, nM(t) noise signals received by the mth array element of the array;
the d (t-tau), k (t-tau) signals are all broadband signals, denoted as
Wherein, wqThe weight of the Q-th sub-band is 1,2, …, Q is the number of sub-bands of the interference signal, f1Carrier frequency, n, of interference signal k (t)qIs the frequency of the q-th segment of the sub-band,
the covariance matrix of the broadband useful signal is therefore expressed as
R1＝E[d(t-τ)d(t-τ)H] (13)
The covariance matrix of the broadband interference signal is expressed as
R2＝E[k(t-τ)k(t-τ)H] (14)
The covariance matrix of the noise signal n (t) is expressed as
Rn＝E[n(t)n(t)H] (15)
Wherein E [ ] is covariance matrix, H is conjugate transpose;
substituting the covariance matrix of the useful signals, the interference signals and the noise signals obtained in the second step into a linear constraint minimum variance criterion to obtain an optimal self-adaptive weight; the specific process is as follows:
let the beam of the array point to theta0The optimal adaptive weight vector w is a column vector with M dimension, and the optimal adaptive weight vector w is obtained based on LCMV criterion, that is
Wherein μ is any constant; α (f, θ)0) A steering vector for the array;
order to
Wherein
Spherical wave received by the ith array element, f is working frequency, then
where K is the number of samples, will be alpha (f, theta)0) And
and (5) carrying in (16), namely obtaining the optimal adaptive weight w for the digital beam forming of the broadband signal with uniform power spectral density.
2. The method for designing optimal weights for adaptive beamforming of uniform power broadband signals according to claim 1, wherein: the power spectral density function of said signal s (t) is ps(f)；
Wherein t is a time variable, ps(f) Is a function of the power spectral density of signal s (t), f is a frequency variable, and p is the power spectral density of signal s (t).
3. The method for designing optimal weights for adaptive beamforming of uniform power broadband signals according to claim 2, wherein: frequency of the l-th sub-band
4. The method according to claim 3, wherein the method for designing optimal weights for adaptive beamforming of the uniform power broadband signal comprises: the weight w of the l-th sub-bandl＝0.5-0.5cos(2πl/L-1)，l＝1,2,…,L。
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201910190116.9A CN109905158B (en) | 2019-03-13 | 2019-03-13 | Design method for self-adaptive beam forming optimal weight of uniform power broadband signal |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201910190116.9A CN109905158B (en) | 2019-03-13 | 2019-03-13 | Design method for self-adaptive beam forming optimal weight of uniform power broadband signal |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN109905158A CN109905158A (en) | 2019-06-18 |
| CN109905158B true CN109905158B (en) | 2022-05-20 |
Family
ID=66952149
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201910190116.9A Active CN109905158B (en) | 2019-03-13 | 2019-03-13 | Design method for self-adaptive beam forming optimal weight of uniform power broadband signal |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN109905158B (en) |
Families Citing this family (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110365437B (en) * | 2019-07-30 | 2021-05-14 | 电子科技大学 | Fast power spectrum estimation method based on sub-Nyquist sampling |
| CN110673116B (en) * | 2019-10-25 | 2022-05-10 | 海鹰企业集团有限责任公司 | Same frequency interference suppression method |
| CN111224704B (en) * | 2019-11-12 | 2022-10-11 | 电子科技大学 | Distributed self-adaptive reduced rank beam forming method |
| CN111859644B (en) * | 2020-07-07 | 2023-10-20 | 南京国睿防务系统有限公司 | Method for forming receiving wave beam and scanning azimuth of circular phased array antenna |
Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102195701A (en) * | 2011-01-27 | 2011-09-21 | 哈尔滨工业大学 | Method for suppressing side lobes of sum beams and difference beams of planar phased array only by utilizing one kind of analogue weighting |
| CN104506256A (en) * | 2014-12-24 | 2015-04-08 | 南京信息工程大学 | Performance evaluation method for MIMO (Multiple Input Multiple Output) multi-antenna system and multi-antenna system |
| CN104868946A (en) * | 2015-06-12 | 2015-08-26 | 哈尔滨工业大学 | Adaptive weighted interference suppression method of subarray level mixed MIMO-phased array system |
| CN106788653A (en) * | 2016-11-25 | 2017-05-31 | 西安电子科技大学 | A kind of Adaptive beamformer method based on covariance matrix reconstruct |
Family Cites Families (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FI20020461A0 (en) * | 2002-03-12 | 2002-03-12 | Nokia Corp | Communication method and system |
-
2019
- 2019-03-13 CN CN201910190116.9A patent/CN109905158B/en active Active
Patent Citations (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102195701A (en) * | 2011-01-27 | 2011-09-21 | 哈尔滨工业大学 | Method for suppressing side lobes of sum beams and difference beams of planar phased array only by utilizing one kind of analogue weighting |
| CN104506256A (en) * | 2014-12-24 | 2015-04-08 | 南京信息工程大学 | Performance evaluation method for MIMO (Multiple Input Multiple Output) multi-antenna system and multi-antenna system |
| CN104868946A (en) * | 2015-06-12 | 2015-08-26 | 哈尔滨工业大学 | Adaptive weighted interference suppression method of subarray level mixed MIMO-phased array system |
| CN106788653A (en) * | 2016-11-25 | 2017-05-31 | 西安电子科技大学 | A kind of Adaptive beamformer method based on covariance matrix reconstruct |
Non-Patent Citations (2)
| Title |
|---|
| Sidelobe Suppression Approach at Subarray Level for Sum and Difference Beam without Analog Weighting;Hu Hang等;《Proceedings of 2012 5th Global Symposium on Millimeter-Waves》;20120927;全文 * |
| 均匀圆阵波束空间二维DOA估计算法;胡航等;《哈尔滨工业大学学报》;20040630;第36卷(第6期);全文 * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN109905158A (en) | 2019-06-18 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN109905158B (en) | Design method for self-adaptive beam forming optimal weight of uniform power broadband signal | |
| CN108508423B (en) | Subarray digital sum and difference monopulse angle measurement method based on special-shaped array | |
| US7511665B2 (en) | Method and apparatus for a frequency diverse array | |
| CN109407055B (en) | Beam forming method based on multipath utilization | |
| CN111624565B (en) | Multi-area joint beam tracking method for large conformal phased array system | |
| CN113866709A (en) | Phase control array cross multi-beam amplitude comparison direction finding method | |
| CN109143230B (en) | Two-dimensional direction of arrival estimation method applied to S-band through-wall radar | |
| CN112612005A (en) | Radar main lobe interference resisting method based on deep learning | |
| CN113314832A (en) | Millimeter wave vehicle-mounted MIMO radar antenna array device and design method | |
| KR101807499B1 (en) | Apparatus for estimating direction of arrival based on a circularly arraying antenna compensating intermutual interference and method therefor | |
| WO2019200663A1 (en) | Beam control method based on frequency diversity array antenna | |
| CN111693979B (en) | Digital array monopulse angle measurement method based on Taylor expansion | |
| CN110850383B (en) | MIMO radar signal processing method based on conformal array | |
| US11152986B2 (en) | Fast spatial search using phased array antennas | |
| CN111817766A (en) | Beam forming method of combined array antenna unit directional diagram | |
| CN100380134C (en) | Small size antenna array aperture expanding and space signal processing method | |
| CN109239678A (en) | A kind of portable high frequency groundwave radar radio frequency interference suppressing method | |
| CN114814739A (en) | Main lobe interference suppression method based on distributed FDA-MIMO radar | |
| CN111585632B (en) | Broadband self-adaptive beam forming method based on interference suppression model optimization | |
| Deng et al. | Frequency diverse arc array beampattern synthesis analysis with nonlinear frequency offset | |
| CN114488027A (en) | Wave beam zero setting and widening method of conformal array antenna | |
| Zhou et al. | Research on interferometer direction finding technology based on digital beam forming | |
| CN112711018A (en) | Near-field space mesh structure focusing beam forming method | |
| CN110954887B (en) | Phased array MIMO beam forming method based on spherical invariant constraint and antisymmetry | |
| CN114361815B (en) | Use method of sum-difference double-channel sidelobe suppression phased array antenna system |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |