CN109639151A - Constant-current control circuit and constant current control method for LLC resonant converter - Google Patents
Constant-current control circuit and constant current control method for LLC resonant converter Download PDFInfo
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- CN109639151A CN109639151A CN201910054905.XA CN201910054905A CN109639151A CN 109639151 A CN109639151 A CN 109639151A CN 201910054905 A CN201910054905 A CN 201910054905A CN 109639151 A CN109639151 A CN 109639151A
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/338—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement
- H02M3/3385—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement with automatic control of output voltage or current
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- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
- H05B45/30—Driver circuits
- H05B45/37—Converter circuits
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps
- Y02B20/30—Semiconductor lamps, e.g. solid state lamps [SSL] light emitting diodes [LED] or organic LED [OLED]
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Dc-Dc Converters (AREA)
Abstract
This application discloses a kind of constant-current control circuits and constant current control method for LLC resonant converter.The LLC resonant converter includes the first bipolar junction transistor and the second bipolar junction transistor to be worked using self-oscillation mode, the constant-current control circuit includes: switch element, for being shorted the driving current of at least one first bipolar junction transistor and second bipolar junction transistor；And drive module, including exporting current calculation module, for calculate the difference of resonance current signal and the first transformer magnetizing current signal absolute value average value as thermal compensation signal, and the on state of the switch element according to compensation signal control, to realize current constant control.In the complicated circuit for constituting charge pump PFC module and LLC resonant transformation combined application, which still can be improved constant-current control accuracy.
Description
Technical field
The present invention relates to power technique fields, more particularly, to for LLC resonant converter constant-current control circuit and
Constant current control method.
Background technique
LED drive circuit is used to provide average anode current to LED light, shines so that LED light is lighted to as illumination
Light source.The Specifeca tion speeification of LED drive circuit includes power factor (PF) and output current ripple.Power factor characterization is active
The ratio of power and reactive power.The AC compounent of output current ripple characterization average anode current.For example, the AC compounent is
Power frequency component, it will the stroboscopic for leading to LED light not only influences illuminating effect, but also influences the service life of LED light.LED drive circuit
High Power Factor utilization rate of electrical can be improved, low output current ripple can reduce stroboscopic.
High Power Factor and low output current ripple in order to balance, LED drive circuit can use a variety of cascade circuits
Scheme, comprising: the first kind concatenated schemes of single-stage inverse-excitation type primary-side-control constant-current system framework and the ripple circuit composition that disappears；It rises
The Second Type concatenated schemes of pressure topology and inverse-excitation type primary-side-control constant current topology composition；Boost topology and resonance oscillation semi-bridge LLC knot
The third type concatenated schemes of structure composition；4th type cascade side of charge pump PFC module and resonance oscillation semi-bridge LLC structure composition
Case.
The circuit arrangement of four seed types can realize High Power Factor (PF) and low output current ripple (nothing simultaneously above
Stroboscopic).However, the shortcomings that first kind concatenated schemes be disappear ripple circuit on system effectiveness influence it is very big, especially when resonance is defeated
When voltage is relatively low out.The shortcomings that Second Type concatenated schemes is that two-step scheme systematic comparison is complicated, and system cost is higher,
In addition EMI debugging is relatively difficult, and efficiency is not also high.The efficiency of third type concatenated schemes and the 4th type concatenated schemes is than second
Type concatenated schemes it is high-efficient, but system is more complicated and cost is higher.
In the concatenated schemes of the 4th type, mode of resonance switch converters are to obtain square-wave voltage using switching tube and adopt
Resonance is carried out with resonant tank to realize the power inverter of energy transmission.LLC resonant converter has higher power density
And less electronic component quantity, while possessing smooth current waveform, be conducive to improve electromagnetic interference, and can be whole
The zero voltage switching (Zero Voltage Switching, ZVS) and zero current switching of switching tube are realized in a range of operation
(Zero Current Switching, ZCS), helps to obtain high efficiency.Further, on LLC half-bridge driven
Increase the passive PFC of current-type charge pump and the passive PFC combination of voltage-type charge pump, can obtain very high power factor (PF) and
Very low total harmonic distortion (THD).Therefore, the concatenated schemes of the 4th type have apparent advantage in terms of circuit efficiency.
Further, expect to take into account the raising of circuit efficiency and the drop of circuit cost in the concatenated schemes of the 4th type
It is low.
Summary of the invention
In view of the above problems, the application provides the constant-current control circuit and constant current control method for being used for LLC resonant converter,
Wherein, the thermal compensation signal that constant-current control circuit obtains includes resonance current signal and the first transformer magnetizing current signal, thus
Constant-current control accuracy can be improved.
According to an aspect of the present invention, a kind of constant-current control circuit for LLC resonant converter is provided, the LLC is humorous
The converter that shakes includes the first transformer, the first bipolar junction transistor and the second bipolar junction transistor, first bipolar transistor
Pipe and second bipolar junction transistor are worked using self-oscillation mode, so that resonance current and exciting current flow through described the
The primary side winding of one transformer, the constant-current control circuit include: switch element, for being shorted first bipolar junction transistor
With the driving current of at least one second bipolar junction transistor；And drive module, including output current calculation module, it uses
In the difference for calculating resonance current signal and the first transformer magnetizing current signal absolute value average value as thermal compensation signal,
And the on state of the switch element according to compensation signal control is to realize the control of resonance frequency, to realize constant current control
System.
Preferably, the LLC resonant converter further includes the second transformer, and second transformer has load winding,
And the first driving winding and the second driving winding, the load winding of second transformer coupled with the load winding connects
It connects on resonant tank to obtain resonance current, the different name of the Same Name of Ends of the first driving winding and the second driving winding
End is respectively connected to the base stage of first bipolar junction transistor and second bipolar junction transistor, to provide according to
The respective drive electric current that the induced current of resonance current generates.
Preferably, the switch element is when being shorted the driving current by the first driving Same Name of Ends of winding and different
Name end is connected to each other.
Preferably, second transformer further includes control winding, and the switch element is when being shorted the driving current
The Same Name of Ends of the control winding and different name end are connected to each other.
Preferably, the switch element includes: the first transistor and second transistor, and the first transistor is connected to institute
It states between the different name end and ground terminal of the first driving winding, the second transistor is connected to the of the same name of the first driving winding
Between end and ground terminal, the ground terminal is connected in first bipolar junction transistor and second bipolar junction transistor
Intermediate node.
Preferably, further includes: the first operational amplifier and second operational amplifier are connected respectively to the first transistor
It is connected with the control terminal of second transistor, wherein the drive module provides open signal, first operational amplifier and institute
It states second operational amplifier and cut-off signals is provided, the switch control signal of the first transistor and second transistor is described opens
The superposed signal of messenger and the cut-off signals.
Preferably, first operational amplifier and the respective non-inverting input terminal of the second operational amplifier receive negative electricity
Position reference voltage, inverting input terminal is each connected to output end, to realize the Same Name of Ends of the corresponding windings of second transformer
With the negative voltage clamper at different name end.
Preferably, the switch element includes: the first transistor and second transistor, and differential concatenation is connected to described second
Between the Same Name of Ends and ground terminal of the corresponding windings of transformer, the different name end of the corresponding windings of second transformer and described connect
Ground terminal is connected to the intermediate node of first bipolar junction transistor and second bipolar junction transistor.
Preferably, the drive module is connect with the control terminal of the first transistor and second transistor to provide switch
Control signal.
Preferably, first transformer includes primary side winding and vice-side winding, and the primary side winding is as resonant tank
A part, the vice-side winding couples with the primary side winding to provide resonance output voltage, wherein the output galvanometer
It calculates module and institute is obtained according to the current sampling signal of the resonance current and the voltage feedback signal of the resonance output voltage
State thermal compensation signal.
Preferably, the output current calculation module includes: the third crystalline substance that third operational amplifier and output end are connected
Body pipe, for generating the first electric current；The 4th transistor that four-operational amplifier and output end are connected, for generating the second electricity
Stream；Multiple current mirrors, for by first electric current and second current subtraction to generate equivalent charging current；And electricity
Hold, for being integrated to equivalent charging current to generate the thermal compensation signal, wherein the third operational amplifier and described
The non-inverting input terminal of four-operational amplifier receives the first reference voltage and the second reference voltage, the third operation amplifier respectively
The inverting input terminal of one of device and the four-operational amplifier receives the current sampling signal, another inverting input terminal
Ground connection.
Preferably, the output current calculation module further include: first switch, for by the third operational amplifier
Inverting input terminal selectively grounded or receives the current sampling signal；Second switch is used for the 4th operation amplifier
The inverting input terminal of device selectively grounded or receives the current sampling signal；Comparator, by the voltage feedback signal with
Third reference voltage compares, to generate the control signal of the first switch and the second switch.
Preferably, second reference voltage is greater than first reference voltage.
Preferably, the drive module further include: oscillator generates clock letter according to ramp signal and the thermal compensation signal
Number；And logic module, open signal or switch control signal are generated according to the clock signal.
According to another aspect of the present invention, a kind of constant current control method for LLC resonant converter, the LLC are provided
Controlled resonant converter includes the first transformer, the first bipolar junction transistor and the second bipolar junction transistor, the first ambipolar crystalline substance
Body pipe and second bipolar junction transistor are worked using self-oscillation mode so that resonance current and exciting current flow through it is described
The primary side winding of first transformer, the constant current control method include: to calculate resonance current signal and the first static exciter electricity
The average value of the absolute value of the difference of signal is flowed, thermal compensation signal is obtained；According to the conducting of the compensation signal control switch element
State is to realize the control of resonance frequency, to realize current constant control, wherein be shorted described first pair in switching elements conductive
The driving current of at least one bipolar transistor and second bipolar junction transistor.
Preferably, the step of obtaining thermal compensation signal includes: to refer to the current sampling signal of resonance current signal and first
Voltage compares, to generate the first electric current；Second electric current is generated using the second reference voltage；By first electric current and described
Two current subtractions are to generate equivalent charging current；And equivalent charging current is integrated to generate the thermal compensation signal,
In, second reference voltage is greater than first reference voltage.
It preferably, further include switching the path of the current sampling signal, according to the resonance output voltage signal to obtain
Obtain the average value of the absolute value of the difference of the resonance current signal and the first transformer magnetizing current signal.
Preferably, the resonance output voltage signal is compared with third reference voltage to obtain the circuit sampling letter
Number path switching signal.
It preferably, further include being carried out to the control terminal of first bipolar junction transistor and second bipolar junction transistor
Negative voltage clamper.
Constant-current control circuit according to an embodiment of the present invention is shorted the first bipolar junction transistor and second using switch element
The driving current of bipolar junction transistor, so that the switch of first bipolar junction transistor and second bipolar junction transistor is all
Phase follows switch control signal, to realize the control of resonance frequency.The thermal compensation signal characterization that the constant-current control circuit obtains is humorous
The average value of the absolute value for electric current and the first transformer magnetizing current difference of shaking, according to the negative feedback control switch control of average value
The frequency of signal, so as in the output electric current constant current control of the secondary side of the first transformer of the primary side side of the first transformer realization
System.In the complicated circuit for constituting charge pump PFC module and LLC resonant converter combined application, the constant-current control circuit is still
Constant-current control accuracy so can be improved.
In a preferred embodiment, the control circuit in constant-current control circuit can directly control the first bipolar junction transistor
It is ambipolar to be indirectly controlled second using the coupling between the first driving winding and the second driving winding for the driving current of base stage
The driving current of transistor base.The constant-current control circuit is not necessarily to provide additional control circuit for the second bipolar junction transistor,
So as to be further simplified the circuit structure of control circuit and reduce circuit cost.
In a preferred embodiment, control circuit includes being connected between the different name end and Same Name of Ends of the first driving winding
The first transistor and second transistor control the driving electricity of the first bipolar junction transistor base stage for being shorted the first driving winding
Stream.Ground terminal (floating ground) of the intermediate node of first bipolar junction transistor and the second bipolar junction transistor as control circuit.The
One transistor and second transistor are as switch element, for being shorted the first driving winding, because without practical ground connection.The control
Control electric current needed for circuit generates control winding without power supply circuit, thus the power consumption of circuit can be reduced and reduce electricity
Road cost.
Detailed description of the invention
By referring to the drawings to the description of the embodiment of the present invention, above-mentioned and other purposes of the invention, feature and
Advantage will be apparent from, in the accompanying drawings:
Fig. 1 shows the schematic circuit of power supply device according to prior art.
Fig. 2 shows the schematic circuits of LED drive circuit according to a first embodiment of the present invention.
Fig. 3 shows the schematic circuit of LED drive circuit according to a second embodiment of the present invention.
Fig. 4 shows the schematic circuit of control circuit in LED drive circuit shown in Fig. 3.
Fig. 5 shows the working waveform figure of LED drive circuit shown in Fig. 3.
Fig. 6 a to 6c shows the equivalent circuit diagram of LED drive circuit shown in Fig. 3 in the first stage.
Fig. 7 a to 7b shows LED drive circuit shown in Fig. 3 in the equivalent circuit diagram of second stage.
Fig. 8 a to 8c shows LED drive circuit shown in Fig. 3 in the equivalent circuit diagram of phase III.
Fig. 9 a to 9b shows LED drive circuit shown in Fig. 3 in the equivalent circuit diagram of fourth stage.
Figure 10 shows the schematic circuit of control circuit in LED drive circuit according to a third embodiment of the present invention.
Figure 11 shows the working waveform figure of control circuit shown in Figure 10.
Figure 12 shows the detailed circuit block diagram of control circuit shown in Fig. 4.
Figure 13 shows the schematic circuit that current calculation module is exported in control circuit shown in Figure 12.
Figure 14 shows the schematic illustration that control circuit shown in Figure 12 carries out current regulation.
Specific embodiment
The various embodiments that the present invention will be described in more detail that hereinafter reference will be made to the drawings.In various figures, identical element
It is indicated using same or similar appended drawing reference.For the sake of clarity, the various pieces in attached drawing are not necessarily to scale.
Fig. 1 shows the schematic circuit of power supply device according to prior art.Power supply device 100 include rectifier bridge DB,
Filter condenser Ce, charge pump PFC module 110, controlled resonant converter 120.Rectifier bridge DB is for converting AC-input voltage AC
At rectified input voltage.Charge pump PFC module 110 is folded using the resonance output voltage and resonance current obtained from resonant tank
The input terminal of LLC resonant converter 120 is added in realize PFC.Filter condenser Ce converts rectified input voltage
At smooth DC input voitage.DC input voitage is converted into resonance output voltage by controlled resonant converter 120, thus to load
LD power supply.
Charge pump PFC module 110 includes diode DX1 and DX2, diode Di1 and Di2, boost capacitor Ci1 and Ci2.
Current source electric charge pump module includes boost capacitor Ci2 and diode Di2, utilizes resonant inductor Lr and resonant capacitor Cr group
At resonant tank generate resonance current as current source.Voltage source electric charge pump module includes boost capacitor Ci1 and two poles
Pipe Di1, using the end voltage of resonant capacitor Cr as voltage source.
Controlled resonant converter 120 includes control circuit 121, switch element M1 and M2, coupling capacitor Cc, resonant inductor Lr
With resonant capacitor Cr.The on state of control circuit 121 control switch element M1 and M2 generate square-wave voltage.Square wave electricity
Pressure input resonant tank, to generate resonance.The end voltage of resonant capacitor Cr powers to the load.
In the power supply device 100, electric current source charge pump utilizes the conducting of switch element and disconnects the high-frequency current generated
Ring obtains the electric energy of AC-input voltage, the high frequency node electricity that voltage source charge pump is generated using the conducting and disconnection of switch element
Pressure obtains the electric energy of AC-input voltage, to promote the voltage and current of DC input voitage.Switch element M1 and M2 pass through
Switched conductive and off-state are to generate high frequency voltage and electric current.It is formed due to resonant inductor Lr and resonant capacitor Cr
Load of the resonant tank as switch element M1 and M2, therefore, high frequency output electric current are the resonance current under resonance frequency.
In the LED drive circuit, switch element M1 and M2 used in controlled resonant converter are respectively metal oxide half
Conductor field effect transistor (MOSFET) is used as switching tube.Although MOSFET has outstanding switch performance, complexity is needed
Control circuit provides control signal for switching tube, therefore, leads to mentioning for LED drive circuit cost as switching tube using MOSFET
It is high.
Fig. 2 shows the schematic circuits of LED drive circuit according to a first embodiment of the present invention.Power supply device 200 wraps
Include rectifier bridge DB, filter condenser Cht, charge pump PFC module 210, LLC resonant converter 220.
Rectifier bridge DB is used to AC-input voltage AC being converted into rectified input voltage.
Charge pump PFC module 210 includes diode D1 and boost capacitor Cboost.Diode D1 is connected to rectifier bridge DB
Positive output end and LLC resonant converter 220 positive input terminal between, to form rectifier bridge DB to LLC resonant converter 220
Unilateal conduction path.The cathode of diode D1 is connected to the first end of the resonant capacitor Cr in LLC resonant converter 220.
Boost capacitor Cboost is connected between the positive output end and negative output terminal of rectifier bridge DB.Charge pump PFC module 210 using from
The resonance current that resonant tank obtains extracts electric current from rectified input voltage to realize PFC, and gives filtered electrical
Container Cht provides electric current, to realize the function of boosting.
Filter condenser Cht is connected between the output end of charge pump PFC module 210 and the negative output terminal of rectifier bridge DB.
Rectified input voltage is converted into smooth DC input voitage by filter condenser Cht.
LLC resonant converter 220 includes the first transformer T1, the second transformer T2, bipolar junction transistor Q1 and Q2, two poles
Pipe D2 and D3, capacitor Cmid, resonant capacitor Cr and resonant inductor Lr.Diode D2 and D3 respectively with bipolar junction transistor
Q1 and Q2 are connected in inverse parallel, and capacitor Cmid is connected in parallel with bipolar junction transistor Q2.
In the primary side of the first transformer T1, primary side winding Lp, resonant capacitor Cr and the resonant inductance of the first transformer T1
Device Lr forms resonant tank.Between the positive input terminal and negative input end of LLC resonant converter 220, bipolar junction transistor Q1 and
Q2 is connected in series, and the intermediate node of the two is connected to resonant tank.In resonant tank, sampling resistor Rs and primary side winding Lp are gone here and there
Connection connection, it is hereby achieved that the sampled signal for characterizing the inductive current for flowing through primary side winding Lp.Second transformer T2 packet
Include four windings around same iron core, i.e. load winding W1, driving winding W2 and W3, control winding W4.In resonant tank,
Load winding W1 and primary side winding Lp is connected in series.Meanwhile drive winding W2 and W3 respectively with bipolar junction transistor Q1 and Q2
Base stage coupling, but it is contrary.That is, the Same Name of Ends of driving winding W2 is connected to the base stage of bipolar junction transistor Q1, driving around
The different name end of group W3 is connected to the base stage of bipolar junction transistor Q2.These windings are for providing necessary electric current to drive ambipolar crystalline substance
The base stage of body pipe Q1 and Q2, to realize that self-oscillation drives (SOC, Self-Oscillating Converter).From exciting
It swings under the control of driving signal, DC input voitage is converted into square wave by bipolar junction transistor Q1 and Q2 alternate conduction and shutdown
Voltage.The square-wave voltage inputs resonant tank, to generate the resonance current under resonance frequency.Therefore, pass through resonant tank, electric energy
The secondary side of the first transformer T1 is transferred to from the primary side of the first transformer T1.
On the secondary side of the first transformer T1, diode D4 and D5 form rectification circuit.The both ends of vice-side winding are separately connected
The anode of diode D4 and D5, the centre tap ground connection of vice-side winding.Output capacitance C1 is connected to the cathode of diode D4 and D5
Between ground, process resonance output voltage is provided at its both ends.
LLC resonant converter 220 further includes constant-current control circuit 221.The constant-current control circuit 221 is from controlled resonant converter
The current sampling signal CS that resonance current is obtained on 220 sampling resistor Rs, from the first transformer T1's of controlled resonant converter 220
The voltage feedback signal FB of auxiliary winding Lf acquisition resonance output voltage.The constant-current control circuit 221 includes being respectively connected to the
The different name end of the control winding W4 of two transformer T2 and the driving end DR1 and DR2 of Same Name of Ends.The constant-current control circuit 221 passes through
The connection relationship of control driving end DR1 and DR2, so that resonance frequency is controlled, to control resonance current.
During operation, DC input voitage is converted into resonance output voltage by LLC resonant converter 220, thus to LED
Load supplying.The switch commutation of bipolar junction transistor Q1 and Q2 are spontaneous in LLC resonant converter 220, are intrinsic
SOC frequency of oscillation.However, when LLC resonant converter 220 works, it is also necessary to adjust its switching frequency, the frequency is general
Higher than intrinsic SOC frequency of oscillation.
LED drive circuit is real using the concatenated schemes of charge pump PFC module and LLC resonant converter according to this embodiment
Existing AC-DC voltage transformation, powers to LED load, it is hereby achieved that very high power factor (PF) and very low total harmonic wave lose
Very (THD).Using bipolar junction transistor as switching tube in LLC resonant converter, using self-oscillation control switch pipe
Conducting and off-state, and control the short circuit of control winding and discharge short-circuit condition in the suitable time, carry out control switch
Pipe alternate conduction, thus to control resonance frequency, thus realize current constant control and simplified control circuit and reduce circuit at
This.
Fig. 3 shows the schematic circuit of LED drive circuit according to a second embodiment of the present invention.Power supply device 300 wraps
Include rectifier bridge DB, filter condenser Cht, charge pump PFC module 310, LLC resonant converter 320.
Rectifier bridge DB is used to AC-input voltage AC being converted into rectified input voltage.
Charge pump PFC module 310 includes diode D1 and boost capacitor Cboost.Diode D1 is connected to rectifier bridge DB
Positive output end and LLC resonant converter 320 positive input terminal between, to form rectifier bridge DB to LLC resonant converter 320
Unilateal conduction path.The anode of diode D1 is connected to the first end of the resonant capacitor Cr in LLC resonant converter 320.
Boost capacitor Cboost is connected between the positive output end and negative output terminal of rectifier bridge DB.Charge pump PFC module 310 using from
The resonance current that resonant tank obtains extracts electric current from rectified input voltage to realize PFC, and gives filtered electrical
Container Cht provides electric current, to realize the function of boosting.
Filter condenser Cht is connected between the output end of charge pump PFC module 310 and the negative output terminal of rectifier bridge DB.
Rectified input voltage is converted into smooth DC input voitage by filter condenser Cht.
LLC resonant converter 320 includes the first transformer T1, the second transformer T2, bipolar junction transistor Q1 and Q2, two poles
Pipe D2 and D3, capacitor Cmid, resonant capacitor Cr and resonant inductor Lr.Diode D2 and D3 respectively with bipolar junction transistor
Q1 and Q2 are connected in inverse parallel, and capacitor Cmid is connected in parallel with bipolar junction transistor Q2.
In the primary side of the first transformer T1, primary side winding Lp, resonant capacitor Cr and the resonant inductance of the first transformer T1
Device Lr forms resonant tank.Between the positive input terminal and negative input end of LLC resonant converter 320, bipolar junction transistor Q1 and
Q2 is connected in series, and the intermediate node of the two is connected to resonant tank.In resonant tank, sampling resistor Rs and primary side winding Lp are gone here and there
Connection connection, it is hereby achieved that the sampled signal for characterizing the inductive current for flowing through primary side winding Lp.Second transformer T2 packet
Include three windings around same iron core, i.e. load winding W1, driving winding W2 and W3.In resonant tank, load winding W1
It is connected in series with primary side winding Lp.Meanwhile winding W2 and W3 being driven to couple respectively with the base stage of bipolar junction transistor Q1 and Q2, but
It is contrary.That is, the Same Name of Ends of driving winding W2 is connected to the base stage of bipolar junction transistor Q1, the different name end of winding W3 is driven
It is connected to the base stage of bipolar junction transistor Q2.These windings are for providing necessary electric current to drive bipolar junction transistor Q1 and Q2
Base stage, to realize that self-oscillation drives (SOC, Self-Oscillating Converter).In self-oscillation driving signal
Under control, DC input voitage is converted into square-wave voltage by bipolar junction transistor Q1 and Q2 alternate conduction and shutdown.Square wave electricity
Pressure input resonant tank, to generate the resonance current under resonance frequency.Therefore, by resonant tank, electric energy is from the first transformer
The primary side of T1 is transferred to the secondary side of the first transformer T1.
On the secondary side of the first transformer T1, diode D4 and D5 form rectification circuit.The both ends of vice-side winding are separately connected
The anode of diode D4 and D5, the centre tap ground connection of vice-side winding.Output capacitance C1 is connected to the cathode of diode D4 and D5
Between ground, process resonance output voltage is provided at its both ends.
LLC resonant converter 320 further includes constant-current control circuit 321.The constant-current control circuit 221 is from controlled resonant converter
The current sampling signal CS that resonance current is obtained on 220 sampling resistor Rs, from the first transformer T1's of controlled resonant converter 220
The voltage feedback signal FB of auxiliary winding Lf acquisition resonance output voltage.The constant-current control circuit 321 includes being respectively connected to the
The different name end of the driving winding W2 of two transformer T2 and the driving end DR1 and DR2 of Same Name of Ends, and it is connected to bipolar junction transistor
The ground terminal GND of the intermediate node of Q1 and Q2.The constant-current control circuit 321 passes through control driving end DR1 and DR2 and ground terminal
The connection relationship of GND, so that resonance frequency is controlled, to control resonance current.
During operation, DC input voitage is converted into resonance output voltage by LLC resonant converter 320, thus to LED
Load supplying.The switch commutation of bipolar junction transistor Q1 and Q2 are spontaneous in LLC resonant converter 320, are intrinsic
SOC frequency of oscillation.However, when LLC resonant converter 320 works, it is also necessary to adjust its switching frequency, the frequency is general
Higher than intrinsic SOC frequency of oscillation.
LED drive circuit is real using the concatenated schemes of charge pump PFC module and LLC resonant converter according to this embodiment
Existing AC-DC voltage transformation, powers to LED load, it is hereby achieved that very high power factor (PF) and very low total harmonic wave lose
Very (THD).Using bipolar junction transistor as switching tube in LLC resonant converter, using self-oscillation control switch pipe
Conducting and off-state, and control the short circuit of at least one driving winding and discharge short-circuit condition in the suitable time, come
Control switch pipe alternate conduction, thus to control resonance frequency, to realize current constant control and simplified control circuit and reduction
Circuit cost.
Fig. 4 shows the schematic circuit of control circuit in LED drive circuit shown in Fig. 3.
Constant-current control circuit 321 includes transistor M1 and M2, operational amplifier U1 and U2, drive module 3211.In the reality
It applies in example, transistor M1 and M2 are, for example, MOSFET.Further, the first end and second end of transistor M1 is connected to drive
Between moved end DR1 and ground terminal GND, the first end and second end of transistor M2 is connected to driving end DR2 and ground terminal GND
Between.
Drive module 3211 obtains the current sampling signal of resonance current from the sampling resistor Rs of controlled resonant converter 320
CS obtains the voltage feedback signal FB of resonance output voltage from the auxiliary winding Lf of the first transformer T1 of controlled resonant converter 320,
And thermal compensation signal Vcomp is generated according to current sampling signal CS and voltage feedback signal FB.Drive module 3211 and transistor
M1 is connected with the control terminal of M2, for providing open signal VG1 and VG2, operational amplifier U1 to transistor M1 and M2 respectively
Cut-off signals are provided to transistor M1 and M2 with U2, therefore, the switch control signal of transistor M1 and M2 are open signal and pass
The superposed signal of break signal.The non-inverting input terminal of operational amplifier U1 receives negative potential reference voltage-Vref, preferably -0.1V,
Inverting input terminal is connected with output end, and further, the output end of operational amplifier U1 is connected with the control terminal of transistor M1
It connects, operational amplifier U1 is other than it can control M1 shutdown, and when negative pressure occurs in the end DR1, operational amplifier U1 is controlled at M1
In magnifying state, it is ensured that the end DR1 voltage is not less than -0.1V.The non-inverting input terminal of operational amplifier U2 receives negative potential with reference to electricity
Pressure, preferably -0.1V, inverting input terminal are connected with output end, further, the output end and transistor of operational amplifier U2
The control terminal of M2 is connected, and operational amplifier U2 is other than it can control M2 shutdown, and when negative pressure occurs in the end DR2, operation is put
Big device U2 control M2 is in magnifying state, it is ensured that the end DR2 voltage is not less than -0.1V.
Fig. 5 shows the working waveform figure of LED drive circuit shown in Fig. 3.It is shown in figure the humorous of the acquisition of drive module 3211
Shake the exciting current CT1 of current sampling signal CS, voltage feedback signal FB, clock signal clk and the first transformer T1, the second change
The exciting current CT2 of depressor T2 changes with time relationship.
The exciting current CT2 of resonance current sampled signal CS and the second transformer T2 intersect at A, B, C point.Clock signal
CLK has a level of high and low (1,0) two states, and resonance current sampled signal CS also has the electricity of positive and negative (>0,<0) two states
Flat, combination of two, there are four types of different states altogether, to generate the different circuit stages.
It is open-minded that drive module 3211 in the low level state of clock signal clk, constant-current control circuit 321 generates first
Signal VG1, so that transistor M1 is connected, transistor M2 then has operational amplifier U2 control, and can be there are two state: one be off shape
State；Second is that negative voltage clamping state.Drive module in the high level state of clock signal clk, constant-current control circuit 321
3211 generate the second open signal VG2, so that transistor M2 is connected, transistor M1 then has operational amplifier U1 control, have two
A state: first is that off state；Second is that negative voltage clamping state.
It is converted into just in the current sampling signal CS of the low level time section of clock signal clk, resonance current from negative current
Electric current.Negative electricity is converted into from positive current in the current sampling signal CS of the high level time section of clock signal clk, resonance current
Stream.
Therefore, the first stage of LED drive circuit corresponds to the time period t 0 in figure to t1, and second stage corresponds in figure
Time period t 1 to t2, the phase III corresponds to the time period t 2 in figure to t3, and fourth stage corresponds to the time period t 3 in figure
To t4.
Fig. 6 a to 6c shows the equivalent circuit diagram of LED drive circuit shown in Fig. 3 in the first stage.As shown, clock
Signal CLK is low level, is the first stage of circuit when resonance current is negative current.In the first phase, constant-current control circuit
Transistor M1 conducting in 321, transistor M2 negative voltage clamper.The current path of LED drive circuit 300 is sampled because of resonance current
The change of the charged state of the variation and capacitor Cmid of the difference of the exciting current CT2 of signal CS and the second transformer T2
And it changes.
Rectifier bridge DB includes four diode D11 to D14 for forming bridge, in the positive output end and negative output of rectifier bridge DB
Rectified input voltage is provided between end.
In moment t0, bipolar junction transistor Q1 and Q2 are off state.AC-input voltage is via resonant tank to electricity
Container Cmid charging.During the charging of capacitor Cmid, the end voltage Vmid of capacitor Cmid is gradually risen.Resonance current is anti-
To the load winding W1 of the primary side winding Lp for flowing through the first transformer T1 and the second transformer T2, that is, in the inside of corresponding windings
Same Name of Ends is flowed to from different name end, can judge to drive according to the difference of the exciting current CT2 of resonance current CS and the second transformer T2
The current internal of dynamic winding W2 and W3 are that different name end is flowed to from Same Name of Ends, since the voltage difference of the two ends of driving winding W2 only has
0.1V, therefore may determine that inside driving winding W3 be not no electric current flowing.
As shown in Figure 6 a, the resonance current path in LED drive circuit 300 are as follows: from the positive output end of rectifier bridge DB, via
Resonant capacitor Cr, resonant inductor Lr, the primary side winding Lp of the first transformer T1, the second transformer T2 load winding W1,
Sampling resistor Rs, capacitor Cmid return to the negative output terminal of rectifier bridge DB.Further, since the crystal in constant-current control circuit 321
Pipe M2 negative voltage clamper, effect are similar to the electricity between the driving end DR2 and ground terminal GND for being connected to constant-current control circuit 321
Potential source, therefore, the driving current path of bipolar junction transistor Q1 are as follows: from the driving end DR2 of constant-current control circuit 321, via
The driving end DR1 of driving the winding W2 and constant-current control circuit 321 of two transformer T2, return to the ground connection of constant-current control circuit 321
GND is held, current loop is formed.The driving current path of bipolar junction transistor Q2 disconnects.
Then, when voltage Vmid is greater than the end voltage of filter condenser Cht, resonance current path changes.At this point,
The base-collector junction afterflow of bipolar junction transistor Q1, so that work is connected in reverse phase.Resonance current is no longer to capacitor Cmid
Charging, but charge via bipolar junction transistor Q1 to filter condenser Cht.
As shown in Figure 6 b, the resonance current path in LED drive circuit 300 are as follows: from the positive output end of rectifier bridge DB, via
Resonant capacitor Cr, resonant inductor Lr, the primary side winding Lp of the first transformer T1, the second transformer T2 load winding W1,
Sampling resistor Rs, bipolar junction transistor Q1, filter condenser Cht return to the negative output terminal of rectifier bridge DB.In addition, ambipolar crystalline substance
The driving current path of body pipe Q1 remains unchanged, and the driving current path of bipolar junction transistor Q2 disconnects.
Then, in Fig. 5 after A point, occur in the exciting current difference of resonance current CS and the second transformer T2 by bearing
Become timing, the current direction of driving the winding W2 and W3 of the second transformer T2 will also change, and flow direction is same from internal different name end
Name end, at this point, transistor M2's constant-current control circuit 321 inside becomes off state by negative voltage clamping state, resonance is electric
The one part of current of stream reversely flows through the driving winding W2 of the second transformer T2, i.e., flows in the inside of corresponding windings from different name end
Same Name of Ends, and the base-collector junction of bipolar junction transistor Q1 is flowed through, so that bipolar junction transistor Q1 is completely reversed conducting.It is humorous
Another part of vibration electric current charges to filter condenser Cht via bipolar junction transistor Q1.
As fig. 6 c, the resonance current path in LED drive circuit 300 are as follows: from the positive output end of rectifier bridge DB, via
Resonant capacitor Cr, resonant inductor Lr, the primary side winding Lp of the first transformer T1, the second transformer T2 load winding W1,
Sampling resistor Rs, bipolar junction transistor Q1, filter condenser Cht return to the negative output terminal of rectifier bridge DB.In addition, ambipolar crystalline substance
The driving current path of body pipe Q1 via the second transformer T2 driving winding W2 and bipolar junction transistor Q1 base collector
The driving current path of knot, bipolar junction transistor Q2 disconnects.
Terminate in the negative current stage of moment t1, resonance current, the first stage of LED drive circuit 300 terminates.
Fig. 7 a to 7b shows LED drive circuit shown in Fig. 3 in the equivalent circuit diagram of second stage.As shown, clock
Signal CLK is low level, is the second stage of circuit when resonance current is positive current.In second stage, constant-current control circuit
Transistor M1 conducting, transistor M2 shutdown in 321.The current path of LED drive circuit 300 is because of resonance current sampled signal CS
With the variation of the difference of the exciting current CT2 of the second transformer T2.
In moment t1, resonance current CS is positive, and the driving winding W2 of the second transformer T2 obtains reversed driving current, makes
Obtaining bipolar junction transistor Q1 is on state.Bipolar junction transistor Q2 is maintained off state.Resonance current is via diode D1
It charges to boost capacitor Cboost.During the charging of capacitor Cboost, the end voltage of boost capacitor Cboost
Vboost gradually rises.Resonance current forward direction flow through the load of the primary side winding Lp and the second transformer T2 of the first transformer T1 around
Group W1, that is, different name end is flowed to from Same Name of Ends in the inside of corresponding windings, it can be according to resonance current CS's and the second transformer T2
The current internal of difference judgement driving the winding W2 and W3 of exciting current CT2 are to flow to Same Name of Ends from different name end.
As shown in Figure 7a, the resonance current path in LED drive circuit 300 are as follows: from the first end of resonant inductor Lr, warp
By resonant capacitor Cr, boost capacitor Cboost, filter condenser Cht, bipolar junction transistor Q1, sampling resistor Rs, second
The primary side winding Lp of the load winding W1 of transformer T2, the first transformer T1 return to the second end of resonant inductor Lr.In addition,
Since the driving winding W2 of the second transformer T2 obtains reversed driving current, the driving current of bipolar junction transistor Q1
Path are as follows: returned from the Same Name of Ends of the driving winding W2 of the second transformer T2 via the base emitter junction of bipolar junction transistor Q1
Go back to the different name end of the driving winding W2 of the second transformer T2.The driving current path of bipolar junction transistor Q2 disconnects.
Then, when the end voltage Vboost of boost capacitor Cboost is greater than the end voltage Vht of filter condenser Cht,
Resonance current path changes.At this point, diode D1 is connected.Resonance current no longer charges to boost capacitor Cboost, and
It is the collector that bipolar junction transistor Q1 is flowed to via diode D1.
As shown in Figure 7b, the resonance current path in LED drive circuit 300 are as follows: from the first end of resonant inductor Lr, warp
By resonant capacitor Cr, diode D1, bipolar junction transistor Q1, sampling resistor Rs, the load winding W1 of the second transformer T2,
The primary side winding Lp of one transformer T1 returns to the second end of resonant inductor Lr.Further, since the driving of the second transformer T2 around
Group W2 obtains reversed driving current, and therefore, the driving current path of bipolar junction transistor Q1 remains unchanged.Bipolar junction transistor
The driving current path of Q2 disconnects.
Terminate in the low level stage of moment t2, clock signal clk, the second stage of LED drive circuit 300 terminates.
Fig. 8 a to 8c shows LED drive circuit shown in Fig. 3 in the equivalent circuit diagram of phase III.As shown, clock
Signal CLK is high level, and resonance current CS is the phase III of circuit when being positive current.In the phase III, current constant control electricity
Transistor M1 negative voltage clamper, transistor M2 conducting in road 321.The conducting of transistor M2 so that bipolar junction transistor Q1 base
Pole emitter is shorted, and therefore, bipolar junction transistor Q1 is in off state always.The current path of LED drive circuit 300 is because of electricity
Flow the variation of the difference of the exciting current CT2 of sampled signal CS and the second transformer T2 and the charged state of capacitor Cmid
Change and change.
In moment t2, clock signal clk is turned into high level from low level, and bipolar junction transistor Q1 and Q2 are cut-off shape
State.Capacitor Cmid discharges via resonant tank.During the electric discharge of capacitor Cmid, the end voltage Vmid of capacitor Cmid is gradually
It reduces.Resonance current forward direction flows through the load winding W1 of the primary side winding Lp and the second transformer T2 of the first transformer T1, that is,
The inside of corresponding windings flows to different name end from Same Name of Ends, can be according to the exciting current of resonance current CS and the second transformer T2
The current internal of difference judgement driving the winding W2 and W3 of CT2 are to flow to Same Name of Ends from different name end, due to the voltage difference of the two ends of W2
Only 0.1V, therefore may determine that inside W3 it is not no electric current flowing.
As shown in Figure 8 a, the resonance current path in LED drive circuit 300 are as follows: from the first end of resonant inductor Lr, warp
By resonant capacitor Cr, diode D1, filter condenser Cht, capacitor Cmid, sampling resistor Rs, the second transformer T2 it is negative
The primary side winding Lp of winding W1, the first transformer T1 are carried, the second end of resonant inductor Lr is returned.Further, since current constant control
Transistor M1 negative voltage clamper in circuit 321, effect are similar to connecing for the driving end DR1 for being connected to constant-current control circuit 321
Voltage source between ground terminal GND, therefore, the driving current path of bipolar junction transistor Q1 are as follows: from the drive of constant-current control circuit 321
Moved end DR1 returns to current constant control via the driving end DR2 of driving the winding W2 and constant-current control circuit 321 of the second transformer T2
The ground terminal GND of circuit 321 forms current loop.The driving current path of bipolar junction transistor Q2 disconnects.
Then, when being less than the voltage of ground terminal GND of constant-current control circuit 321 in voltage Vmid, bipolar junction transistor Q2
Driving current path change.At this point, the base-collector junction afterflow of bipolar junction transistor Q2, so that work is led in reverse phase
It is logical.Capacitor Cmid no longer discharges via resonant tank.
As shown in Figure 8 b, the one part of current forward direction of resonance current flows through the driving winding W3 of the second transformer T2, that is, exists
The inside of corresponding windings flows to different name end from Same Name of Ends, and flows through the base-collector junction of bipolar junction transistor Q2, so that double
Bipolar transistor Q2 is completely reversed conducting.Another part of resonance current flows to resonant tank via bipolar junction transistor Q2.This
Outside, the driving current path of bipolar junction transistor Q1 remains unchanged.
Then, in Fig. 5 after B point, occur in the exciting current difference of resonance current and the second transformer T2 by just becoming
When negative, the current direction of driving the winding W2 and W3 of the second transformer T2 will also change, at this point, flowing to from internal Same Name of Ends
Different name end, transistor M1's inside constant-current control circuit 321 becomes off state by negative voltage clamping state.
As shown in Figure 8 c, the resonance current path in LED drive circuit 300 are as follows: from the first end of resonant inductor Lr, warp
By the load of resonant capacitor Cr, diode D1, filter condenser Cht, triode Q2, sampling resistor Rs, the second transformer T2
The primary side winding Lp of winding W1, the first transformer T1 return to the second end of resonant inductor Lr.In addition, bipolar junction transistor Q1
Driving current path disconnect, the driving current path of bipolar junction transistor Q2 via bipolar junction transistor Q2 base collector
Knot.
Terminate in the positive current stage of moment t3, resonance current, the phase III of LED drive circuit 300 terminates.
Fig. 9 a to 9b shows LED drive circuit shown in Fig. 3 in the equivalent circuit diagram of fourth stage.As shown, clock
Signal CLK is high level, is the fourth stage of circuit when resonance current is negative current.In fourth stage, constant-current control circuit
Transistor M1 shutdown in 321, transistor M2 conducting.The conducting of transistor M2 so that bipolar junction transistor Q1 Base-Emitter
It is shorted, therefore, bipolar junction transistor Q1 is in off state always.The current path of LED drive circuit 300 is believed because of current sample
The variation of the difference of the exciting current CT2 of number CS and the second transformer.
In moment t3, resonance current is reversed, and the driving winding W3 of the second transformer T2 obtains positive driving current, so that
Bipolar junction transistor Q2 is on state.Bipolar junction transistor Q1 is maintained off state.Boost capacitor Cboost is via humorous
The electric discharge of vibration circuit, the end voltage Vboost of boost capacitor Cboost are gradually decreased.Resonance current reversely flows through the first transformer
The load winding W1 of the primary side winding Lp of T1 and the second transformer T2, that is, flow in the inside of corresponding windings from different name end of the same name
End can judge the electricity of driving winding W2 and W3 according to the difference of the exciting current CT2 of resonance current CS and the second transformer T2
It is that different name end is flowed to from Same Name of Ends that stream is internal.
As illustrated in fig. 9, the resonance current path in LED drive circuit 300 are as follows: from the first of boost capacitor Cboost
End, via the load of resonant capacitor Cr, resonant inductor Lr, the primary side winding Lp of the first transformer T1, the second transformer T2
Winding W1, sampling resistor Rs, bipolar junction transistor Q2 return to the second end of boost capacitor Cboost.Further, since second becomes
The driving winding W3 of depressor T2 obtains positive driving current, and the driving current path of bipolar junction transistor Q2 is via ambipolar crystalline substance
The driving current path of the base emitter junction of body pipe Q2, bipolar junction transistor Q2 forward conduction, bipolar junction transistor Q1 disconnects.
Then, when the end voltage Vboost of boost capacitor Cboost is less than AC-input voltage, resonance current path
It changes.AC-input voltage is powered to resonant tank.
As shown in figure 9b, the resonance current path in LED drive circuit 300 are as follows: from the positive output end of rectifier bridge DB, via
Resonant capacitor Cr, resonant inductor Lr, the primary side winding Lp of the first transformer T1, the second transformer T2 load winding W1,
Sampling resistor Rs, bipolar junction transistor Q2 return to the negative output terminal of rectifier bridge DB.Further, since the driving of the second transformer T around
Group W3 obtains positive driving current, and the driving current path of bipolar junction transistor Q2 is sent out via the base stage of bipolar junction transistor Q2
The driving current path of emitter-base bandgap grading knot, bipolar junction transistor Q2 forward conduction, bipolar junction transistor Q1 disconnects.
Terminate in the high level stage of moment t4, clock signal clk, the fourth stage of LED drive circuit 300 terminates.
Figure 10 shows the illustrative circuitry of constant-current control circuit in LED drive circuit according to a third embodiment of the present invention
Figure.LED drive circuit and second embodiment according to a third embodiment of the present invention are the difference is that constant-current control circuit
Circuit structure, remaining aspect is then identical as second embodiment, below the difference of both main descriptions.
Constant-current control circuit 421 includes transistor M1 and M2 and drive module 4211.In the embodiment, transistor M1
It is, for example, MOSFET with M2.Further, the driving end DR1 of constant-current control circuit 421 is directly shorted with ground terminal GND, crystal
Pipe M1 is connected with M2 differential concatenation, organizes pairs of top switch, is connected to the driving end DR2 and ground terminal GND of constant-current control circuit 421
Between.That is, the first end of transistor M1 is connected to the driving end DR2 of constant-current control circuit 421, the first end of transistor M2 connects
It is connected to the ground terminal GND of constant-current control circuit 421, the second end of transistor M1 and M2 are connected to each other.
Drive module 4211 obtains the current sampling signal of resonance current from the sampling resistor Rs of controlled resonant converter 220
CS obtains the voltage feedback signal FB of resonance output voltage from the additional winding Lf of the first transformer T1 of controlled resonant converter 220,
And the switch control signal of transistor M1 and M2 are generated according to current sampling signal CS and voltage feedback signal FB.Drive module
4211 are connected with the control terminal of transistor M1 and M2, for providing the same switch control signal VG to transistor M1 and M2.
LED drive circuit according to this embodiment, a driving end and ground terminal in control circuit are shorted, another drive
Differential concatenation connection transistor M1 and M2 are used as to top switch, so as to save in control circuit between moved end and ground terminal
Negative voltage clamper module (for example, operational amplifier), to simplify circuit structure and reduce circuit cost.
Figure 11 shows the working waveform figure of control circuit shown in Figure 10.It is shown in figure the electric current of the acquisition of drive module 4211
Sampled signal CS, voltage feedback signal FB, clock signal clk and the first transformer T1 exciting current CT1 and the second transformer
The exciting current CT2 of T2 changes with time relationship.
The exciting current CT2 of current sampling signal CS and the second transformer T2 intersect at A, B, C point.Clock signal clk has
The level of high and low (1,0) two states, current sampling signal CS also have the voltage of positive and negative (>0,<0) two states, two-by-two group
It closes, there are four types of different states altogether, to generate the different circuit stages.
Drive module 4211 in the rising edge or failing edge of clock signal clk, constant-current control circuit 421 generates switch
Signal VG is controlled, so that transistor M1 and M2 are connected, it will be short between the driving end DR2 and ground terminal GND of constant-current control circuit 421
It connects.Drive module 4211 in the rising edge or failing edge of voltage feedback signal FB, constant-current control circuit 421 generates switch control
Signal VG processed, so that transistor M1 and M2 end, by the interruption of the driving end DR2 of constant-current control circuit 421 and ground terminal GND
It opens.
Therefore, the first stage of LED drive circuit corresponds to the time period t 0 in figure to t1, and second stage corresponds in figure
Time period t 1 to t2, the phase III corresponds to the time period t 2 in figure to t3, and fourth stage corresponds to the time period t 3 in figure
To t4.
Figure 12 shows the detailed circuit block diagram of constant-current control circuit 321 shown in Fig. 4.The constant-current control circuit 321 is, for example,
The chip of single package.Referring to fig. 4, constant-current control circuit 321 includes transistor M1 and M2, operational amplifier U1 and U2, driving
Module 3211.
Drive module 3211 obtains the current sampling signal of resonance current from the sampling resistor Rs of controlled resonant converter 320
CS obtains the voltage feedback signal FB of DC output voltage from the additional winding Lf of the first transformer T1 of controlled resonant converter 320,
And open signal VG1 and VG2 are provided respectively to transistor M1 and M2.
Further, as shown in figure 12, the drive module 3211 of constant-current control circuit 321 includes output current calculation module
11, peak value current-limiting protection module 12, oscillator 13, logic module 14 and driving stage 15, capacitor C12, constant-current source I11.
It exports current calculation module 11 and thermal compensation signal is generated according to voltage feedback signal FB and resonance current sampling signal CS
Vcomp。
Constant-current source I11 and capacitor C12 are connected in series between feeder ear and ground, are generated slope in the intermediate node of the two and are believed
Number.Two input terminals of oscillator 13 receive ramp signal and thermal compensation signal Vcomp respectively, generate clock signal according to the two
CLK.Logic module 14 generates open signal VG1 and VG2 according to clock signal clk.
In the constant-current control circuit 321, the frequency and resonance current and the first transformer T1 excitation of switch control signal
The average value of the absolute value of electric current CT1 difference is related, that is, according to the frequency of the negative feedback control switch control signal of average value
Rate, so as in the output electric current current constant control of the secondary side of the first transformer T1 of the primary side side of the first transformer T1 realization.
Preferably, constant-current control circuit 321 can also include multiple protective modules, the clamper module including pressure feedback port
16, the clamper module 19 of open-circuit-protection module 17, short circuit protection module 18 and feeder ear, under-voltage locking module 22.In addition,
Constant-current control circuit 321 can also include overvoltage protective module 20, overheat protector module 21.
Figure 13 shows the schematic circuit that current calculation module is exported in constant-current control circuit 321 shown in Figure 12, Figure 14
The schematic illustration that the output electric current of control circuit shown in Figure 12 calculates is shown.
As shown in figure 13, output current calculation module 11 includes operational amplifier AMP1 and AMP2, comparator COMP1, opens
Close K11 and K12, transistor M11 to M16, resistance R11 and R12, capacitor C11.
The non-inverting input terminal of operational amplifier AMP1 receives reference voltage VREF1, and inverting input terminal selectively receives electricity
Stream sampled signal CS or ground connection, output end are connected to the grid of transistor M11.Further, the source electrode of transistor M11 is connected to
The inverting input terminal of operational amplifier AMP1.Transistor M13 and M14 form the first current mirror, transistor M17 and M18 composition the
Two current mirrors, the first current mirror and the second current mirror are coupled to each other.Transistor M11 and transistor M13 is connected in series, and passes through
The electric current of transistor M11 generates the first electric current I11 for flowing through transistor M18 through current mirror coupled.
The non-inverting input terminal of operational amplifier AMP2 receives reference voltage VREF2, and inverting input terminal selectively receives electricity
Stream sampled signal CS or ground connection, output end are connected to the grid of transistor M12.Further, the source electrode of transistor M12 is connected to
The inverting input terminal of operational amplifier AMP2.Transistor M15 and M16 form third current mirror.Transistor M12 and transistor M15
It is connected in series, passes through the electric current of transistor M12 through current mirror coupled, generate the second electric current I12 for flowing through transistor M16.
Transistor M16 and M18 are connected in series, so that the second current mirror and third current mirror are connected in series.Further, electric
Hold C11 to be connected between the intermediate node and ground of transistor M16 and M18, to provide thermal compensation signal at the both ends of capacitor C11
Vcomp。
The non-inverting input terminal and inverting input terminal of comparator COMP1 receives voltage feedback signal FB and reference voltage respectively
VREF3.Switch K11 and K12 are single-pole double-throw switch (SPDT) respectively.The output end of comparator COMP1 is connected to switch K11's and K12
Control terminal, so that switch K11 and K12 switch simultaneously, thus by the inverting input terminal of operational amplifier AMP1 and AMP2 with complementation
Mode is grounded, or be connected to current sampling signal CS via resistance R12 via resistance R11.
In order to be further described, the amplification factor of current mirror is assumed to 1 in following analysis, above-mentioned reference voltage
VREF1, VREF2 and VREF3 are respectively set to 0.85,0.95V, 0.2V.However, the invention is not limited thereto, reference voltage VREF2
It greater than VREF1, and can be any appropriate numerical value respectively.Further, reference voltage signal Vcscc=VREF2- is defined
VREF1。
First stage T1: voltage feedback signal FB is greater than reference voltage VREF3
In the first stage, switch K11 and K12 are switched to the end A by the switch control signal that comparator COMP1 is generated respectively.
The inverting input terminal of operational amplifier AMP2 receives current sampling signal CS, the reverse phase of operational amplifier AMP1 via resistance R12
End is grounded via resistance R11.When current sampling signal CS is zero, operational amplifier AMP2 intermediate node generate second electricity
Flow I12=VREF2/R, wherein the resistance value of R expression resistance R11 and R12.Operational amplifier AMP1 is generated in intermediate node
First electric current I11=VREF1/R, wherein R indicates the resistance value of resistance R11 and R12, therefore, the received electric current of capacitor C11
IDIFF=I12-I11=(VREF2-VREF1)/R=Vcscc/R > 0, charges to capacitor C11.In current sampling signal CS
When less than zero (note represents part of the first stage CS electric current less than 0 at CS1), operational amplifier AMP2 is generated in intermediate node
The second electric current I12=VREF2/R+ | CS1 | Rs/R > 0 *.The first electric current I11 that operational amplifier AMP1 is generated in intermediate node
=VREF1/R, wherein the resistance value of R expression resistance R11 and R12.Therefore, the received electric current IDIFF=I12-I11 of capacitor C11
=(VREF2-VREF1)/R+ | CS1 | * Rs/R=Vcscc/R+ | CS1 | Rs/R > 0 * charges to capacitor C11.
(note represents the part that first stage CS electric current is greater than 0 at CS2), operation when current sampling signal CS is greater than zero
The second electric current I12=VREF2/R-CS2*Rs/R that amplifier AMP2 is generated in intermediate node.Operational amplifier AMP1 is in centre
The I11=VREF1/R that node generates, wherein the resistance value of R expression resistance R11 and R12.Therefore, the received electric current of capacitor C11
IDIFF=I12-I11=(VREF2-VREF1)/R-CS2*Rs/R=Vcscc/R-CS2*Rs/R is in Vcscc/R > CS2*Rs/R
When, received electric current IDIFF=Vcscc/R-CS2*Rs/R > 0 capacitor C11 charges to capacitor C11.Vcscc/R <
When CS2*Rs/R, received electric current IDIFF=Vcscc/R-CS2*Rs/R < 0 capacitor C11 discharges to capacitor C11.
Therefore in the first stage: the average value that capacitor C11 receives electric current IDIFF is equal to:
Since reference voltage signal is DC voltage,
It is the area of S (c) inside Figure 14
It is that the area of S (b) inside Figure 14 is gained knowledge according to geometry it is found that the region area S (a)=S (b)-S
(c).And S (a) is exactly resonance current CS and the first transformer magnetizing current CT1 area defined area, therefore:
Therefore formula (1) becomes:
Definition is Isource to capacitor C11 average eguivalent charging current；Average eguivalent discharge current is Isink
In the first stage, average eguivalent charging current Isource=Vcscc/R.
In the first stage, average eguivalent discharge current Isink is equal to the exciting current of resonance current and the first transformer T1
The average value of the absolute value of CT1 difference:
Wherein, Rs indicates the resistance value of sampling resistor Rs, and R indicates resistance R11 and R12
Resistance value, CS indicates that relevant to resonance current current sampling signal, CT1 indicate that current sample relevant to exciting current is believed
Number.
Second stage T2: voltage feedback signal FB is less than reference voltage VREF3
In the switch control signal that second stage, comparator COMP1 generate, switch K11 and K12 are switched into the end B respectively.
The inverting input terminal of operational amplifier AMP2 is grounded via resistance R11, and the reverse side of operational amplifier AMP1 connects via resistance R12
Receive current sampling signal CS.
When current sampling signal CS is zero, the second electric current I12=that operational amplifier AMP2 is generated in intermediate node
VREF2/R, wherein the resistance value of R expression resistance R11 and R12.The first electric current that operational amplifier AMP1 is generated in intermediate node
I11=VREF1/R, wherein R indicates the resistance value of resistance R11 and R12, therefore, the received electric current IDIFF=I12- of capacitor C11
I11=
(VREF2-VREF1)/R=Vcscc/R > 0 charges to capacitor C11.
(note represents the part that second stage CS electric current is greater than 0 at CS3), operation when current sampling signal CS is greater than zero
The second electric current I12=VREF2/R that amplifier AMP2 is generated in intermediate node.Operational amplifier AMP1 is generated in intermediate node
I11=VREF1-CS3*Rs/R, wherein the resistance value of R expression resistance R11 and R12.Therefore, the received electric current of capacitor C11
IDIFF=I12-I11=(VREF2-VREF1)/R+ | CS | capacitor C11 is filled in Rs/R=Vcscc/R+CS3*Rs/R > 0 *
Electricity.
(note represents part of the second stage CS electric current less than 0 at CS4), operation when current sampling signal CS is less than zero
The second electric current I12=VREF2/R that amplifier AMP2 is generated in intermediate node.Operational amplifier AMP1 is generated in intermediate node
First electric current I11=VREF1/R+ | CS4 | * Rs
/ R > 0, wherein the resistance value of R expression resistance R11 and R12.Therefore, the received electric current IDIFF=of capacitor C11
(VREF2-VREF1)/R- | CS4 | * Rs/R=Vcscc/R- | CS4 | Rs/R > 0 * Vcscc/R > | CS4 | when * Rs/R, capacitor
The received electric current IDIFF of C11
=Vcscc/R- | CS4 | Rs/R > 0 * charges to capacitor C11.
Vcscc/R < | CS4 | when * Rs/R, the received electric current IDIFF of capacitor C11
=Vcscc/R- | CS4 | Rs/R < 0 * discharges to capacitor C11.
Therefore in second stage: the average value that capacitor C11 receives electric current IDIFF is equal to:
Since reference voltage signal is DC voltage,It is similarly available
Therefore formula (3) equally becomes:
In second stage, average eguivalent charging current Isource=Vcscc/R.
In second stage, average eguivalent discharge current Isink is equally equal to the excitation of resonance current and the first transformer T1
The average value of the absolute value of electric current CT1 difference:
Wherein, Rs indicates the resistance value of sampling resistor Rs, and R indicates the resistance value of resistance R11 and R12, and CS is indicated and resonance
The relevant current sampling signal of electric current, CT1 indicate current sampling signal relevant to exciting current.
In output current calculation module 11, the effect of capacitor C11 is in harmonic period to equivalent mean charging current
Isource and average eguivalent discharge current Isink are integrated, to generate thermal compensation signal Vcomp.WhenGreatly
When reference voltage signal Vcscc, at this moment the average value of average eguivalent discharge current Isink is greater than average eguivalent charging current
The average value of Isource, thermal compensation signal Vcomp reduces, so that the frequency of switch control signal reduces, to reduce
WhenWhen less than reference voltage signal Vcscc, at this moment average eguivalent discharge current Isink is averaged
Value is less than the average value of average eguivalent charging current Isource, and thermal compensation signal Vcomp increases, so that the frequency of switch control signal
Rate increases, to increase
WhenWhen equal to reference voltage signal Vcscc, at this moment average eguivalent discharge current Isink is averaged
Value is equal to the average value of average eguivalent charging current Isource, and thermal compensation signal Vcomp remains unchanged.
As shown in figure 14, output current calculation module 11 is according to resonance current signal and the first transformer magnetizing current signal
The average value of absolute value of difference obtain thermal compensation signal, the numerical value of the thermal compensation signal corresponds to resonance current CS and the first transformation
Device exciting current CT1 area defined area S (a)=S (b)-S (c), according to LLC half-bridge principle it is found that the region area with
Export the proportional relationship of electric current.Specifically, switch control signal is generated according to thermal compensation signal, to be shorted the described first ambipolar crystalline substance
The driving current of at least one body pipe and second bipolar junction transistor, to realize the control of resonance frequency.In constant current control
In feedback loop processed, above-mentioned thermal compensation signal Vcomp remains unchanged, that is, maintains corresponding region area to remain unchanged, so that resonance
Frequency maintains constant value corresponding with the output of desired electric current, to realize current constant control.
In the above-described embodiment, the LED drive circuit including charge pump PFC module and LLC resonant converter is described.
It is appreciated that LLC resonant converter can be used alone, and still may be implemented identical based on similar working principle
Technical effect.
In the above-described embodiment, the drive by control upside bipolar junction transistor in LLC resonant converter is described
It moves the short circuit of winding and discharges short-circuit condition in the suitable time, carry out control switch pipe alternate conduction, so that ambipolar
The switch periods of transistor follow the period of circuit internal switch control signal, further according to the negative feedback control of resonance current
Switch control signal obtains frequency, so that resonance frequency maintains constant value corresponding with the output of desired electric current, to realize
Current constant control.However, the invention is not limited thereto.It is appreciated that based on similar working principle, under LLC resonant converter
Identical technical effect also may be implemented in the circuit paths control of the driving winding of side bipolar junction transistor.
It is as described above according to the embodiment of the present invention, these embodiments details all there is no detailed descriptionthe, also not
Limiting the invention is only the specific embodiment.Obviously, as described above, can make many modifications and variations.This explanation
These embodiments are chosen and specifically described to book, is principle and practical application in order to better explain the present invention, thus belonging to making
Technical field technical staff can be used using modification of the invention and on the basis of the present invention well.The present invention is only by right
The limitation of claim and its full scope and equivalent.
Claims (19)
1. a kind of constant-current control circuit for LLC resonant converter, the LLC resonant converter includes the first transformer, the
One bipolar junction transistor and the second bipolar junction transistor, first bipolar junction transistor and second bipolar junction transistor are adopted
It is worked with self-oscillation mode, so that resonance current and exciting current flow through the primary side winding of first transformer, the perseverance
Flow control circuit includes:
Switch element, for being shorted the driving of at least one first bipolar junction transistor and second bipolar junction transistor
Electric current；And
Drive module, including output current calculation module, for calculating resonance current signal and the first transformer magnetizing current letter
Number difference absolute value average value as thermal compensation signal, and the conducting shape of the switch element according to compensation signal control
State is to realize the control of resonance frequency, to realize current constant control.
2. constant-current control circuit according to claim 1, wherein the LLC resonant converter further includes the second transformer,
Second transformer have load winding, and couple with the load winding first driving winding and second drive around
Group, the load winding of second transformer are connected on resonant tank to obtain resonance current, the first driving winding
The different name end of Same Name of Ends and the second driving winding is respectively connected to first bipolar junction transistor and described second bipolar
The base stage of transistor npn npn, to provide the respective drive electric current generated according to the induced current of the resonance current.
3. constant-current control circuit according to claim 2, wherein the switch element will when being shorted the driving current
The Same Name of Ends of the first driving winding and different name end are connected to each other.
4. constant-current control circuit according to claim 2, wherein second transformer further includes control winding, described
The Same Name of Ends of the control winding and different name end are connected to each other by switch element when being shorted the driving current.
5. constant-current control circuit according to claim 3 or 4, wherein the switch element includes:
The first transistor and second transistor, the first transistor are connected to different name end and the ground connection of the first driving winding
Between end, the second transistor is connected between the Same Name of Ends and ground terminal of the first driving winding, and the ground terminal connects
It is connected to the intermediate node of first bipolar junction transistor and second bipolar junction transistor.
6. constant-current control circuit according to claim 5, further includes:
First operational amplifier and second operational amplifier are connected respectively to the control of the first transistor and second transistor
End connection,
Wherein, the drive module provides open signal, and first operational amplifier and the second operational amplifier provide
The switch control signal of cut-off signals, the first transistor and second transistor is the open signal and the cut-off signals
Superposed signal.
7. constant-current control circuit according to claim 6, wherein first operational amplifier and second operation are put
The big respective non-inverting input terminal of device receives negative potential reference voltage, and inverting input terminal is each connected to output end, described in realizing
The Same Name of Ends of the corresponding windings of second transformer and the negative voltage clamper at different name end.
8. constant-current control circuit according to claim 3 or 4, wherein the switch element includes:
The first transistor and second transistor, differential concatenation are connected to the Same Name of Ends of the corresponding windings of second transformer and connect
Between ground terminal, the different name end of the corresponding windings of second transformer and the ground terminal are connected to first bipolar transistor
The intermediate node of pipe and second bipolar junction transistor.
9. constant-current control circuit according to claim 8, wherein the drive module and the first transistor and second
The control terminal of transistor is connected to provide switch control signal.
10. constant-current control circuit according to claim 1, wherein first transformer includes primary side winding and secondary side
Winding, a part of the primary side winding as resonant tank, the vice-side winding couples humorous to provide with the primary side winding
Shake output voltage,
Wherein, the output current calculation module exports electricity according to the current sampling signal of the resonance current and the resonance
The voltage feedback signal of pressure obtains the thermal compensation signal.
11. constant-current control circuit according to claim 10, wherein the output current calculation module includes:
The third transistor that third operational amplifier and output end are connected, for generating the first electric current；
The 4th transistor that four-operational amplifier and output end are connected, for generating the second electric current；
Multiple current mirrors, for by first electric current and second current subtraction to generate equivalent charging current；And
Capacitor, for being integrated to equivalent charging current to generate the thermal compensation signal,
Wherein, the non-inverting input terminal of the third operational amplifier and the four-operational amplifier receives first with reference to electricity respectively
The inverting input terminal of one of pressure and the second reference voltage, the third operational amplifier and the four-operational amplifier receives institute
Current sampling signal is stated, another reverse inter-input-ing ending grounding.
12. constant-current control circuit according to claim 11, wherein the output current calculation module further include:
First switch, for selectively grounded or receiving the electric current and adopting the inverting input terminal of the third operational amplifier
Sample signal；
Second switch, for selectively grounded or receiving the electric current and adopting the inverting input terminal of the four-operational amplifier
Sample signal；
Comparator compares the voltage feedback signal with third reference voltage, to generate the first switch and described
The control signal of second switch.
13. constant-current control circuit according to claim 11, wherein second reference voltage is greater than first reference
Voltage.
14. constant-current control circuit according to claim 1, wherein the drive module further include:
Oscillator generates clock signal according to ramp signal and the thermal compensation signal；And
Logic module generates open signal or switch control signal according to the clock signal.
15. a kind of constant current control method for LLC resonant converter, the LLC resonant converter includes the first transformer, the
One bipolar junction transistor and the second bipolar junction transistor, first bipolar junction transistor and second bipolar junction transistor are adopted
It is worked with self-oscillation mode, so that resonance current and exciting current flow through the primary side winding of first transformer, the perseverance
Method of flow control includes:
The average value of the absolute value of the difference of resonance current signal and the first transformer magnetizing current signal is calculated, compensation letter is obtained
Number；
Realize the control of resonance frequency, according to the on state of the compensation signal control switch element to realize constant current control
System,
Wherein, be shorted in switching elements conductive first bipolar junction transistor and second bipolar junction transistor at least it
One driving current.
16. constant current control method according to claim 15, wherein obtain thermal compensation signal the step of include:
The current sampling signal of resonance current signal is compared with the first reference voltage, to generate the first electric current；
Second electric current is generated using the second reference voltage；
By first electric current and second current subtraction to generate equivalent charging current；And
Equivalent charging current is integrated to generate the thermal compensation signal,
Wherein, second reference voltage is greater than first reference voltage.
17. constant current control method according to claim 16, further includes, institute is switched according to the resonance output voltage signal
The path of current sampling signal is stated, to obtain the difference of the resonance current signal and the first transformer magnetizing current signal
Absolute value average value.
18. constant current control method according to claim 17, wherein refer to the resonance output voltage signal and third
Voltage compares to obtain the path switching signal of the circuit sampling signal.
19. constant current control method according to claim 15 further includes to first bipolar junction transistor and described
The control terminal of two bipolar junction transistors carries out negative voltage clamper.
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Cited By (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN111629496A (en) * | 2020-06-12 | 2020-09-04 | 英飞特电子（杭州）股份有限公司 | Charge pump control circuit and driving power supply |
| WO2022016835A1 (en) * | 2020-07-20 | 2022-01-27 | 华中科技大学 | Current measurement method and apparatus for llc resonant converter |
| CN115025388A (en) * | 2022-06-24 | 2022-09-09 | 深圳邦健生物医疗设备股份有限公司 | Defibrillator high-voltage generation circuit and method based on LLC resonance transformation |
Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN205544948U (en) * | 2015-11-23 | 2016-08-31 | 池州学院 | Hybrid control's LLC resonant transformation ware |
| CN205753505U (en) * | 2016-05-11 | 2016-11-30 | 谢方清 | A kind of resonant type soft-switch charging device |
| CN107395022A (en) * | 2017-07-25 | 2017-11-24 | 杭州士兰微电子股份有限公司 | Mode of resonance switch converters and its control method |
| CN107769563A (en) * | 2016-08-17 | 2018-03-06 | 英飞特电子（杭州）股份有限公司 | A kind of controlled resonant converter |
| US20180183325A1 (en) * | 2016-12-28 | 2018-06-28 | Hangzhou Mps Semiconductor Technology Ltd. | Resonant converter with power factor correction and controller used therein |
| CN209389940U (en) * | 2019-01-21 | 2019-09-13 | 杭州士兰微电子股份有限公司 | Constant-current control circuit for LLC resonant converter |
-
2019
- 2019-01-21 CN CN201910054905.XA patent/CN109639151A/en active Pending
Patent Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN205544948U (en) * | 2015-11-23 | 2016-08-31 | 池州学院 | Hybrid control's LLC resonant transformation ware |
| CN205753505U (en) * | 2016-05-11 | 2016-11-30 | 谢方清 | A kind of resonant type soft-switch charging device |
| CN107769563A (en) * | 2016-08-17 | 2018-03-06 | 英飞特电子（杭州）股份有限公司 | A kind of controlled resonant converter |
| US20180183325A1 (en) * | 2016-12-28 | 2018-06-28 | Hangzhou Mps Semiconductor Technology Ltd. | Resonant converter with power factor correction and controller used therein |
| CN107395022A (en) * | 2017-07-25 | 2017-11-24 | 杭州士兰微电子股份有限公司 | Mode of resonance switch converters and its control method |
| CN209389940U (en) * | 2019-01-21 | 2019-09-13 | 杭州士兰微电子股份有限公司 | Constant-current control circuit for LLC resonant converter |
Cited By (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN111629496A (en) * | 2020-06-12 | 2020-09-04 | 英飞特电子（杭州）股份有限公司 | Charge pump control circuit and driving power supply |
| WO2022016835A1 (en) * | 2020-07-20 | 2022-01-27 | 华中科技大学 | Current measurement method and apparatus for llc resonant converter |
| CN115025388A (en) * | 2022-06-24 | 2022-09-09 | 深圳邦健生物医疗设备股份有限公司 | Defibrillator high-voltage generation circuit and method based on LLC resonance transformation |
| CN115025388B (en) * | 2022-06-24 | 2023-02-28 | 深圳邦健生物医疗设备股份有限公司 | Defibrillator high-voltage generation circuit and method based on LLC resonance transformation |
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