CN109510445A - A kind of inverter change switching frequency modulation method considering inductance saturated characteristic - Google Patents
A kind of inverter change switching frequency modulation method considering inductance saturated characteristic Download PDFInfo
- Publication number
- CN109510445A CN109510445A CN201811289902.6A CN201811289902A CN109510445A CN 109510445 A CN109510445 A CN 109510445A CN 201811289902 A CN201811289902 A CN 201811289902A CN 109510445 A CN109510445 A CN 109510445A
- Authority
- CN
- China
- Prior art keywords
- inductance
- phase
- current
- switching frequency
- switch periods
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
- H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
- H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
- H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/12—Arrangements for reducing harmonics from ac input or output
- H02M1/123—Suppression of common mode voltage or current
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Physics & Mathematics (AREA)
- Electromagnetism (AREA)
- Inverter Devices (AREA)
Abstract
The invention discloses a kind of change switching frequency modulation method for considering inductance saturated characteristic, which controls current ripples distribution by real-time update inverter switching frequency based on the current ripples prediction model for considering inductance saturation.Using change switching frequency modulation method of the invention, in the case where filter inductance saturation, it remains to effectively predict current ripples in each switch periods, and the shake of switching frequency is carried out to inverter using current ripples as control object, traditional PWM method relative to fixed switching frequency, average frequency of switching reduces, directly reduction inverter system switching loss；Secondly, the shake of switching frequency, can effectively reduce system conductive EMI.
Description
Technical field
The invention belongs to field of power electronics, open more particularly, to a kind of inverter change for considering inductance saturated characteristic
Close frequency modulating method.
Technical background
In the inverter system using pulse modulation technology, due to the excitation of PWM voltage, design filter is generally required
Disinthibite high-frequency harmonic.However, the inductance value of design is frequently not an ideal definite value, saturated characteristic of the meeting according to inductance,
With electric current real-time change.When actual inductance value wide variation, the prediction calculating of current ripples can be also affected by it.Existing electricity
The prediction of flow liner wave calculates, and often assumes that the inductance value of filter inductance immobilizes in any operating condition, and there is no consider electricity
The influence that inductance value variation calculates current ripples prediction.In the case where actual inductance saturation, need to propose new computation model,
Current ripples are accurately calculated.Based on this, the change switching frequency PWM based on current ripples prediction model is designed.
Summary of the invention
It is directed to the Improvement requirement of the prior art, the present invention provides a kind of change switching frequencies for considering inductance saturated characteristic
Modulator approach is that the prediction for solving existing current ripples calculates, and there is no consider that inductance value changes to predict to count to current ripples
The technical issues of influence of calculation accurately cannot be predicted and be controlled to current ripples.
To achieve the above object, the present invention provides a kind of inverter change switching frequency modulation side for considering inductance saturated characteristic
Method includes the following steps:
Step (1) determines equivalent inductance, the equivalent inductance of each phase is each according to the relationship of actual inductance and load current
It further include each phase actual inductance on the basis of phase actual inductance to the derivative of load current and the product of load current；
Step (2) carries out time slice to switch periods, predicts the electricity in each split time based on the equivalent inductance
The slope of flow liner wave, and the slope of the time slice situation based on the switch periods and the current ripples of prediction, predicted current
Ripple peak value；
Step (3), current ripples peak value and current ripples limit value based on prediction update the switching frequency of inverter, make
The current ripples peak value for obtaining the next period is equal to current ripples limit value.
In an optional example, in the step (1), kth equivalent inductanceC) are as follows:
Wherein, LkFor kth phase inductance actual value, the load current by being passed through inductance is determined；It is actual inductance to negative
The product of current-carrying derivative and load current, the load current by being passed through inductance determine.
In an optional example, if in the single switch period, equivalent inductanceIt is considered as definite value.
In an optional example, the step (2) specifically comprises the following steps:
Step (2-1) determines each equivalent inductanceThree-phase duty ratio da,db,dcAnd threephase load voltage
ea,eb,ec, within the single switch period, switch periods are divided into multi-section time；
Step (2-2) is based on current ripples prediction model and threephase load voltage ea,eb,ec, determine each mutually every time
The voltage on inductance is acted on, obtains each phase in the current ripples slope of every time；The current ripples prediction model is to examine
Consider the single-phase prediction model of current ripples of inductance saturated characteristic；
Mutually current ripples slope in every time is kth
Wherein, ik_rippleIndicate current ripples, VKN(t) output end between inverter K potential point and N potential point is indicated
Voltage (K=A, B, C), A, B, C and N are the potential point of inverter, VON(t) powder inverter common-mode voltage instantaneous value is indicated,Indicate inverter output end voltage average value in a switch periods,Indicate inverter output end voltage
The average value in a switch periods, t indicate time variable, TsIndicate fixed switch periods；
Step (2-3) passes through three-phase duty ratio and fixed switch periods TS, every section of action time after being segmented is found out respectively
Length；
Step (2-4), the current ripples slope based on every section of action time length and each phase in every time find out each phase
Current ripples peak value.
In an optional example, the equivalent inductance determined based on step (1) determines the common-mode voltage instantaneous value:
In an optional example, the equivalent inductance determined based on step (1) determines the common-mode voltage single switch
Average value in period:
Wherein, t1And t2Used parameter, t when for switch periods being divided into multi-section time1=(dmax-dmid)Ts, t2=
(dmid-dmin)Ts dmax, dmid, dminRespectively three-phase largest duty cycle value, median and minimum value,Indicate a phase load
Average value in the voltage single switch period,Indicate average value in the b phase load voltage single switch period,Indicate c
Average value in the phase load voltage single switch period.
In an optional example, the step (3) includes the following steps:
Switch periods are updated:
Wherein, TNFor the switch periods of update, TsFor fixed switch periods, RrequireFor the corresponding electricity of fixed switch periods
Flow liner wave limit value, Rpeak_kPeak value is predicted for each phase current ripple under fixed switch periods.
In general, through the invention it is contemplated above technical scheme is compared with the prior art, can obtain down and show
Beneficial effect:
(1) the change switching frequency modulation method provided by the invention for considering inductance saturated characteristic, based on consideration inductance saturation
Current ripples prediction model, inductance saturation in the case where, prediction calculating effectively can be carried out to current ripples, be based on electric current line
Wave predicted value makes under the premise of meeting the restriction of current ripples peak value, and realization makes full use of current ripples peak value to limit space, together
When in order to adjust current ripples peak value, having reached reduces switching frequency, realizes effective control to switching frequency.
(2) the change switching frequency modulation method provided by the invention for considering inductance saturated characteristic, switching frequency real-time change,
Corresponding current harmonics possesses wider range of frequency spectrum, to consider that the change switching frequency PWM (VSFPWM) of inductance saturation can be obvious
Reduction fixed switching frequency PWM (CSFPWM) EMI noise peak value, reduce switching loss, improve inverter performance.
Detailed description of the invention
Fig. 1 is typical two level three-phase inverter topological structure schematic diagram；
Fig. 2 is that the general monophase current ripple prediction circuit model of inductance saturated characteristic the considerations of realization according to the present invention shows
It is intended to；
Fig. 3 is the inductance saturation curve schematic diagram of typical magnetic powder core material；
Fig. 4 is to predict schematic diagram according to the current ripples in the single switch period of the present invention；
Fig. 5 is that block diagram is realized in the change switching frequency PWM control of the considerations of present invention realizes inductance saturated characteristic；
Fig. 6 be according to the present invention realize busbar voltage 200V, the index of modulation 0.7 operating condition under three-phase current ripple show
It is intended to；
Fig. 7 is that the operating condition lower switch frequency variation in busbar voltage 200V, the index of modulation 0.7 realized according to the present invention is shown
It is intended to；
Fig. 8 be according to the present invention realize busbar voltage 200V, the index of modulation 0.7 operating condition under EMI comparison schematic diagram.
Specific embodiment
In order to make the objectives, technical solutions, and advantages of the present invention clearer, with reference to the accompanying drawings and embodiments, right
The present invention is further elaborated.It should be appreciated that described herein, specific examples are only used to explain the present invention, not
For limiting the present invention.As long as in addition, technical characteristic involved in the various embodiments of the present invention described below that
Not constituting conflict between this can be combined with each other.
Present invention firstly provides the current ripples prediction models for considering inductance saturation, the prediction model are based on, with three-phase
Current ripples peak value is control object, inverter switching frequency is adjusted in real time, to reduce inverter system switching loss and EMI
Noise peak.
To achieve the above object, it is special to propose a kind of consideration inductance saturation for a technical solution according to the invention
The change switching frequency modulation method of property, the modulator approach include the following steps:
(1) inductance parameters recognize
(2) current ripples are predicted
(3) switch periods update
Further, the step (1) further includes following sub-step:
(1-1) is based on inductance value and current relationship curve, according to threephase load current sampling data, calculates real-time inductance value；
Further, the step (2) further includes following sub-step:
(2-1) is based on prediction model as follows, calculates every section of voltage acted on inductance, and it is oblique to obtain current ripples
Rate；
(2-2) passes through three-phase duty ratio and fixed switch periods TS, every practical function time is found out respectively.
Further, the step (3) further includes following sub-step:
According to the three-phase ripple current peak value relationship directly proportional to switch periods, switch periods are updated:
Next by taking general two level three-phase inverter as an example, Fig. 1 is typical two level three-phase inverter topology for analysis
Structural schematic diagram, as shown in Figure 1, three-phase circuit has by Kirchhoff's second law:
Wherein, R indicates line equivalent resistance, ψaIndicate a phase inductance magnetic linkage, ψbIndicate b phase inductance magnetic linkage, ψcIndicate c phase electricity
Feel magnetic linkage, ia(t) a phase current, i are indicatedb(t) b phase current, i are indicatedc(t) c phase current is indicated.
The right Section 2 can be further written as:
Wherein, the present invention defines three-phase equivalent inductance parameter:
In above formula, LkFor inductance actual value, as shown in Fig. 2, being determined by the load current for being passed through inductance；ikdLk/dikFor reality
Border inductance is equally determined the derivative of load current and the product of load current by the load current for being passed through inductance.Such as Fig. 3 is
The inductance saturation curve schematic diagram of typical magnetic powder core material is based on inductance value and current relationship curve, foundation as seen from Figure 3
Threephase load current sampling data can calculate real-time inductance value, can also calculate actual inductance to the derivative of load current.For letter
Change analytical calculation, it is assumed that in the single switch period, equivalent inductanceIt can be considered definite value.
The single-phase universal current ripples prediction model proposed by the present invention for considering inductance saturation, as shown in Figure 2.Wherein, VKN
For kth phase inverter output end voltage, VONFor three-phase inverter transient common mode voltage,WithRespectively hold
Voltage and the common-mode voltage average value in a switch periods,For equivalent inductance.
Without loss of generality, by taking a phase as an example.In the single switch period, there are three under rising edge and three altogether for three-phase PWM pulse
Edge drops, for entire switch periods are divided into seven sections, as shown in figure 4, every section of time span can be set successively are as follows:Every section of corresponding action time can calculate as follows:
In above formula, dmax, dmid, dminRespectively three-phase largest duty cycle value, median and minimum value, TsFor fixed switch
Period.
Based on current ripples prediction model as shown in Figure 2, in every section of action time, the slope of kth phase current ripple can
It is acquired by following formula:
Compared with the prior art,It is no longer a steady state value, seeking for common-mode voltage is different.
Further, three-phase voltage equation is writeable are as follows:
Common-mode voltage instantaneous value may be expressed as:
Average value is as follows in the common-mode voltage single switch period:
Slope in every section of action time of current known ripple and this section of action time then can be predicted and calculate current ripples
Peak value.
Fig. 5 is that block diagram is realized in the change switching frequency PWM control proposed by the present invention for considering inductance saturated characteristic, is broadly divided into
Three parts 1) 2) current ripples prediction 3) switch periods update for inductance parameters identification.
When inductance parameters identification module receives sampler signal, from controller reading threephase load electric current, according to inductance
Value and load current relation curve, are updated equivalent inductance parameters, are sent into electricity with three-phase duty ratio and threephase load voltage
Flow liner wave prediction module；According to current ripples prediction technique above-mentioned, prediction calculating is carried out to current ripples value, by three-phase current
Ripple predicts that peak value is sent into switch periods update module；
In order to enable the three-phase current ripple peak value in next period is equal to current ripples limit value, mould is updated in switch periods
In block, the switch periods of update are as follows:
In above formula, TNFor the switch periods of update, TsFor fixed switch periods, RrequireIt is corresponding for fixed switch periods
Current ripples limit value, Rpeak_kPeak value is predicted for three-phase current ripple under fixed switch periods.When a complete carrier wave generates
When completion, sampler issues signal, controls into next switch periods.
It is to be appreciated that due to the slope of each section of the current ripples in each switch periods be it is fixed,
If wanting to increase the current ripples peak value of certain phase, need to increase switch time, i.e. increase switch periods.
As shown in fig. 6, Fig. 6 b is to control three-phase current ripple according to the change switching frequency PWM (VSFPWM) that the present invention realizes
As a result, becoming under switching frequency PWM control, either switch compared to fixed switching frequency PWM (CSFPWM) traditional shown in Fig. 6 a
In period, current ripples peak value is still equal to limit value.
The present invention takes full advantage of current ripples head room, and as shown in Figure 6 a, traditional C/S FPWM controls lower current ripples
There are also certain spaces for peak value, and the present invention is by being based on RrequireAnd Rpeak_kUpdate switch periods TN, so that current ripples peak value is more
The nearly current ripples peak limit value of adjunction takes full advantage of current ripples peak value and limits space, while in order to adjust current ripples
Peak value, due to RrequireGreater than Rpeak_k, then switch periods T is increasedN, that is, switching frequency is reduced, when switching frequency reduces then
Switching loss can be reduced, while the EMI noise value of VSFPWM can be reduced.For details, reference can be made to shown in Fig. 7 and Fig. 8.
As shown in fig. 7, be according to the VSFPWM that realizes of the present invention compared with CSFPWM switching frequency.Relative to
CSFPWM, VSFPWM switching frequency change between 13.2kHz and 15kHz, and average frequency of switching reduces by 8%, to directly drop
Low inverter switching device loss.
As shown in figure 8, being the comparison for the VSFPWM and CSFPWM Conducted Electromagnetic Interference (EMI) realized according to the present invention.
VSFPWM can effectively cut the EMI noise peak value of low CSFPWM, and low-frequency range (150kHz-800kHz) EMI noise peak value is reduced
10dB.
Change switching frequency PWM provided by the invention based on current ripples prediction, by changing inverter switching device frequency in real time
Rate controls current ripples distribution, can reduce system switching loss, EMI noise peak value, optimization system performance.Inductance saturation can influence
The prediction of current ripples needs to analyze it calculating so as to directly affect the control accuracy for becoming switching frequency PWM.
As it will be easily appreciated by one skilled in the art that the foregoing is merely illustrative of the preferred embodiments of the present invention, not to
The limitation present invention, any modifications, equivalent substitutions and improvements made within the spirit and principles of the present invention should all include
Within protection scope of the present invention.
Claims (7)
1. a kind of inverter for considering inductance saturated characteristic becomes switching frequency modulation method, which comprises the steps of:
Step (1) determines that equivalent inductance, the equivalent inductance of each phase are real in each phase according to the relationship of actual inductance and load current
It further include each phase actual inductance on the basis of the inductance of border to the derivative of load current and the product of load current；
Step (2) is carried out time slice to switch periods, the electric current line in each split time is predicted based on the equivalent inductance
The slope of wave, and the slope of the time slice situation based on the switch periods and the current ripples of prediction, predicted current ripple
Peak value；
Step (3), current ripples peak value and current ripples limit value based on prediction update the switching frequency of inverter, so that under
The current ripples peak value in a period is equal to current ripples limit value.
2. inverter according to claim 1 becomes switching frequency modulation method, which is characterized in that in the step (1), the
K equivalent inductanceAre as follows:
Wherein, LkFor kth phase inductance actual value, the load current by being passed through inductance is determined；It is actual inductance to load electricity
The derivative of stream and the product of load current, the load current by being passed through inductance determine.
3. inverter according to claim 1 or 2 becomes switching frequency modulation method, which is characterized in that set single switch week
In phase, equivalent inductanceIt is considered as definite value.
4. inverter according to claim 2 becomes switching frequency modulation method, which is characterized in that the step (2) is specific
Include the following steps:
Step (2-1) determines each equivalent inductanceThree-phase duty ratio da,db,dcAnd threephase load voltage ea,eb,
ec, within the single switch period, switch periods are divided into multi-section time；
Step (2-2) is based on current ripples prediction model and threephase load voltage ea,eb,ec, determine each mutually every section of time effect
Voltage on inductance obtains each phase in the current ripples slope of every time；The current ripples prediction model is to consider electricity
Feel the single-phase prediction model of current ripples of saturated characteristic；
Mutually current ripples slope in every time is kth
Wherein, ik_rippleIndicate current ripples, VKN(t) output end voltage (K between inverter K potential point and N potential point is indicated
=A, B, C), A, B, C and N are the potential point of inverter, VON(t) powder inverter common-mode voltage instantaneous value is indicated,
Indicate inverter output end voltage average value in a switch periods,Indicate inverter output end voltage at one
Average value in switch periods, t indicate time variable, TsIndicate fixed switch periods；
Step (2-3) passes through three-phase duty ratio and fixed switch periods TS, every section of action time length after being segmented is found out respectively；
Step (2-4), the current ripples slope based on every section of action time length and each phase in every time find out each phase current
Ripple peak value.
5. inverter according to claim 4 becomes switching frequency modulation method, which is characterized in that determined based on step (1)
Equivalent inductance determine the common-mode voltage instantaneous value:
6. inverter according to claim 4 becomes switching frequency modulation method, which is characterized in that determined based on step (1)
Equivalent inductance determine average value in the common-mode voltage single switch period:
Wherein, t1And t2Used parameter, t when for switch periods being divided into multi-section time1=(dmax-dmid)Ts, t2=(dmid-
dmin)Tsdmax, dmid, dminRespectively three-phase largest duty cycle value, median and minimum value,Indicate a phase load voltage list
Average value in a switch periods,Indicate average value in the b phase load voltage single switch period,Indicate c phase load
Average value in the voltage single switch period.
7. inverter according to claim 1 becomes switching frequency modulation method, which is characterized in that the step (3) includes
Following steps:
Switch periods are updated:
Wherein, TNFor the switch periods of update, TsFor fixed switch periods, RrequireFor the corresponding electric current line of fixed switch periods
Wave limit value, Rpeak_kPeak value is predicted for each phase current ripple under fixed switch periods.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201811289902.6A CN109510445B (en) | 2018-10-31 | 2018-10-31 | Inverter variable switching frequency modulation method considering inductance saturation characteristic |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201811289902.6A CN109510445B (en) | 2018-10-31 | 2018-10-31 | Inverter variable switching frequency modulation method considering inductance saturation characteristic |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN109510445A true CN109510445A (en) | 2019-03-22 |
| CN109510445B CN109510445B (en) | 2020-02-14 |
Family
ID=65747291
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201811289902.6A Active CN109510445B (en) | 2018-10-31 | 2018-10-31 | Inverter variable switching frequency modulation method considering inductance saturation characteristic |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN109510445B (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110417333A (en) * | 2019-07-30 | 2019-11-05 | 南京航空航天大学 | A kind of half primitive period of switching frequency segmentation random space vector pulse duration modulation method |
Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2012531183A (en) * | 2009-06-22 | 2012-12-06 | ボルボ テクノロジー コーポレイション | Method for damping electromechanical vibrations in an electromechanical system and vibration damping system using such a method |
| US20160172841A1 (en) * | 2014-12-10 | 2016-06-16 | Fairchild Korea Semiconductor Ltd. | Protection mode control circuit, switch control circuit including the protection mode control circuit and power supply device including the switch control circuit |
| CN106385196A (en) * | 2016-09-27 | 2017-02-08 | 华中科技大学 | Current ripple real-time prediction model-based three-level voltage source variable switching frequency control method |
| CN106385215A (en) * | 2016-09-27 | 2017-02-08 | 华中科技大学 | Variable switching frequency PWM torque ripple control method for alternating-current motor |
| CN108054948A (en) * | 2017-12-26 | 2018-05-18 | 西安理工大学 | NPC current transformer randomized switching frequency modulator approaches based on ripple current peak value |
-
2018
- 2018-10-31 CN CN201811289902.6A patent/CN109510445B/en active Active
Patent Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2012531183A (en) * | 2009-06-22 | 2012-12-06 | ボルボ テクノロジー コーポレイション | Method for damping electromechanical vibrations in an electromechanical system and vibration damping system using such a method |
| US20160172841A1 (en) * | 2014-12-10 | 2016-06-16 | Fairchild Korea Semiconductor Ltd. | Protection mode control circuit, switch control circuit including the protection mode control circuit and power supply device including the switch control circuit |
| CN106385196A (en) * | 2016-09-27 | 2017-02-08 | 华中科技大学 | Current ripple real-time prediction model-based three-level voltage source variable switching frequency control method |
| CN106385215A (en) * | 2016-09-27 | 2017-02-08 | 华中科技大学 | Variable switching frequency PWM torque ripple control method for alternating-current motor |
| CN108054948A (en) * | 2017-12-26 | 2018-05-18 | 西安理工大学 | NPC current transformer randomized switching frequency modulator approaches based on ripple current peak value |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110417333A (en) * | 2019-07-30 | 2019-11-05 | 南京航空航天大学 | A kind of half primitive period of switching frequency segmentation random space vector pulse duration modulation method |
Also Published As
| Publication number | Publication date |
|---|---|
| CN109510445B (en) | 2020-02-14 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| Lange et al. | Three-level single-phase bridgeless PFC rectifiers | |
| CN106385196A (en) | Current ripple real-time prediction model-based three-level voltage source variable switching frequency control method | |
| Blasko et al. | A new mathematical model and control of a three-phase AC-DC voltage source converter | |
| Biela et al. | Optimal design of a compact 99.3% efficient single-phase PFC rectifier | |
| CN102025145B (en) | Bidirectional chopper-based direct current active power filter and control method thereof | |
| CN109510499B (en) | One kind being suitable for shunt chopper circulation ripple peak control methods and control system | |
| CN109831094A (en) | A kind of the model-free predictive-current control system and its control method of Boost pfc converter | |
| CN108494259A (en) | A kind of control method of high-voltage DC power supply connection in series-parallel combined system | |
| CN112054694B (en) | Bidirectional converter optimization control method and device based on minimum current stress | |
| Kasper et al. | Ultra-high power density server supplies employing GaN power semiconductors and PCB-integrated magnetics | |
| CN109510445A (en) | A kind of inverter change switching frequency modulation method considering inductance saturated characteristic | |
| CN105305483A (en) | Inverter grid connected type power source impedance measurement model optimization method under the condition of external disturbance signals | |
| Şehirli | Examining the impacts of DM filters to PFC isolated Ćuk converter for DCM operation by comparing Si and SiC MOSFET | |
| CN107785905A (en) | A kind of self-adapting type power grid harmonic suppression device integrated system | |
| Fang et al. | Hybrid mode‐hopping modulation for LLC resonant converter achieving high efficiency and linear behaviour | |
| CN103427619B (en) | The PFC control circuit that can boost flexibly and control method thereof | |
| Agrawal | Circulating power flow restricted operation of the isolated bi-directional dual-active bridge DC-DC converter for battery charging applications | |
| Ahmad et al. | Emi standard compliance of three-phase buck type pfc rectifier for application in aircraft | |
| Fan et al. | Zero voltage switching non-isolated bidirectional DC–DC converter with transient current built-up technique | |
| Dai | Modeling and efficiency-based control of interleaved LLC converters for PV DC microgrid | |
| Russi et al. | A simple approach to detect ZVT and determine its time of occurrence for PWM converters | |
| CN106505621B (en) | Inverter and its control method and device | |
| CN112467776B (en) | Current source type converter system, control method and space vector modulation method | |
| KR101062386B1 (en) | Power Factor Correction Method of Matrix Converter and Its System | |
| CN103023291A (en) | Resonance suppression method based on voltage-type grid-connected inverter |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |