CN109412447B - Phase-shifting three-phase high-frequency chain matrix inverter topological structure and modulation method - Google Patents
Phase-shifting three-phase high-frequency chain matrix inverter topological structure and modulation method Download PDFInfo
- Publication number
- CN109412447B CN109412447B CN201811433103.1A CN201811433103A CN109412447B CN 109412447 B CN109412447 B CN 109412447B CN 201811433103 A CN201811433103 A CN 201811433103A CN 109412447 B CN109412447 B CN 109412447B
- Authority
- CN
- China
- Prior art keywords
- switch tube
- controllable switch
- load
- capacitor
- transformer
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
- H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
- H02M5/10—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using transformers
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
- H02M5/02—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc
- H02M5/04—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters
- H02M5/22—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M5/275—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M5/293—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into dc by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
- H02M1/007—Plural converter units in cascade
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Ac-Ac Conversion (AREA)
- Inverter Devices (AREA)
Abstract
The invention discloses a phase-shifting three-phase high-frequency chain matrix type inverter topological structure and a modulation method, wherein the phase-shifting three-phase high-frequency chain matrix type inverter topological structure is formed by sequentially connecting a full-bridge inverter, a transformer T, a matrix converter and an LC filter; the topology structure suppresses voltage oscillation and voltage spike by introducing two clamping diodes and a resonant inductor; the adopted unipolar phase shift control strategy is different from the traditional phase shift control strategy in that the phase shift angle of the unipolar phase shift control strategy is continuously changed, a matrix converter at the rear stage of the transformer can be decoupled into two common voltage type inverters for control, and under the decoupling unipolar phase shift modulation method, zero voltage switching of all controllable switching tubes in the topology can be realized, the loss of the switching tubes can be reduced, and the efficiency of the converter can be improved. The invention has the advantages of less power conversion level, simple control method, high circuit stability and the like.
Description
Technical Field
The invention relates to the field of power electronic power converter topology and modulation, in particular to a phase-shifting three-phase high-frequency chain matrix inverter topology structure and a modulation method.
Background
An inverter is a topological device that converts direct current electrical energy into alternating current electrical energy. The high-frequency chain inverter adopts the transformer to replace the power frequency transformer, and overcomes the defects of large volume, large noise, high cost and the like of the traditional transformer. The conversion process of the high-frequency chain matrix inverter has three power characteristics of DC/HFAC/LFAC, wherein HFAC: high frequency ac, LFAC: low frequency ac. It is known that in such inverters a DC/AC, i.e. direct current/alternating current inversion link occurs, which is located at the primary side of the transformer, and an AC/AC, i.e. alternating current/alternating current conversion link, which is also commonly referred to as a matrix converter link, is present at the secondary side of the transformer. Compared with the traditional converter, the matrix converter has no intermediate energy storage link, adopts a bidirectional switch, can realize bidirectional flow of energy, has compact structure, small volume and high efficiency, and can independently control the amplitude and the frequency of output voltage.
Because of the existence of leakage inductance of the transformer, when the high-frequency chain matrix inverter commutates, larger voltage overshoot is generated on the power tube of the secondary side matrix converter of the transformer, so the safe commutation of the secondary side matrix converter of the transformer is always a technical difficulty for restricting the high-frequency chain inverter to realize large-scale popularization. At present, the following safe commutation strategies mainly exist: (1) the soft switch can be realized by adding active clamp to restrain voltage overshoot, but the introduced clamp circuit increases cost, and the added controllable power tube also makes control more complex; (2) the unipolar and bipolar phase-shifting control strategy realizes the natural commutation of the inductive current by means of the commutation overlapping of the matrix converter, and realizes the ZVS of the power tube, but has the problems of difficult control of the commutation overlapping time and the like; (3) the series resonant circuit is introduced into the pre-stage inverter to realize soft commutation of the power tube, at the moment, the switching of the power tube is required to occur at the zero current moment, and the resonant working state of the resonant circuit needs to be judged for controlling the output energy, so that the control mode is complex.
However, although the above strategy can realize safe commutation, the modulation and control of the inverter are more complex, resulting in reduced system reliability, which affects the popularization and use of such converters.
Disclosure of Invention
Aiming at the technical problems, the invention aims to provide a topological structure of a phase-shifting three-phase high-frequency chain matrix inverter and a modulation method. The invention has the advantages of low power conversion level and simple modulation, and can realize the unipolar front-stage phase-shifting modulation method by combining voltage type decoupling.
In order to achieve the above object, the present invention is realized according to the following technical scheme:
the invention relates to a topological structure of a phase-shifting three-phase high-frequency chain matrix inverter, which is characterized by comprising a full-bridge inverter, a transformer T, a matrix converter and an LC filter, wherein all components are sequentially connected;
wherein the full-bridge inverter is formed by direct-current input voltage U i Controllable switch tube S 1 Controllable switch tube S 2 Controllable switch tube S 3 Controllable switch tube S 4 Diode D C1 Diode D C2 Inductance L r Composition;
the matrix converter is formed by a controllable switch tube S 1a Controllable switch tube S 4b Controllable switch tube S 4a Controllable switch tube S 1b Controllable switch tube S 3a Controllable switch tube S 6b Controllable switch tube S 6a Controllable switch tube S 3b Controllable switch tube S 5a Controllable switch tube S 2b Controllable switch tube S 2a Controllable switch tube S 5b Composition;
the LC type filter consists of a first inductor L f1 Second inductance L f2 Third inductance L f3 First capacitor C f1 A second capacitor C f2 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Composition;
the DC input voltage U i Positive pole of (a) is respectively connected with the controllable switch tube S 1 Collector, diode D of (c) C1 Positive electrode of (a) controllable switch tube S 3 Is connected with the collector of the DC input voltage U i The negative pole of (a) is respectively connected with the controllable switch tube S 2 Emitter, diode D of (2) C2 Is a negative electrode of a controllable switch tube S 4 Is connected with the emitter of the (C);
controllable switch tube S 1 Emitter of (a) is respectively connected with inductance L r One end of a controllable switch tube S 2 Is connected with the collector of the capacitor; controllable switch tube S 3 The emitter of the transformer T is respectively connected with one end of the primary side of the transformer T and the controllable switch tube S 4 Is connected with the collector of the inductor L r Respectively with diode D C1 Cathode of (D) diode D C2 The positive electrode of the transformer T is connected with the other end of the primary side of the transformer T;
one end of the secondary side of the transformer T is respectively connected with a controllable switch tube S 1a Collector of (a) controllable switch tube S 3a Collector of (a) controllable switch tube S 5a The other end of the secondary side of the transformer T is respectively connected with a controllable switch tube S 1b Collector of (a) controllable switch tube S 3b Collector of (a) controllable switch tube S 5b Is connected with the collector of the capacitor; controllable switch tube S 1a Emitter and controllable switch tube S 4b Emitter of (C) is connected with a controllable switch tube S 3a Emitter and controllable switch tube S 6b Emitter of (C) is connected with a controllable switch tube S 5a Emitter and controllable switch tube S 2b Is connected with the emitter of the (C); controllable switch tube S 1b Emitter and controllable switch tube S 4a Emitter of (C) is connected with a controllable switch tube S 3b Emitter and controllable switch tube S 6a Emitter of (C) is connected with a controllable switch tube S 5b Emitter and controllable switch tube S 2a Is connected with the emitter of the (C);
controllable switch tube S 4a Collector and controllable switch tube S 4b Is connected with the collector of the first inductor L f1 One end is connected with the first inductance L f1 The other end is connected with the first capacitor C f1 One end, load R 1 One end is connected with a load R 1 The other end is respectively connected with the load R 2 Load R 3 Are connected; first capacitor C f1 The other end is respectively connected with the second capacitor C f2 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Are connected;
controllable switch tube S 6a Collector and controllable switch tube S 6b Is connected with the second inductance L f2 One end is connected with a second inductor L f2 The other end is connected with a second capacitor C f2 Load R 2 One end is connected with a load R 2 The other end is respectively connected with the load R 1 Load R 3 Are connected; second capacitor C f2 The other end is respectively connected with the first capacitor C f1 Third capacitor C f3 Load(s)R 1 Load R 2 Load R 3 Are connected;
controllable switch tube S 2a Collector and controllable switch tube S 2b Is connected with the third inductance L f3 One end is connected with the third inductor L f3 The other end is connected with a third capacitor C f3 Load R 3 One end is connected with a load R 3 The other end is respectively connected with the load R 1 Load R 2 Are connected; third capacitor C f3 The other end is respectively connected with the first capacitor C f1 A second capacitor C f2 Load R 1 Load R 2 Load R 3 Are connected.
The invention provides a modulation method for modulating a topological structure of a phase-shifting three-phase high-frequency chain matrix inverter, which is characterized by comprising the following steps of: the phase-shifting matrix inverter adopts PWM control of duty ratio change, and the controllable switch tube S 1 Controllable switch tube S 4 Bridge arm and controllable switch tube S 2 Controllable switch tube S 3 The bridge arms are alternately conducted; the series resonance inductor realizes zero-voltage switching of the switching tube; the voltage of the primary side of the transformer T connected with the two clamping diodes is clamped at 0 and U by the clamping diodes i Between them; the matrix converter of the rear stage of the transformer T is equivalent to two groups of common voltage type inverters for decoupling control, and the controllable switch tube S is controlled according to a voltage type decoupling logic modulation circuit 1a Controllable switch tube S 6a Controllable switch tube S 1b Controllable switch tube S 6b Control is performed to convert the high-frequency resonant current outputted from the transformer T into a low-frequency pulsating current.
Further, the θ angle is continuously changed by unipolar phase shift control in the duty ratio, where o is the modulation degree m<m<1, the variation range is 2 (1-m) 180 DEG<θ<180 °; in a switching period T s During the time, switch tube S 1 And S is equal to 4 、S 2 And S is equal to 3 The common on time of (2) is: t (T) com ＝T s (180°-θ)/360°。
Further, the matrix converter of the rear stage of the transformer T adopts voltage type decoupling controlThe switching tube of the matrix converter is divided into positive and negative groups, namely a controllable switching tube S 1a ∽S 6a And a controllable switch tube S 1b ∽S 6b When the positive group switching tube works, the negative group switching tube is all conducted, and when the negative group switching tube works, the positive group is all conducted, and the rear-stage matrix converter is equivalent to two groups of common current type inverters.
Compared with the prior art, the invention has the following advantages:
the application of the resonant inductor of the invention enables the primary side power switching tube of the transformer to realize zero-voltage switching, and the two clamping diodes are introduced to inhibit voltage oscillation and voltage spike of the secondary side switching tube of the transformer, thus reducing very high voltage spike generated at two ends of the switching tube due to breaking the flow path of the leakage inductance and the filter inductance current of the transformer, and inhibiting series resonance caused by the leakage inductance of the transformer and parasitic capacitance of the switching tube, so that the loss of the switch is reduced, and the reliability and the efficiency of the circuit are improved.
Drawings
In order to more clearly illustrate the embodiments of the invention or the technical solutions in the prior art, the drawings that are required in the embodiments or the description of the prior art will be briefly described, it being obvious that the drawings in the following description are only some embodiments of the invention, and that other drawings may be obtained from these drawings without inventive effort for a person skilled in the art.
FIG. 1 is a schematic diagram of an inverter circuit topology according to the present invention;
FIG. 2 is a schematic waveform diagram of the inverter operating state in a high frequency period according to the present invention;
FIG. 3 is a schematic diagram of a secondary matrix transformer in voltage-type decoupling;
fig. 4 (a) -4 (l) are schematic diagrams of a front-stage phase-shifting three-phase high-frequency chain matrix inverter according to the present invention in a high-frequency cycle.
Detailed Description
For the purpose of making the objects, technical solutions and advantages of the embodiments of the present invention more apparent, the technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention, and it is apparent that the described embodiments are some embodiments of the present invention, but not all embodiments of the present invention.
As shown in fig. 1, the topological structure of the inverter comprises a full-bridge inverter, a transformer T, a matrix converter and an LC filter, which are sequentially connected to form;
the full bridge inverter is powered by DC input voltage U i Controllable switch tube S 1 Controllable switch tube S 2 Controllable switch tube S 3 Controllable switch tube S 4 Diode D C2 Diode D C2 Inductance L r Composition;
the matrix converter is formed by a controllable switch tube S 1a Controllable switch tube S 4b Controllable switch tube S 4a Controllable switch tube S 1b Controllable switch tube S 3a Controllable switch tube S 6b Controllable switch tube S 6a Controllable switch tube S 3b Controllable switch tube S 5a Controllable switch tube S 2b Controllable switch tube S 2a Controllable switch tube S 5b Composition;
the LC filter is composed of a first inductor L f1 Second inductance L f2 Third inductance L f3 First capacitor C f1 A second capacitor C f2 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Composition;
DC input voltage U i Positive pole of (a) is respectively connected with the controllable switch tube S 1 Collector, diode D of (c) C1 Positive electrode of (a) controllable switch tube S 3 Is connected with the collector of the DC input voltage U i The negative pole of (a) is respectively connected with the controllable switch tube S 2 Emitter, diode D of (2) C2 Is a negative electrode of a controllable switch tube S 4 Is connected with the emitter of the (C);
controllable switch tube S 1 Emitter of (a) is respectively connected with inductance L r One end of a controllable switch tube S 2 Is connected with the collector of the capacitor; controllable switch tube S 3 The emitter of the transformer T is respectively connected with one end of the primary side of the transformer T and the controllable switch tube S 4 Is connected with the collector of the capacitor;inductance L r And diode D C1 Cathode of (D) diode D C2 The positive electrode of the transformer T is connected with one end of the primary side of the transformer T;
one end of the secondary side of the transformer T is respectively connected with a controllable switch tube S 1a Collector of (a) controllable switch tube S 3a Collector of (a) controllable switch tube S 5a The other end of the secondary side of the transformer T is respectively connected with a controllable switch tube S 1b Collector of (a) controllable switch tube S 3b Collector of (a) controllable switch tube S 5b Is connected with the collector of the capacitor; controllable switch tube S 1a Emitter and controllable switch tube S 4b Emitter of (C) is connected with a controllable switch tube S 3a Emitter and controllable switch tube S 6b Emitter of (C) is connected with a controllable switch tube S 5a Emitter and controllable switch tube S 2b Is connected with the emitter of the (C); controllable switch tube S 1b Emitter and controllable switch tube S 4a Emitter of (C) is connected with a controllable switch tube S 3b Emitter and controllable switch tube S 6a Emitter of (C) is connected with a controllable switch tube S 5b Emitter and controllable switch tube S 2a Is connected with the emitter of the (C);
controllable switch tube S 4a Collector and controllable switch tube S 4b Is connected with the collector of the first inductor L f1 One end is connected with the first inductance L f1 The other end is connected with the first capacitor C f1 One end, load R 1 One end is connected with a load R 1 The other end is respectively connected with the load R 2 Load R 3 Are connected; first capacitor C f1 The other end is respectively connected with the second capacitor C f2 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Are connected;
controllable switch tube S 6a Collector and controllable switch tube S 6b Is connected with the collector of the second inductor L f2 One end is connected with a second inductor L f2 The other end is connected with a second capacitor C f2 Load R 2 One end is connected; load R 2 The other end is respectively connected with the load R 1 Load R 3 Are connected; second capacitor C f2 The other end is respectively connected with the firstCapacitor C f1 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Are connected;
controllable switch tube S 2a Collector and controllable switch tube S 2b Is connected with the collector of the third inductor L f3 One end is connected with the third inductor L f3 The other end is connected with a third capacitor C f3 Load R 3 One end is connected with a load R 3 The other end is respectively connected with the load R 1 Load R 2 Are connected; third capacitor C f3 The other end is respectively connected with the first capacitor C f1 A second capacitor C f2 Load R 1 Load R 2 Load R 3 Are connected.
The invention adds clamping diode and resonant inductance to the front-stage high-frequency inverter of the transformer, adopts a PWM control method with a certain duty ratio to drive the right bridge arm driving signals S of the front and the inverter 4 (S 3 ) Relative to left arm drive signal S 1 (S 2 ) There is a phase shift angle θ. The matrix converter of the rear stage of the transformer adopts a modulation method combining PWM and voltage type decoupling to convert high-frequency alternating current output by the transformer into power frequency current.
The invention provides a modulation method for modulating a topological structure of a three-phase high-frequency chain matrix inverter by utilizing the phase shift, which comprises the following steps: the phase-shifting matrix inverter adopts PWM control of duty ratio change, and the controllable switch tube S 1 Controllable switch tube S 4 Bridge arm and controllable switch tube S 2 Controllable switch tube S 3 The bridge arms are alternately conducted; the series resonance inductor realizes zero-voltage switching of the switching tube; the voltage of the primary side of the transformer T connected with the two clamping diodes is clamped at 0 and U by the clamping diodes i Between them; the matrix converter of the rear stage of the transformer T is equivalent to two groups of common voltage type inverters for decoupling control, and the controllable switch tube S is controlled according to a voltage type decoupling logic modulation circuit 1a Controllable switch tube S 6a Controllable switch tube S 1b Controllable switch tube S 6b Control is performed to convert the high-frequency resonant current outputted from the transformer T into a low-frequency pulsating current.
The angle theta of the unipolar phase shift control of the duty cycle is constantly changing, where o is in the case of a modulation degree of m<m<1, the variation range is 2 (1-m) 180 DEG<θ<180 °; in a switching period T s During the time, switch tube S 1 And S is equal to 4 、S 2 And S is equal to 3 The common on time of (2) is: t (T) com ＝T s (180°-θ)/360°。
The matrix converter of the later stage of the transformer T adopts voltage type decoupling control to decompose the switching tube of the matrix converter into positive and negative groups, namely a controllable switching tube S 1a ∽S 6a And a controllable switch tube S 1b ∽S 6b When the positive group switching tube works, the negative group switching tube is all conducted, and when the negative group switching tube works, the positive group is all conducted, and the rear-stage matrix converter is equivalent to two groups of common voltage type inverters.
Fig. 2 is a schematic waveform diagram of the working state in a high frequency period according to the present invention. S in the figure 1 、S 4 And S is 2 、S 3 Is a driving signal of a power tube of a front-stage high-frequency inverter of the transformer, S 1a ∽S 6a 、S 1b ∽S 6b Is a driving signal of a power tube of a transformer rear-stage matrix converter. i.e p Current waveform of primary side of transformer, i DC1 、i DC2 For flowing through diode D C1 、D C2 Is set in the above-described range). As can be seen from fig. 2, the duty cycle of the pre-stage power switch tube driving signal is varied; at the same time, it can be seen that the diode D is in one period of the topology C1 、D C2 Each turn on once, clamp the voltage of the primary side of the transformer, the input voltage of the primary side of the transformer is zero, and the switching moment of the positive and negative groups of the switching tube is provided for the later stage, so that the voltage peak generated by the leakage current of the transformer can be avoided.
Fig. 3 is a schematic diagram of circuit decoupling of a transformer post matrix converter. The modulation method enables the matrix converter to be equivalently decomposed into two common voltage-type inverters. S of the positive group inverter when the input current of the transformer is positive 1a 、S 2a 、S 3a 、S 4a 、S 5a 、S 6a In a three-phase modulation state, a negative group inverter S 1b 、S 2b 、S 3b 、S 4b 、S 5b 、S 6b Is in a conducting state; when the input current signal of the transformer is negative, the negative group inverter S 1b 、S 2b 、S 3b 、S 4b 、S 5b 、S 6b Three-phase modulation state, S of positive group inverter 1a 、S 2a 、S 3a 、S 4a 、S 5a 、S 6a In an on state.
Fig. 4 is a schematic diagram of a mode circuit of the phase-shifting three-phase high-frequency chain matrix inverter in a high-frequency period, wherein the high-frequency chain is a circuit structure of replacing a power frequency transformer with a high-frequency transformer in a circuit. FIG. 4 (a) -FIG. 4 (l) shows the following working modes 1-12, respectively. Assuming that all components in the topology are ideal components, according to the working principle, 12 working states exist in a high-frequency period, and the specific modal analysis is as follows:
(1) Working modality 1[t 0 -t 1 ]，t 0 S before moment 1 、S 4 Has been in a conductive state, i p Warp S 1 、S 4 The circulation starts to transfer energy to the rear stage, the positive group inverter in the matrix converter works, the switching tubes in the negative group inverter are all in a conducting state, and the inductance current i is filtered Lf The linear rise starts.
(2) Working modality 2[t 1 -t 2 ]，t 1 Time switch tube S 1 Triggering turn-off, switching tube S for preventing bridge arm from direct connection 2 Should add an appropriate dead zone. Primary current i p For C 1 Charging while giving C 2 And (5) discharging. Because the primary and secondary side voltages of the transformer are zero in the stage, the switching tube S 1 Zero voltage turn-off can be achieved. Output filter inductor current i due to primary side current clamping Lf The state of the last mode is still maintained, the linear decline is started, the b-phase capacitor in the filter network is charged, the a-phase capacitor and the c-phase capacitor are discharged, and at the moment, the switching tubes of the matrix converter are all conducted.
(3) Working modality 3[t 2 -t 3 ]，t 2 Time capacitor C 1 、C 2 End of charge and discharge，i p Warp D 2 、S 4 Circulation, this stage triggers the switching tube S 2 The switch is turned on, and zero voltage switch-on can be realized. In this stage, the primary and secondary side voltages of the transformer are zero, and the switching tubes of the matrix converter are all conducted to output the filter inductance current i Lf And the filter network is still in the follow current state of the previous stage, the b and c phase capacitors are charged, and the a phase capacitor is discharged.
(4) Working modality 4[t 3 -t 4 ]，t 3 Time switch tube S 4 Triggering off, primary side current i p For C 4 Charging while giving C 3 Discharging, switching tube S 4 Zero voltage turn-off can be achieved. Switching tube S of matrix converter at this stage 4b 、S 5b 、S 6b And closing, and keeping the state of the last mode by the output filter network.
(5) Working modality 5[t 4 -t 5 ]，t 4 Time capacitor C 4 、C 3 At the end of charge and discharge, the power tube S in this stage 3 Realize zero voltage turn-on, the leakage inductance of the transformer and parasitic capacitance of the post-stage switching tube generate resonance, and meanwhile, the voltage at two ends of the transformer is-U i Therefore, the voltage at the point B is zero, so that the clamping diode D C2 Conducting, U is BC Clamped at-U i . Due to t 4 At the moment when the secondary side voltage of the transformer is zero, the input direct current voltage is reversely added to Lr, ir and ip rapidly drop to zero, and reversely increase. Due to the action of the resonant inductor, the voltage at the two ends of the transformer rises in a reverse linear way. The output filter network maintains the state of the previous mode.
(6) Working modality 6[t 5 -t 6 ]During this phase, the primary side provides energy to the secondary side, i p Equal to i r Filtering inductor current i Lf The output filter network maintains the state of the previous mode in a linear rising manner.
(7) Working modality 7[t 6 -t 7 ]，t 6 At the moment, switch tube S 1b Turn off, S 4b And conducting, charging the phase a and the phase b of the capacitor in the filter network, and discharging the phase c of the capacitor.
(8) Working modality 8[t 7 -t 8 ]，t 7 Time to openClosing tube S 2 Triggering off, primary side current i p For C 2 Charging while giving C 1 Discharging, switching tube S 2 Zero voltage turn-off can be realized, the primary side voltage and the secondary side voltage of the transformer are zero in the stage, and the switching tubes of the matrix converter are all conducted to output filter inductance current i Lf And the filter network is still in the circulation state of the last stage, the b-phase capacitor is charged, and the a-phase capacitor and the c-phase capacitor are discharged.
(9) Working modality 9[t 8 -t 9 ]，t 8 Time capacitor C 2 、C 1 End of charge and discharge, i p Warp D 1 、S 3 Flow through, trigger power tube S in this stage 1 And the power-on can be realized, and the zero voltage power-on can be realized. The voltage at two ends of the transformer is zero in this stage, the switching tubes of the matrix converter are all conducted, and the filter inductance current i is output Lf And the filter network is still in the follow current state of the previous stage, the b and c phase capacitors are charged, and the a phase capacitor is discharged.
(10) Working mode 10[ t ] 9 -t 10 ]，t 9 Time switch tube S 3 Triggering off, primary side current i p For C 3 Charging while giving C 4 Discharging, switching tube S 3 Zero voltage turn-off can be achieved. In this stage, the primary and secondary voltages of the transformer are all zero, and the output filter network maintains the state of the previous mode.
(11) Working mode 11[ t ] 10 -t 11 ]，t 10 Time of day, capacitance C 3 、C 4 At the end of charge and discharge, the power tube S in this stage 4 Realizing zero voltage turn-on, and simultaneously, the voltage at two ends of the transformer is U i The voltage at the point B is U i Make the clamping diode D C1 Conducting, U is BC Clamped at U i 。i p After rapidly falling to zero, the forward direction rapidly increases, and the matrix converter switching tube S 4a 、S 5a 、S 6a And closing. And the b and c phase capacitors in the filter network are charged, and the a phase capacitor is discharged.
(12) Working mode 12[ t ] 11 -t 12 ]，t 11 At the moment, clamping diode D C1 Closing, the output filter network keeps the state of the last mode, and the power is suppliedThe way goes to the next cycle.
According to the working process, the three-phase high-frequency chain inverter with the clamping diode by adopting the decoupling unipolar phase-shift modulation method can realize zero-voltage switching of the switching tube in topology, and voltage oscillation caused by leakage inductance of the transformer and parasitic capacitance of the switching tube is greatly reduced.
While the invention has been described in detail in the foregoing general description and specific examples, it will be apparent to those skilled in the art that modifications and improvements can be made thereto. Accordingly, such modifications or improvements may be made without departing from the spirit of the invention and are intended to be within the scope of the invention as claimed.
Claims (4)
1. The topological structure of the phase-shifting three-phase high-frequency chain matrix inverter is characterized by comprising a full-bridge inverter, a transformer T, a matrix converter and an LC filter, wherein the components are sequentially connected;
wherein the full-bridge inverter is formed by direct-current input voltage U i Controllable switch tube S 1 Controllable switch tube S 2 Controllable switch tube S 3 Controllable switch tube S 4 Diode D C1 Diode D C2 Inductance L r Composition;
the matrix converter is formed by a controllable switch tube S 1a Controllable switch tube S 4b Controllable switch tube S 4a Controllable switch tube S 1b Controllable switch tube S 3a Controllable switch tube S 6b Controllable switch tube S 6a Controllable switch tube S 3b Controllable switch tube S 5a Controllable switch tube S 2b Controllable switch tube S 2a Controllable switch tube S 5b Composition;
the LC type filter consists of a first inductor L f1 Second inductance L f2 Third inductance L f3 First capacitor C f1 A second capacitor C f2 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Composition;
the DC input voltage U i Positive pole of (a) is respectively connected with the controllable switch tube S 1 Collector, diode D of (c) C1 Positive electrode of (a) controllable switch tube S 3 Is connected with the collector of the DC input voltage U i The negative pole of (a) is respectively connected with the controllable switch tube S 2 Emitter, diode D of (2) C2 Is a negative electrode of a controllable switch tube S 4 Is connected with the emitter of the (C);
controllable switch tube S 1 Emitter of (a) is respectively connected with inductance L r One end of a controllable switch tube S 2 Is connected with the collector of the capacitor; controllable switch tube S 3 The emitter of the transformer T is respectively connected with one end of the primary side of the transformer T and the controllable switch tube S 4 Is connected with the collector of the inductor L r Respectively with diode D C1 Cathode of (D) diode D C2 The positive electrode of the transformer T is connected with the other end of the primary side of the transformer T;
one end of the secondary side of the transformer T is respectively connected with a controllable switch tube S 1a Collector of (a) controllable switch tube S 3a Collector of (a) controllable switch tube S 5a The other end of the secondary side of the transformer T is respectively connected with a controllable switch tube S 1b Collector of (a) controllable switch tube S 3b Collector of (a) controllable switch tube S 5b Is connected with the collector of the capacitor; controllable switch tube S 1a Emitter and controllable switch tube S 4b Emitter of (C) is connected with a controllable switch tube S 3a Emitter and controllable switch tube S 6b Emitter of (C) is connected with a controllable switch tube S 5a Emitter and controllable switch tube S 2b Is connected with the emitter of the (C);
controllable switch tube S 1b Emitter and controllable switch tube S 4a Emitter of (C) is connected with a controllable switch tube S 3b Emitter and controllable switch tube S 6a Emitter of (C) is connected with a controllable switch tube S 5b Emitter and controllable switch tube S 2a Is connected with the emitter of the (C);
controllable switch tube S 4a Collector and controllable switch tube S 4b Is connected with the collector of the first inductor L f1 One end is connected with the first inductance L f1 The other end is connected with the first capacitor C f1 One end is provided withLoad R 1 One end is connected with a load R 1 The other end is respectively connected with the load R 2 Load R 3 Are connected; first capacitor C f1 The other end is respectively connected with the second capacitor C f2 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Are connected;
controllable switch tube S 6a Collector and controllable switch tube S 6b Is connected with the second inductance L f2 One end is connected with a second inductor L f2 The other end is connected with a second capacitor C f2 Load R 2 One end is connected with a load R 2 The other end is respectively connected with the load R 1 Load R 3 Are connected; second capacitor C f2 The other end is respectively connected with the first capacitor C f1 Third capacitor C f3 Load R 1 Load R 2 Load R 3 Are connected;
controllable switch tube S 2a Collector and controllable switch tube S 2b Is connected with the third inductance L f3 One end is connected with the third inductor L f3 The other end is connected with a third capacitor C f3 Load R 3 One end is connected with a load R 3 The other end is respectively connected with the load R 1 Load R 2 Are connected; third capacitor C f3 The other end is respectively connected with the first capacitor C f1 A second capacitor C f2 Load R 1 Load R 2 Load R 3 Are connected.
2. A modulation method for topological structure modulation of a phase-shifting three-phase high-frequency chain matrix inverter according to claim 1, characterized by comprising the steps of: the phase-shifting matrix inverter adopts PWM control of duty ratio change, and the controllable switch tube S 1 Controllable switch tube S 4 Bridge arm and controllable switch tube S 2 Controllable switch tube S 3 The bridge arms are alternately conducted; the series resonance inductor realizes zero-voltage switching of the switching tube; the voltage of the primary side of the transformer T connected with the two clamping diodes is clamped at 0 and U by the clamping diodes i Between them; the matrix converter of the rear stage of the transformer T is equivalent to two groups of common voltagesThe decoupling control is carried out on the inverter, and the controllable switch tube S is controlled according to the voltage type decoupling logic modulation circuit 1a Controllable switch tube S 6a Controllable switch tube S 1b Controllable switch tube S 6b Control is performed to convert the high-frequency resonant current outputted from the transformer T into a low-frequency pulsating current.
3. The phase shift modulation method according to claim 2, wherein: the angle theta is continuously changed by adopting unipolar phase shift control in the duty ratio, and in the case of a modulation degree of m, o is<m<1, the variation range is 2 (1-m) 180 DEG<θ<180 °; in a switching period T s Controllable switch tube S in time 1 And S is equal to 4 、S 2 And S is equal to 3 The common on time of (2) is: t (T) com ＝T s (180°-θ)/360°。
4. The modulation method according to claim 2, characterized in that: the matrix converter of the later stage of the transformer T adopts voltage type decoupling control to decompose the switching tube of the matrix converter into positive and negative groups, namely a controllable switching tube S 1a ∽S 6a And a controllable switch tube S 1b ∽S 6b When the positive group switching tube works, the negative group switching tube is all conducted, and when the negative group switching tube works, the positive group is all conducted, and the rear-stage matrix converter is equivalent to two groups of common voltage type inverters.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201811433103.1A CN109412447B (en) | 2018-11-28 | 2018-11-28 | Phase-shifting three-phase high-frequency chain matrix inverter topological structure and modulation method |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN201811433103.1A CN109412447B (en) | 2018-11-28 | 2018-11-28 | Phase-shifting three-phase high-frequency chain matrix inverter topological structure and modulation method |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN109412447A CN109412447A (en) | 2019-03-01 |
| CN109412447B true CN109412447B (en) | 2023-08-22 |
Family
ID=65456008
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN201811433103.1A Active CN109412447B (en) | 2018-11-28 | 2018-11-28 | Phase-shifting three-phase high-frequency chain matrix inverter topological structure and modulation method |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN109412447B (en) |
Families Citing this family (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN112713795A (en) * | 2020-12-30 | 2021-04-27 | 燕山大学 | Single-phase high-frequency chain matrix type inverter and modulation method |
Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2002204579A (en) * | 2001-01-09 | 2002-07-19 | Fuji Electric Co Ltd | Control method for inverter |
| CN103138614A (en) * | 2013-03-01 | 2013-06-05 | 南京航空航天大学 | Double-normal-shock inverter |
| WO2016150245A1 (en) * | 2015-03-23 | 2016-09-29 | 深圳市皓文电子有限公司 | Dc/dc converter |
| CN106787914A (en) * | 2017-03-03 | 2017-05-31 | 燕山大学 | LC series resonance-type three-phases high frequency chain matrix inverter topology and modulator approach |
| CN208890679U (en) * | 2018-11-28 | 2019-05-21 | 燕山大学 | A kind of phase shift type three-phase high frequency chain matrix inverter topological structure |
Family Cites Families (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US9118259B2 (en) * | 2007-12-03 | 2015-08-25 | Texas Instruments Incorporated | Phase-shifted dual-bridge DC/DC converter with wide-range ZVS and zero circulating current |
| US20140119088A1 (en) * | 2012-10-29 | 2014-05-01 | Huawei Technologies Co., Ltd. | Three-level inverter and power supply equipment |
-
2018
- 2018-11-28 CN CN201811433103.1A patent/CN109412447B/en active Active
Patent Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2002204579A (en) * | 2001-01-09 | 2002-07-19 | Fuji Electric Co Ltd | Control method for inverter |
| CN103138614A (en) * | 2013-03-01 | 2013-06-05 | 南京航空航天大学 | Double-normal-shock inverter |
| WO2016150245A1 (en) * | 2015-03-23 | 2016-09-29 | 深圳市皓文电子有限公司 | Dc/dc converter |
| CN106787914A (en) * | 2017-03-03 | 2017-05-31 | 燕山大学 | LC series resonance-type three-phases high frequency chain matrix inverter topology and modulator approach |
| CN208890679U (en) * | 2018-11-28 | 2019-05-21 | 燕山大学 | A kind of phase shift type three-phase high frequency chain matrix inverter topological structure |
Non-Patent Citations (1)
| Title |
|---|
| 三相高频链矩阵式逆变器的HPWM调制策略原理与实现;闫朝阳;贾民立;张纯江;邬伟扬;;电工技术学报(11);全文 * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN109412447A (en) | 2019-03-01 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN106208769B (en) | Power conversion device | |
| CN107968471B (en) | LCLC resonance circuit, wide-range constant-power output direct-current charger and control method | |
| US9543823B2 (en) | Power conversion apparatus having a switching circuit unit that includes a switching device and an auxiliary switching device | |
| CN110504852B (en) | Single-phase soft switch charger topology with voltage decoupling function and modulation method thereof | |
| CN110768549B (en) | Single-phase zero-voltage soft switching charger topology and modulation method thereof | |
| CN103001523B (en) | Zero-voltage switching energy storage bridge-type inverter without additional voltage and modulation method for inverter | |
| CN108631604B (en) | Environment-friendly double-transformer type zero-current resonance three-level direct current converter | |
| CN108736756B (en) | Improved double-auxiliary resonant-pole three-phase soft switching inverter circuit | |
| CN109698627B (en) | Full-bridge DC/DC converter based on switched capacitor and modulation strategy thereof | |
| CN102969925B (en) | Without auxiliary voltage zero voltage switch energy storage semi-bridge type inverter and modulator approach | |
| CN102983767B (en) | Low auxiliary voltage zero voltage switch Stockbridge vibration inverter and modulator approach | |
| CN110572063A (en) | Asymmetric input multi-level converter and control method | |
| CN108667301B (en) | Full-bridge converter with follow current path | |
| CN212367151U (en) | Inverter circuit | |
| CN109412447B (en) | Phase-shifting three-phase high-frequency chain matrix inverter topological structure and modulation method | |
| CN117175951A (en) | LCL resonant current source DC-DC converter topological structure | |
| CN109742957B (en) | Double-ring full-resonance type soft switching converter | |
| CN113630032B (en) | Soft switching three-phase current type high-frequency chain matrix inverter topology and modulation method | |
| CN102843054A (en) | Single-phase five-level inverter | |
| CN111555633B (en) | Direct-current transformer for new energy direct-current grid connection and control method thereof | |
| CN208890679U (en) | A kind of phase shift type three-phase high frequency chain matrix inverter topological structure | |
| CN111342690B (en) | Modulation method of split capacitor power unit multilevel converter | |
| CN110943641B (en) | Pulse width modulation method of current type three-phase high-frequency link matrix inverter | |
| CN210724578U (en) | Rectangular wave crude oil dehydration power supply system | |
| CN115296542B (en) | Modulation strategy of active clamping three-tube push-pull full-bridge bidirectional DC-DC converter |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |