CN109347562A - A kind of CO-OFDM system phase noise optimization compensation method - Google Patents

A kind of CO-OFDM system phase noise optimization compensation method Download PDF

Info

Publication number
CN109347562A
CN109347562A CN201811167870.2A CN201811167870A CN109347562A CN 109347562 A CN109347562 A CN 109347562A CN 201811167870 A CN201811167870 A CN 201811167870A CN 109347562 A CN109347562 A CN 109347562A
Authority
CN
China
Prior art keywords
phase noise
symbol
ofdm
matrix
channel
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
CN201811167870.2A
Other languages
Chinese (zh)
Other versions
CN109347562B (en
Inventor
田飞燕
曹婧
毛怡帆
任宏亮
卢瑾
覃亚丽
乐孜纯
胡卫生
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Zhejiang University of Technology ZJUT
Original Assignee
Zhejiang University of Technology ZJUT
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Zhejiang University of Technology ZJUT filed Critical Zhejiang University of Technology ZJUT
Priority to CN201811167870.2A priority Critical patent/CN109347562B/en
Publication of CN109347562A publication Critical patent/CN109347562A/en
Application granted granted Critical
Publication of CN109347562B publication Critical patent/CN109347562B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B10/00—Transmission systems employing electromagnetic waves other than radio-waves, e.g. infrared, visible or ultraviolet light, or employing corpuscular radiation, e.g. quantum communication
    • H04B10/25—Arrangements specific to fibre transmission
    • H04B10/2507—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion
    • H04B10/2513—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion due to chromatic dispersion
    • H04B10/2525—Arrangements specific to fibre transmission for the reduction or elimination of distortion or dispersion due to chromatic dispersion using dispersion-compensating fibres
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/024—Channel estimation channel estimation algorithms
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03159—Arrangements for removing intersymbol interference operating in the frequency domain

Abstract

A kind of CO-OFDM system phase noise optimization compensation method, by the constraint condition for reducing phase noise spectrum structure, whether its channel estimation is accurately or not, its optimization algorithm can residual Amplitude Modulation Noise Limitation after effective compensation channel equalization, therefore final phase noise estimated accuracy is substantially better than other phase noise algorithm for estimating, and achieves breakthrough phase noise compensation effect.Phase noise dimensionality reduction model is used in the present invention, the phase noise optimization method computation complexity can be tolerated completely for CO-OFDM system.The present invention invention can be greatly facilitated application of the CO-OFDM system in long range access net and Metropolitan Area Network (MAN).

Description

A kind of CO-OFDM system phase noise optimization compensation method
Technical field
The invention belongs to optical communication network technology field, in particular to the phase noise of a kind of CO-OFDM system, which optimizes, to be mended Compensation method.
Background technique
Electrical domain orthogonal frequency division multiplexing (OFDM) modulation technique is combined in coherent optical communication system, forms coherent light orthogonal frequency Multiplexing (Coherent Optical Orthogonal Frequency Division Multiplexing, CO-OFDM) is divided to pass Transferring technology, with there is good inhibiting effect to fibre-optical dispersion and polarization mode dispersion, neatly compensated with Digital Signal Processing The advantages that ability of system injury, high spectrum utilization, it has also become the fields such as long range high-speed communication system and optical access network are standby One of concerned technology.
CO-OFDM system structure is as shown in Figure 1, can be divided into 5 modules: CO-OFDM system transmitting terminal mould by its function Block 101, optical modulator module 102, optical fiber transmission module 103, Photoelectric Detection module 104 and CO-OFDM system receiving terminal module Up-conversion of the electrical domain signal that 105, CO-OFDM transmitting end modules generate Jing Guo Electro-optical Modulation becomes the CO-OFDM signal of area of light, CO-OFDM signal transmits through optical fiber, after balanced detector through photoelectric conversion at the signal of electrical domain, CO-OFDM is docked receiving end again The electric signal received carries out signal processing to restore original transmission segment data.In conjunction with Fig. 1, to the course of work of whole system It is stated in detail.The data 106 of CO-OFDM system serial input pass through serioparallel exchange module 107, become parallel N number According to；The signal after serioparallel exchange is subjected to digital modulation 108 according to different modulation formats；Inverse fast Fourier transform IFFT mould Block 109 realizes conversion of the signal from frequency domain to time domain；Cyclic prefix CP 110 is added；Obtained electrical domain signal is carried out and is gone here and there to turn Change 111.The in-phase component and orthogonal component signal of above-mentioned signal pass through digital analog converter 112,113 respectively and are transformed to analog signal And pass through low-pass filter 114,115；The in-phase component 116 of signal and quadrature component 117 are amplified and injected using amplifier Realize in-phase component I and quadrature component Q to the orthogonal modulation of optical signal into I/Q modulator；I/Q modulator is by 3 both arms Mach increases Dare MZM modulator 120,121 and 122 and forms, and two of them modulator realizes the modulation to signal, third modulation Device 122 controls the phase difference of the in-phase component I and quadrature component Q of light modulation；The direct current of two modulators 120,121 is adjusted respectively Biasing guarantees to realize that the modulator of signal modulation works in minimum power point, and the modulator work of third control phase difference exists There are 90 ° of phase differences to guarantee two paths of signals for orthogonal points；118 indicate the emitting laser of CO-OFDM system, pass through splitter 119 are divided into the same laser of two beams, for driving two optical modulators 120 and 121.The signal of two optical modulators output passes through Bundling device 123 becomes the optical signal of single channel, is then inputted into fiber channel and is transmitted.The CO-OFDM signal of generation is in optical fiber In 124 after the transmission of long-distance, by direct light -125 compensated optical fiber of image intensifer-erbium-doped fiber amplifier (EDFA) It is transmitted again after loss, indicates the optical fiber of long range, 126 indicate optical band pass filter.After the transmission of the optical fiber of long-distance, Area of light signal is transformed into the signal of electrical domain by Photoelectric Detection module.127 indicate the local laser of CO-OFDM system receiving terminal, It is divided into the same laser of two beams by splitter, 128 indicate one 90 ° of phase-shifter；129 and 130 indicate two couplers, drive Move 4 photodiodes (PD) 131,132,133 and 134.135 and 136 indicate two subtracters, respectively correspond output and receive letter Number in-phase component I and quadrature component Q.Obtained in-phase component I and quadrature component Q passes through low-pass filter 137,138 and mould Number converter 139,140 enters the receiving end CO-OFDM after converting.The receiving end CO-OFDM carries out Digital Signal Processing 141, carries out The inverse process of CO-OFDM transmitting terminal carries out serioparallel exchange 142, removes cyclic prefix CP 143, then carries out FFT transform 144, right CO-OFDM signal carries out digital demodulation 145, and it is defeated finally to obtain original transmitting terminal serial data by the recovery of parallel-serial conversion 146 Out 147.
As multicarrier transmission systems, CO-OFDM (such as coherent light quadrature amplitude compared with single carrier coherent optical communication system (QAM) system of modulation), since its symbol period is long, subcarrier spacing is small, to laser phase noise and optical fiber it is non-linear more Sensitivity, so the development of CO-OFDM transmission technology is highly dependent on the compensation of laser phase noise and nonlinear fiber damage Precision.Machine learning at present has been proposed for mitigating nonlinear influence in CO-OFDM system.Although in CO-OFDM system Middle nonlinear impairments and linear damage can be compensated simultaneously by nonlinear equalizers such as neural networks, but such method It is to use a large amount of training symbol and huge computation complexity as cost.In the present invention, we are dedicated in big laser High-precision phase noise compensation method is realized in the CO-OFDM system of line width and high order modulation, can be applied to some make With big line width, the Metro access networks of low-cost laser.
The digital signal processing algorithm of phase noise compensation is used in many of CO-OFDM system.Traditional algorithm can To be divided into three types: blind phase noise compensation algorithm (document 1, Son T.Le, Paul A.Haigh, Andrew D.Ellis, Sergei K.Turitsyn.Blind phase noise compensation for CO-OFDM Transmission.Journal of Lightwave Technology, 2016,34 (2): 745-753., that is, Son T.Le, The blind phase noise of Paul A.Haigh, Andrew D.Ellis, Sergei K.Turitsyn.CO-OFDM Transmission system is mended It repays, lightwave technology journal, 2016,34 (2): 745-753.)；Decision feedback algorithms (document 2, Xuezhi Hong, Xiaojian Hong,Sailing He.Linearly interpolated sub-symbol optical phase noise Suppression in CO-OFDM system.Optics Express, 2015,23 (4): 4691-4702., that is, Xuezhi Linear interpolation and subsymbol light phase noise suppressed, light in Hong, Xiaojian Hong, Sailing He, CO-OFDM system Learn flash report, 2015,23 (4): 4691-4702.)；Algorithm based on pilot tone, including being led with the pilot configuration and radio frequency of dispersion Frequently.When CO-OFDM system uses big line width laser and high-order modulating, the pilot sub-carrier using dispersion is that reality is answered The most general and practical phase noise estimation method in.In numerous algorithms based on pilot tone, linear interpolation and subsymbol light Phase noise compensation (LI-SCPEC) be one of most representative algorithm (document 2, Xuezhi Hong, Xiaojian Hong, Sailing He.Linearly interpolated sub-symbol optical phase noise suppression In CO-OFDM system.Optics Express, 2015,23 (4): 4691-4702., that is, Xuezhi Hong, Xiaojian Linear interpolation and subsymbol light phase noise suppressed in Hong, Sailing He, CO-OFDM system, optics letter, 2015,23 (4): 4691-4702.), which first uses Least Square Method common phase noise (common phase error, CPE):
φn=angle (Yn/Xn), i=0,1 ..., Ns
Wherein Xn、YnIt sent for n-th, receive symbol.
Linear interpolation obtains the phase noise on each subcarrier, N againcpFor the length of cyclic prefix:
Then carry out phase correction.By the signal of channel equalization and phase correction are as follows:
Here ()*Indicate complex-conjugate matrix,Indicate that associate matrix, F are FFT matrix, phase noise matrix
Operation Q () is made decisions to above-mentioned processed signal again, and is IFFT and transforms to time domain:
Symbol after signaling channel is balancedBy ai、biIt is divided into NBA subsymbol, then i-th subsymbol can be with It respectively indicates are as follows:
bn=[bn(il+0),bn(il+1),...,bn(il+l-1)]T
an=[an(il+0),an(il+1),...,an(il+l-1)]T
Here i ∈ [0, NB- 1], l=N/NBIndicate the length of each subsymbol.
The then common phase noise of n-th of OFDM symbol, i-th of subsymbolAre as follows:
Phase noise matrix at this timeThis method (linearly interpolated sub-symbol optical PNC scheme, LI-SCPEC) passes through above step reality Existing phase noise estimation, but its phase noise compensation performance is big and when order of modulation is high is ineffective in system laser line width.
Recently, there is scholar in Wireless OFDM System, phase noise spectrum structure and dimensionality reduction models coupling, by phase noise Estimation problem proposes there is very big breakthrough (document 3, Pramod in phase noise estimated accuracy as optimization problem Mathecken,Taneli Riihonen,Stefan Werner,Risto Wichman.Phase noise estimation in OFDM:utilizing its associated spectral geometry.IEEE Transactions on Signal Processing, 2016,64 (8): 1999-2012., that is, Pramod Mathecken, Taneli Riihonen, The phase noise compensation in ofdm system: Stefan Werner, Risto Wichman utilizes phase noise spectrum structure, IEEE Signal processing transactions, 2016,64 (8): 1999-2012. document 4, Pramod Mathecken, Taneli Riihonen, Stefan Werner,Risto Wichman.Constrained phase noise estimation in OFDM using scattered pilots without decision feedback.IEEE Transactions on Signal Processing, 2017,65 (9): 2348-2362., that is, Pramod Mathecken, Taneli Riihonen, Stefan Werner, Risto Wichman utilize scattered pilot and the phase noise of with constraint conditions without decision-feedback in ofdm system Estimation, IEEE signal processing transactions, 2017,65 (9): 2348-2362.).Document 3 is by phase noise spectrum architectural characteristic (phase noise spectral geometry, PSNG) is described in detail, i.e., the phase noise of each OFDM symbol Compose vector δnMeet:
ΛlIt is Kronecker delta function (Λ0=1, Λl=0, l=1,2 ..., Nc- 1), Pl=(P1)lFor displacement Matrix is by Nc×NcThe P of dimension1It constitutes, P1First row be Nc× 1 vector [0,1,0 ..., 0]T, jth is arranged to be recycled downwards by it It shifts j-1 times and obtains.When l=0, P0For Nc×NcUnit matrix.It can be by N using dimensionality reduction modelcTie up phase noise spectrum vector δn It is down to N-dimensional γn, therefore PNSG becomes corresponding N-dimensional equation:The removable writing two of the equation A constraint condition:WithDocument 4 describes Two kinds of phase noise optimization algorithms, a kind of ULS estimation to constrain without PNSG, another kind is with two constraint condition of PNSG GLS estimation.It is K × N that matrix K is defined in the present inventioncMatrix, each column are all unit vectors, and the position of element 1, which corresponds to, to be sent out The position of sending end record pilot tone, therefore pilot symbol transmitted are as follows:
Xn,p=KXn
HereIt is transmitting terminal frequency domain symbol.Therefore the estimation of receiving end frequency pilot sign Value isChannel frequency response matrix HnIt is that diagonal element isDiagonal matrix, matrix ΥnIt is by column Circular matrix, its first row are the reception data of corresponding symbolSubsequent jth column use first row Downward cyclic shift j-1 times,It is δnEstimated value.
The then phase noise estimate cost function of above-mentioned ULS algorithm are as follows:
Wherein It is γnEstimated value.
The cost function of GLS algorithm is then
(P):
Wherein
It is inspired by this phase noise optimization method based on pilot tone, the present invention will propose that a kind of phase based on pilot tone is made an uproar Sound Optimization Compensation algorithm is used for high-precision phase position noise compensation of the CO-OFDM system in big laser linewidth and high order modulation. In CO-OFDM system, phase noise Optimization Compensation algorithm needs to carry out its channel estimation and equilibrium, therefore channel estimation is general Using traditional least-squares estimation (least square channel equalization, LS-CE).That is:
WhereinAs nth symbol, the channel frequency response of k-th of subcarrier, Xn,k, Yn,kRespectively send and Received nth symbol, the data on k-th of subcarrier, V 'n,kFor noise item.In LS algorithm for estimating, noise item is not considered It influences.If wanting to improve precision of channel estimation, scalar Kalman filtering is added and reduces noise and inter-carrier interference The influence of (intercarrier interference, ICI) can slightly improve algorithm performance.It is embodied as follows:
Initial rough channel estimation valueOutput by Kalman filtering at each subcarrier, after smoothing processing It is defined asHere k indicates that subcarrier, n indicate initial training symbol.Filter equation writingWherein:
Status predication equation
Least mean-square error (MSE) predictive equation
Kalman gain equation
K (n)=P (n | n-1)/(1+P (n | n-1))
State renewal equation
Least mean-square error renewal equation
P (n | n)=(1-K (n)) P (n | n-1)
Here Kgain, KaAnd KbIt is all constant.
Summary of the invention
It is a primary object of the present invention to overcome existing algorithm phase noise in big line width, long range CO-OFDM system Estimate the not high problem for causing phase noise compensation effect poor of accuracy, proposes a kind of phase noise Optimization Compensation method (modified geometry constrained least square estimation, M-GLS) is used for big line width CO- The phase noise high-accuracy compensation of ofdm system.
In order to solve the above technical problem, the present invention provides the following technical solutions:
A kind of CO-OFDM system phase noise optimization compensation method, comprising the following steps:
(1) receiving end initial signal is handled, and process is as follows:
1-1, receiving end carry out coherent detection reception to the CO-OFDM signal received, then carry out analog-to-digital conversion, obtain The signal of electrical domain；Electrical domain optical fiber dispersion compensation is then carried out, by the analytical form of fiber channel frequency domain transfer function through Fourier Time domain is transformed to, designs the long unit impulse response of time-domain finite (FIR) filter to realize, the order of the filter is tired with dispersion It accumulates and increases；
1-2, cyclic prefix CP is removed；
1-3, signal is become from time domain by frequency domain using Fast Fourier Transform (FFT) (FFT)；
(2) LS channel estimation (LS-CE) with it is balanced, process is as follows:
2-1, LS rough channel estimation.In transmitting terminal, N before being arrangedpA symbol is training symbol, in receiving end, according to LS :
WhereinAs nth symbol, the channel frequency response of k-th of subcarrier, Xn,k, Yn,kRespectively send and Received nth symbol, the data on k-th of subcarrier, V 'n,kDo not consider in rough LS algorithm for estimating for noise item The influence of noise item；
2-2, reduce the interference between adjacent sub-carriers with the average method of frequency domain in symbol (ISFA), obtain k-th of son The accurate valuation of the channel transfer functions of carrier wave
Here t is the adjacent sub-carrier number of channel for participating in channel estimation；
2-3, channel equalization, in each OFDM frame, to the N of receiving endsAfter a OFDM data symbol carries out channel equalization, N-th of OFDM symbol, k-th of frequency domain data Y 'n,kFor,
(3) M-GLS phase noise is estimated: the constraint condition by reducing phase noise spectrum structure, regardless of its channel estimation Whether accurate, optimization algorithm can residual Amplitude Modulation Noise Limitation after effective compensation channel equalization, pass through construction cost function and choosing Constraint condition is selected, dual problem is write out and finally finds out phase noise fine estimation；
(4) final phase noise compensation: carrying out following phase noise compensation to its frequency-region signal,
Wherein Y 'nAs nth symbol of the receiving end after channel equalization,Indicate the phase noise spectrum of nth symbol Vector estimated value.
Further, the step (3) the following steps are included:
3-1, phase noise spectrum vector dimensionality reduction, it is assumed that in CO-OFDM system, at digital signal of the receiving end needed for other Adjustment method is all without error, then n-th of OFDM symbol of received time domain, m-th of sampled point yn,mIt indicates are as follows:
Here xn,m, hn,mAnd vn,mRespectively time domain sends signal, channel impulse response and noise item ' '
The phase noise of obvious each sampled pointAll meet:
It enablesIt isDiscrete Fourier transform (DFT), the present invention in It is referred to as phase noise spectrum vector, NcThe number of subcarriers ' ' for including for each OFDM symbol
Since phase noise can be considered as low-pass signal in CO-OFDM system, estimation is only needed to make an uproar positioned at phase Sound spectrum vectorTop and bottom element, that is, low-frequency component.Therefore make phase noise by realizing Architectural characteristic keeps transformation (PNSG preserving transformations, PPT) matrix, and dimensionality reduction can be realized, make δn's Dimension is by NcIt is reduced to N, thus computation complexity also substantially reduces.
The architectural characteristic PNSG of phase noise frequency domain is as follows:
HereIt is column vector, ΛlIt is Kronecker delta function (Λ0=1, Λl= 0, l=1,2 ..., Nc- 1), Pl=(P1)lIt is permutation matrix by Nc×NcThe P of dimension1It constitutes, P1First row be Nc× 1 vector [0,1,0,...,0]T, jth is arranged to be obtained by its downward cyclic shift j-1 times.When l=0, P0For Nc×NcUnit matrix.In formulaIndicate conjugate transposition.T is NcThe PPT matrix of × N realizes the dimensionality reduction of phase noise spectrum vector, meets δn=T γn.After dimensionality reduction Phase noise spectrum by N-dimensional vector γnIt indicates, meets N-dimensional spectrum structure characteristic:
HereIt is P respectivelyl, ΛlCorresponding N-dimensional matrix.
Above-mentioned characteristic includes l=0 and l ≠ 0 two condition:
HereIt is respectivelyReal and imaginary parts；
3-2, the cost function for constructing M-GLS, definition matrix K are K × NcMatrix, by j unit vectorj∈{1,2,...,NcObtain, therefore pilot symbol transmitted are as follows:
Xn,p=KXn
HereIt is transmitting terminal frequency domain symbol.According to system model it is found that receiving end is led The estimated value of frequency symbol is thenChannel frequency response matrix HnIt is that diagonal element isDiagonal matrix, Matrix ΥnIt is by column circular matrix, its first row is the reception data of corresponding symbolThen Jth is arranged with first row downward cyclic shift j-1 times,For δnEstimated value；
Then the cost function of phase noise estimation is defined as:
Wherein It is γnEstimated value, T is NcThe PPT of × N Matrix realizes the dimensionality reduction of phase noise spectrum vector, meets δn=T γn；
The algorithm and its constraint condition of 3-3, M-GLS
Then the optimization problem of M-GLS is to have the form of PNSG partially restrained condition below:
(P):
HerePay attention to constraint condition when constraint condition only takes；
3-4, dual problem is write out, writes out corresponding dual problem using S-procedure:
(D):Maximizeτ
Here τ, αl, βlIt is all optimized variable；
3-5, its optimal solution is sought are as follows:
Wherein ()+It is defined as pseudo inverse matrix, αl ◇,βl ◇It is the minimum value obtained by solution (D).
Technical concept of the invention are as follows: phase noise Optimization Compensation method is inserted into several training in each OFDM frame of transmitting terminal Symbol is inserted into several pilot sub-carriers as expense at regular intervals in each OFDM data symbol.Phase noise optimization Compensation method carries out the behaviour of cyclic prefix, FFT, dispersion compensation, Kalman filtering channel estimation and equalization receiving end data After work, dimensionality reduction, greatly reduction computation complexity are realized first with the low-pass characteristic of phase noise spectrum vector.Later according to hair The smallest principle of minimum mean-square error between received frequency pilot sign is sent, the cost function of phase noise optimal estimating is write out.This When, using the partial condition of phase noise architectural characteristic as constraint, S-procedure is recycled to write out the antithesis of the optimization problem Problem acquires optimal solution with the tool box CVX in MATLAB.This optimal value is finally subjected to phase noise compensation.This method compared with Pilot tone algorithm LI-SCPEC and other optimization algorithms are compared, and preferable compensation effect is achieved.It is in 40Gb/s, transmission range Algorithm of the invention is verified in the CO-OFDM system of 400km, modulation system is 16 rank quadrature amplitude modulations of circle (C-16QAM), when joint laser linewidth is 1.8MHz, bit error rate performance just reaches the hard-decision forward error correction upper limit (HD- FEC), which can be greatly facilitated application of the CO-OFDM system in long range access net and Metropolitan Area Network (MAN).
Compared with the prior art, the invention has the following advantages and beneficial effects:
1. the CO-OFDM system of pair high-order digit modulation and big line width laser, the amendment of phase noise estimation of the present invention Method obtains preferable PNC effect, such as M-GLS-ICE to C-16QAM, 400km, and the joint laser linewidth upper limit is reachable 1.8MHz.Pilot tone subcarrier spacing used in the present invention is larger, uses less pilot sub-carrier than intra-class correlation algorithm, compared with The availability of frequency spectrum of system is improved greatly.
2. M-GLS proposed by the present invention is compared with traditional PNC method, although its complexity is increased slightly, as generation What valence obtained is significantly improving for bit error rate performance, in the CO-OFDM system of big line width laser, high order modulation and long distance transmission In system, this method compensation effect tool has an enormous advantage, such as the M-GLS- to 16QAM, 0.6MHz joint laser linewidth ICE, Networks of Fiber Communications of the transmission range up to 600km, suitable for practical middle and long distance.
Detailed description of the invention
Fig. 1 is the schematic diagram of CO-OFDM system in the prior art.
Fig. 2 is the schematic diagram of phase noise Optimization Compensation algorithm M-GLS proposed by the present invention.
Fig. 3 be in the embodiment of the present invention 1 C-16QAM, transmission range be 0 (back-to-back system (CCC-0)), channel estimation using pass When least-squares estimation (LS-CE) of system, ULS (the not optimization algorithm of belt restraining), GLS are (with the excellent of whole PNSG constraint condition Change algorithm), the bit error rate performance of M-GLS, LI-SCPEC with joint laser linewidth variation when relation curve.
Fig. 4 be in the embodiment of the present invention 1 C-16QAM, transmission range 400km, joint laser linewidth be 0.6MHz When, M-GLS-ICE algorithm bit error rate performance is with covariance initial value in Kalman filtering and parameter KbThree-dimensional curve when variation.
Fig. 5 be in the embodiment of the present invention 1 when C-16QAM, transmission range 0, LS-CE and ICE being respectively adopted, ULS, The relation curve when bit error rate performance of GLS, M-GLS, LI-SCPEC are with joint laser linewidth variation.
Fig. 6 be in the embodiment of the present invention 1 when C-16QAM, transmission range 400km, LS-CE and ICE being respectively adopted, The relation curve when bit error rate performance of ULS, GLS, M-GLS, LI-SCPEC are with joint laser linewidth variation.
Fig. 7 be in the embodiment of the present invention 1 C-16QAM, transmission range 400km, joint laser linewidth be 1.4MHz When, planisphere that tetra- kinds of phase noise compensation algorithms of ULS-ICE, GLS-ICE, M-GLS-ICE, LI-SCPEC-ICE obtain.
Fig. 8 be in the embodiment of the present invention 1 C-16QAM, joint laser linewidth be 0.6MHz, optical signal to noise ratio 32dB, Relationship when LS-CE and ICE is respectively adopted, when the bit error rate performance of ULS, GLS, M-GLS, LI-SCPEC change with transmission range Curve, while also showing joint laser linewidth is 0MHz, that is, does not need what bit error rate performance when PNC changed with transmission range Curve, this is the upper limit of theoretically algorithm performance.
Fig. 9 be in the embodiment of the present invention 1 C-16QAM, transmission range 400km, joint laser linewidth be 0.6MHz, When LS-CE and ICE is respectively adopted, when the bit error rate performance of ULS, GLS, M-GLS, LI-SCPEC change with optical signal to noise ratio (OSNR) Relation curve, while also show joint laser linewidth be 0MHz, that is, bit error rate performance changes with OSNR when not needing PNC Curve, this for theoretically algorithm performance the upper limit.
Specific embodiment
The present invention is described in further detail below with reference to examples and drawings, but embodiments of the present invention are unlimited In this.
Referring to Fig. 2~Fig. 9, a kind of CO-OFDM system phase noise optimization compensation method relates generally to coherent light orthogonal frequency The signal processing problems for dividing multiplexing CO-OFDM system receiving terminal, with reference to retouching in detail to CO-OFDM system structure in background technique It states.
As shown in Figure 1, CO-OFDM system includes CO-OFDM system transmitting end module 101, CO-OFDM optical modulator module 102, optical fiber transmission module 103, Photoelectric Detection module 104 and CO-OFDM system receiving terminal module 105, system transmitting terminal produce The up-conversion that raw signal have passed through light modulation becomes the CO-OFDM signal of area of light, and CO-OFDM signal is transmitted through optical fiber, balanced Through photoelectric conversion at the signal of electrical domain after detector, system receiving terminal again to the electrical domain signal received carry out signal processing to Restore original transmission end data.Initial 40Gb/s pseudo noise code binary data stream is mapped to high-order QAM modulation (16QAM) On 512 subcarriers, the points of FFT or IFFT are 1024.Cyclic prefix CP length in each OFDM data symbol is 128 Point.Every 50km single mode optical fiber is followed by an erbium-doped optical fiber amplifier EDFA, which is 10dB, noise coefficient 4dB. Entire optical fiber link shares 8 sections of 50km single mode optical fibers and amplifier EDFA is added to constitute (when transmission range is 400km).The single mode optical fiber Abbe number be 17ps/nmkm, chromatic dispersion gradient be 0.075ps/ (nm2Km), nonlinear factor 1.3W-1·km-1, PMD coefficient isLoss factor is 0.2dB/km.C- first is carried out to binary system pseudo noise code before OFDM modulation 16QAM mapping.Transmitting terminal laser and coherent reception end laser line width having the same and wavelength, wavelength 1550nm. Laser optimum transmission power is 0dBm.The bandwidth of optical band pass filter is 40GHz.By each OFDM frame 4 OFDM symbols first Number it is training symbol, is divided into 16 between each OFDM symbol pilot frequency sequence.Each OFDM symbol divides sub- symbol in LI-SCPEC method Number mesh is NB=4.
Below with reference to Fig. 2, a kind of CO-OFDM system phase noise optimization compensation method step of the invention is carried out detailed Explanation, comprising the following steps:
S201: the processing of receiving end initial signal, process are as follows:
S201-1, receiving end carry out coherent detection reception to the CO-OFDM signal received, then carry out analog-to-digital conversion, Obtain the signal of electrical domain.Electrical domain optical fiber dispersion compensation is then carried out, by the analytical form of fiber channel frequency domain transfer function through Fu Vertical leaf transformation designs the long unit impulse response of time-domain finite (FIR) filter to time domain to realize, the order of the filter is with color It dissipates accumulation and increases.
S201-2, cyclic prefix CP is removed.
S201-3, signal is become from time domain by frequency domain using Fast Fourier Transform (FFT) (FFT).
S202: LS channel estimation (LS-CE).It is assumed that an OFDM frame includes N in time domains=128 OFDM symbols Number, preceding Np=4 are training symbol, and each OFDM symbol includes N in frequency domainc=512 subcarriers, process are as follows:
S202-1, LS rough channel estimation.In transmitting terminal, N before being arrangedpA symbol is training symbol, in receiving end, according to LS is obtained:
WhereinAs nth symbol, the channel frequency response of k-th of subcarrier, Xn,kFor n-th of symbol of transmission Number, the data on k-th of subcarrier, V 'n,kFor noise item.In rough LS algorithm for estimating, the influence of noise item is not considered.
S202-2, reduce the interference between adjacent sub-carriers with the average method of frequency domain in symbol (ISFA), obtain k-th The accurate valuation of the channel transfer functions of subcarrier
Here t is the adjacent sub-carrier number of channel for participating in channel estimation.
S203, channel equalization.In each OFDM frame, to the N of receiving endsA OFDM data symbol carries out channel equalization Afterwards, n-th of OFDM symbol, k-th of frequency domain data Y 'n,kFor,
S204:M-GLS phase noise optimal estimating and compensation, process are as follows:
S204-1, phase noise spectrum vector dimensionality reduction.Assuming that in CO-OFDM system, in number letter of the receiving end needed for other Number Processing Algorithm is all without error, then n-th of OFDM symbol of received time domain, m-th of sampled point yn,mIt can indicate are as follows:
Here xn,m, hn,mAnd vn,mRespectively time domain sends signal, channel impulse response and noise item.
The phase noise of obvious each sampled pointAll meet:
It enablesIt isDiscrete Fourier transform (DFT), the present invention in It is referred to as phase noise spectrum vector, NcThe number of subcarriers for including for each OFDM symbol.
Since phase noise can be considered as low-pass signal in CO-OFDM system, estimation is only needed to make an uproar positioned at phase Sound spectrum vectorTop and bottom element, that is, low-frequency component.Therefore make phase noise by realizing Architectural characteristic keeps transformation (PNSG preserving transformations, PPT) matrix, and dimensionality reduction can be realized, make δn's Dimension is by NcIt is reduced to N, thus computation complexity also substantially reduces.
The architectural characteristic PNSG of phase noise frequency domain is as follows:
HereIt is column vector, ΛlIt is Kronecker delta function (Λ0=1, Λl= 0, l=1,2 ..., Nc- 1), Pl=(P1)lIt is permutation matrix by Nc×NcThe P of dimension1It constitutes, P1First row be Nc× 1 vector [0,1,0,...,0]T, jth is arranged to be obtained by its downward cyclic shift j-1 times.When l=0, P0For Nc×NcUnit matrix.In formulaIndicate conjugate transposition.T is NcThe PPT matrix of × N realizes the dimensionality reduction of phase noise spectrum vector, meets δn=T γn.After dimensionality reduction Phase noise spectrum by N-dimensional vector γnIt indicates, meets N-dimensional spectrum structure characteristic:
HereIt is P respectivelyl, ΛlCorresponding N-dimensional matrix.
Above-mentioned characteristic includes l=0 and l ≠ 0 two condition:
HereIt is respectivelyReal and imaginary parts.
Here PPT takes following form:
Wherein F is Nc×NcDimension,For N × N-dimensional DFT matrix.It isComplete 1 vector of dimension, 0 is null vector, i.e., There is δn=T γn。
S204-2, M-GLS cost function.Definition matrix K is K × NcMatrix, by j unit vectorj∈{1,2,...,NcObtain, therefore pilot symbol transmitted are as follows:
XN, p=KXn
HereIt is transmitting terminal frequency domain symbol.Therefore the estimation of receiving end frequency pilot sign Value isChannel frequency response matrix HnIt is that diagonal element isDiagonal matrix, matrix ΥnIt is by column Circular matrix, its first row are the reception data of corresponding symbolSubsequent jth is arranged with first Arrange downward cyclic shift j-1 times,It is δnEstimated value.
Then the cost function of phase noise estimation is defined as:
Wherein It is γnEstimated value, T be PPT matrix.
S204-3, M-GLS constraint condition.
The optimization problem of M-GLS is to have the form of PNSG partially restrained condition below:
(P):
HerePay attention to constraint condition when constraint condition only takes l ≠ 0.
S204-4, dual problem.Its dual problem is write out using S-Procedure, as follows:
(D):Maximizeτ
Here τ, αl, βlIt is all optimized variable.
S204-5, its optimal solution is sought are as follows:
Wherein ()+It is defined as pseudo inverse matrix, αl ◇,βl ◇It is the minimum value obtained by solution (D).
S205: carrying out following phase noise compensation to its frequency-region signal,
Wherein Y 'nAs nth symbol of the receiving end after channel equalization,Indicate the phase noise spectrum of nth symbol Vector estimated value.
Simulation numerical verifying is carried out to the phase noise Optimization Compensation method that the invention proposes.
Fig. 3 show C-16QAM, transmission range be 0 (back-to-back system (CCC-0)), channel estimation using LS-CE when, ULS, The relation curve when bit error rate performance of GLS, M-GLS, LI-SCPEC are with joint laser linewidth variation.Pilot interval is at this time 16, dimensionality reduction Dimensionality optimization is N=16.As seen from the figure, when joint laser linewidth is in 0.8MHz~3.2MHz, M-GLS's is excellent More property is very clear, and in 3.2MHz, M-GLS just reaches the hard-decision forward error correction upper limit (HD-FEC), i.e. BER=3.8e-03, And in 0.8MHz or less, 0 is shown as with the bit error rate that direct comparison error bit method calculates M-GLS.And have whole PNSG Its performance is much worse than ULS and M-GLS during line width constantly increases by the GLS of constraint, and the bigger difference of line width is more, The error correction upper limit is reached when 1.4MHz.
Fig. 4 is shown in Kalman filtering channel estimation (ICE), in fixed KgainAnd KaWhen all taking 1, covariance initial value P (0 | 0) and KbInfluence of the value of two parameters to error performance.Z axis is 1/BER, the corresponding parameter in the highest point of three-dimensional curve diagram Value is as optimal.
When laser optimum transmission power 0dBm, Figures 5 and 6 show that in C-16QAM, transmission range be 0 (system back-to-back System) and 400km, when LS-CE and ICE is respectively adopted, the bit error rate performance of ULS, GLS, M-GLS, LI-SCPEC is with combining laser Relation curve when device line width variation.As seen from the figure, transmission range is the rule substantially one that 0 or 400km curve shows It causes, the case where explanation of Fig. 3 had analyzed back-to-back system (CCC-0), here the long distance transmission of Main Analysis 400km.Same M-GLS- The performance of ICE is better than other all algorithms, just reaches the HD-FEC upper limit when combining laser linewidth and being 1.8MHz.For all Phase noise compensation algorithm, obtained obviously with the performance that ICE ratio LS-CE can obtain preferably performance, especially GLS Improvement, this is illustrated in CO-OFDM system, and the GLS all constrained with PNSG is very sensitive to the precision of channel estimation. For ULS, M-GLS and LI-SCPEC, there is slight improvement compared to original LS-CE with the performance after ICE, indicate that they are right The precision of channel estimation is not especially sensitive.
It is 0dBm in laser optimum transmission power that Fig. 7, which is shown, and transmission range is 400km, combines laser linewidth When for 1.4MHz, what tetra- kinds of phase noise compensation algorithms of ULS-ICE, GLS-ICE, M-GLS-ICE, LI-SCPEC-ICE obtained is connect Receiving end C-16QAM planisphere.16QAM signaling point after the OFDM demodulation of receiving end is by laser phase noise and fibre-optical dispersion It influences, serious rotation and diverging has occurred.Limit for length's unit is had according to the design of the time domain specification of fiber channel first in receiving end Impulse response (FIR) filtering carries out electrical domain dispersion compensation, followed by phase noise compensation.The signaling point that Fig. 7 is shown is Equilibrium is 16 block number strong points, can be seen that whether ICI phase noise is effectively inhibited by the degree of divergence of planisphere, from And reduce the probability that mistake in judgment occurs in pre- judgement.The inhibitory effect of obvious M-GLS-ICE is best.
In order to explore M-GLS proposed by the present invention accessible upper limit in transmission range, Fig. 8 show C-16QAM, When joint laser linewidth is 0.6MHz, LS-CE and ICE is respectively adopted, the error code of ULS, GLS, M-GLS, LI-SCPEC are forthright Relation curve when can change with transmission range.With the increase of transmission range, the non-linear damage of optical fiber in CO-OFDM system Wound becomes more and more significant, but it can be seen from the figure that M-GLS-ICE can preferably inhibit the influence of nonlinear fiber, Transmit 300km than pilot tone algorithm LI-SCPEC-ICE when reaching the HD-FEC upper limit, the Metro access networks suitable for long range more.
Influence of the Fig. 9 to transmitting terminal optical signal to noise ratio (OSNR) to system performance is discussed, it is shown that C-16QAM, When transmission range is 400km, joint laser linewidth is 0.6MHz, LS-CE and ICE is respectively adopted, ULS, GLS, M-GLS, LI- The relation curve when bit error rate performance of SCPEC changes with OSNR.As seen from Figure 9, even if M-GLS proposed by the present invention exists Still it is better than other all algorithms in the case where low OSNR.
The dimensionality reduction dimension N=16 of optimization algorithm in the present invention, algorithm complexity are O (N4.5), therefore can hold in complexity In the case where bearing, the present invention provides a kind of suitable for big line width, long range, the CO-OFDM system of high order modulation, and PNC Traditional algorithm based on pilot tone algorithm can be far superior to.
The phase noise Optimization Compensation in coherent light orthogonal frequency division multiplexing CO-OFDM system of the present invention is calculated above Method has carried out introduction in detail, above example illustrate to be merely used to help understand method and its core concept of the invention and Non- to be limited, others are any to be made to change without departing from the spirit and principles of the present invention, modifies, substitutes, group It closes, simplify, should be equivalent substitute mode, be included within the scope of the present invention.

Claims (2)

1. a kind of CO-OFDM system phase noise optimization compensation method, which is characterized in that the described method comprises the following steps:
(1) receiving end initial signal is handled, and process is as follows:
1-1, receiving end carry out coherent detection reception to the CO-OFDM signal received, then carry out analog-to-digital conversion, obtain electrical domain Signal；Electrical domain optical fiber dispersion compensation is then carried out, by the analytical form of fiber channel frequency domain transfer function through Fourier transform To time domain, design the long unit impulse response of time-domain finite (FIR) filter to realize, the order of the filter accumulate with dispersion and Increase；
1-2, cyclic prefix CP is removed；
1-3, signal is become from time domain by frequency domain using Fast Fourier Transform (FFT)；
(2) LS channel estimation with it is balanced, process is as follows:
2-1, LS rough channel estimation, in transmitting terminal, N before being arrangedpA symbol is that training symbol is obtained in receiving end according to LS:
WhereinAs nth symbol, the channel frequency response of k-th of subcarrier, Xn,k, Yn,kRespectively send and receive Nth symbol, the data on k-th of subcarrier, V 'n,kFor noise item；
2-2, reduce the interference between adjacent sub-carriers with the average method of frequency domain in symbol, obtain the channel of k-th of subcarrier The accurate valuation of transfer function
Here t is the adjacent sub-carrier number of channel for participating in channel estimation；
2-3, channel equalization, in each OFDM frame, to the N of receiving endsAfter a OFDM data symbol carries out channel equalization, n-th K-th of frequency domain data Y ' of OFDM symboln,kFor,
(3) M-GLS phase noise is estimated: the constraint condition by reducing phase noise spectrum structure, regardless of its channel estimation is accurate Whether, optimization algorithm can residual Amplitude Modulation Noise Limitation after effective compensation channel equalization, about by construction cost function and selection Beam condition writes out dual problem and finally finds out phase noise fine estimation；
(4) final phase noise compensation: carrying out following phase noise compensation to its frequency-region signal,
Wherein Y 'nAs nth symbol of the receiving end after channel equalization,Indicate the phase noise spectrum vector of nth symbol Estimated value.
2. a kind of CO-OFDM system phase noise optimization compensation method as described in claim 1, which is characterized in that the step Suddenly (3) the following steps are included:
3-1, phase noise spectrum vector dimensionality reduction, it is assumed that in CO-OFDM system, in receiving end, the Digital Signal Processing needed for other is calculated Method is all without error, then n-th of OFDM symbol of received time domain, m-th of sampled point yn,mIt indicates are as follows:
Here xn,m, hn,mAnd vn,mRespectively time domain sends signal, channel impulse response and noise item；
The phase noise of obvious each sampled pointAll meet:
It enablesIt isDiscrete Fourier transform, it is referred to as phase noise spectrum Vector, NcThe number of subcarriers for including for each OFDM symbol；
Since phase noise can be considered as low-pass signal in CO-OFDM system, estimation is only needed to be located at phase noise spectrum VectorTop and bottom element, that is, low-frequency component；Therefore make phase noise structure by realizing Characteristic keeps transformation matrix, and dimensionality reduction can be realized, make δnDimension by NcIt is reduced to N；
The architectural characteristic PNSG of phase noise frequency domain is as follows:
HereIt is column vector, ΛlIt is Kronecker delta function (Λ0=1, Λl=0, l= 1,2,...,Nc- 1), Pl=(P1)lIt is permutation matrix by Nc×NcThe P of dimension1It constitutes, P1First row be Nc× 1 vector [0,1, 0,...,0]T, jth is arranged to be obtained by its downward cyclic shift j-1 times；When l=0, P0For Nc×NcUnit matrix, in formulaTable Show conjugate transposition；T is NcThe PPT matrix of × N realizes the dimensionality reduction of phase noise spectrum vector, meets δn=T γn, phase after dimensionality reduction Position noise spectrum is by N-dimensional vector γnIt indicates, meets N-dimensional spectrum structure characteristic:
HereIt is P respectivelyl, ΛlCorresponding N-dimensional matrix；
Above-mentioned characteristic includes l=0 and l ≠ 0 two condition:
HereIt is respectivelyReal and imaginary parts；
3-2, the cost function for constructing M-GLS, definition matrix K are K × NcMatrix, by j unit vectorj∈{1,2,...,NcObtain, therefore pilot symbol transmitted are as follows:
Xn,p=KXn
HereIt is transmitting terminal frequency domain symbol, according to system model it is found that receiving end pilot tone accords with Number estimated value be thenChannel frequency response matrix HnIt is that diagonal element isDiagonal matrix, square Battle array ΥnIt is by column circular matrix, its first row is the reception data of corresponding symbolSubsequent J is arranged with first row downward cyclic shift j-1 times,For δnEstimated value；
Then the cost function of phase noise estimation is defined as:
Wherein It is the estimated value of γ n, T is NcThe PPT square of × N Battle array realizes the dimensionality reduction of phase noise spectrum vector, meets δn=T γn；
The algorithm and its constraint condition of 3-3, M-GLS
Then the optimization problem of M-GLS is to have the form of PNSG partially restrained condition below:
HerePay attention to constraint condition when constraint condition only takes；
3-4, dual problem is write out, writes out corresponding dual problem using S-procedure:
(D):Maximizeτ
Here τ, αl, βlIt is all optimized variable；
3-5, its optimal solution is sought are as follows:
Wherein ()+It is defined as pseudo inverse matrix, αl ◇,βl ◇It is the minimum value obtained by solution (D).
CN201811167870.2A 2018-10-08 2018-10-08 CO-OFDM system phase noise optimization compensation method Active CN109347562B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN201811167870.2A CN109347562B (en) 2018-10-08 2018-10-08 CO-OFDM system phase noise optimization compensation method

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN201811167870.2A CN109347562B (en) 2018-10-08 2018-10-08 CO-OFDM system phase noise optimization compensation method

Publications (2)

Publication Number Publication Date
CN109347562A true CN109347562A (en) 2019-02-15
CN109347562B CN109347562B (en) 2020-05-05

Family

ID=65308372

Family Applications (1)

Application Number Title Priority Date Filing Date
CN201811167870.2A Active CN109347562B (en) 2018-10-08 2018-10-08 CO-OFDM system phase noise optimization compensation method

Country Status (1)

Country Link
CN (1) CN109347562B (en)

Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110011734A (en) * 2019-04-18 2019-07-12 杭州电子科技大学 CPE compensation method in CO-OFDM system based on pilot tone and two-dimensional projection's histogram
CN110190906A (en) * 2019-04-15 2019-08-30 浙江工业大学 A kind of nonlinear equalization method for high-order QAM coherent light system
CN110233812A (en) * 2019-04-28 2019-09-13 国网安徽省电力有限公司信息通信分公司 CO-OFDM system and its receiving end and nonlinear equalization method, apparatus
CN110830398A (en) * 2019-11-13 2020-02-21 湖南师范大学 Frequency domain average channel estimation method in symbol applied in optical fiber DMT system
CN111917487A (en) * 2020-08-12 2020-11-10 中国科学院长春光学精密机械与物理研究所 Laser communication interference elimination method and device and computer readable storage medium
CN112769497A (en) * 2021-02-23 2021-05-07 苏州大学 Method for carrying out nonlinear compensation on high-capacity high-order QAM coherent light
CN113114602A (en) * 2021-04-09 2021-07-13 重庆邮电大学 Phase noise compensation method based on combination of cyclic prefix interpolation and CKF in CO-OFDM system
CN114095322A (en) * 2021-09-28 2022-02-25 超讯通信股份有限公司 Phase noise suppression method, phase noise suppression device and storage medium
CN114465864A (en) * 2022-02-17 2022-05-10 哲库科技(北京)有限公司 Phase noise compensation method, phase noise compensation device, terminal, storage medium, and program product
CN115778319A (en) * 2022-11-09 2023-03-14 山东大学 Visible light OCT system based on double-spectrometer light source noise compensation

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8760201B1 (en) * 2013-03-11 2014-06-24 Analog Devices Technology Digitally programmed capacitance multiplication with one charge pump
CN104601239A (en) * 2015-01-12 2015-05-06 西南交通大学 Optical fiber adaptive nonlinear compensation scheme based on intensity noise variance and low-pass filter
CN105187345A (en) * 2015-09-23 2015-12-23 浙江工业大学 Non-iterative blind phase noise compensation method suitable for CO-OFDM system
CN105187344A (en) * 2015-09-23 2015-12-23 浙江工业大学 Decision auxiliary time domain average approximation ICI phase noise compensation method suitable for CO-OFDM system
CN107171735A (en) * 2017-05-16 2017-09-15 浙江工业大学 A kind of big line width CO OFDM phase noise compensation methods of time-frequency domain Kalman filtering
CN107395282A (en) * 2017-07-20 2017-11-24 浙江工业大学 A kind of big line width CO ofdm systems phase noise compensation method of time domain Unscented kalman filtering

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8760201B1 (en) * 2013-03-11 2014-06-24 Analog Devices Technology Digitally programmed capacitance multiplication with one charge pump
CN104601239A (en) * 2015-01-12 2015-05-06 西南交通大学 Optical fiber adaptive nonlinear compensation scheme based on intensity noise variance and low-pass filter
CN105187345A (en) * 2015-09-23 2015-12-23 浙江工业大学 Non-iterative blind phase noise compensation method suitable for CO-OFDM system
CN105187344A (en) * 2015-09-23 2015-12-23 浙江工业大学 Decision auxiliary time domain average approximation ICI phase noise compensation method suitable for CO-OFDM system
CN107171735A (en) * 2017-05-16 2017-09-15 浙江工业大学 A kind of big line width CO OFDM phase noise compensation methods of time-frequency domain Kalman filtering
CN107395282A (en) * 2017-07-20 2017-11-24 浙江工业大学 A kind of big line width CO ofdm systems phase noise compensation method of time domain Unscented kalman filtering

Cited By (17)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN110190906A (en) * 2019-04-15 2019-08-30 浙江工业大学 A kind of nonlinear equalization method for high-order QAM coherent light system
CN110011734B (en) * 2019-04-18 2020-06-30 杭州电子科技大学 CPE compensation method in CO-OFDM system based on pilot frequency and two-dimensional projection histogram
CN110011734A (en) * 2019-04-18 2019-07-12 杭州电子科技大学 CPE compensation method in CO-OFDM system based on pilot tone and two-dimensional projection's histogram
CN110233812B (en) * 2019-04-28 2022-05-24 国网安徽省电力有限公司信息通信分公司 CO-OFDM system and receiving end thereof, and nonlinear equalization method and device
CN110233812A (en) * 2019-04-28 2019-09-13 国网安徽省电力有限公司信息通信分公司 CO-OFDM system and its receiving end and nonlinear equalization method, apparatus
CN110830398A (en) * 2019-11-13 2020-02-21 湖南师范大学 Frequency domain average channel estimation method in symbol applied in optical fiber DMT system
CN111917487A (en) * 2020-08-12 2020-11-10 中国科学院长春光学精密机械与物理研究所 Laser communication interference elimination method and device and computer readable storage medium
CN111917487B (en) * 2020-08-12 2021-08-03 中国科学院长春光学精密机械与物理研究所 Laser communication interference elimination method and device and computer readable storage medium
CN112769497A (en) * 2021-02-23 2021-05-07 苏州大学 Method for carrying out nonlinear compensation on high-capacity high-order QAM coherent light
CN112769497B (en) * 2021-02-23 2022-02-22 苏州大学 Method for carrying out nonlinear compensation on high-capacity high-order QAM coherent light
CN113114602B (en) * 2021-04-09 2022-06-21 重庆邮电大学 Phase noise compensation method based on CP interpolation and CKF in CO-OFDM system
CN113114602A (en) * 2021-04-09 2021-07-13 重庆邮电大学 Phase noise compensation method based on combination of cyclic prefix interpolation and CKF in CO-OFDM system
CN114095322A (en) * 2021-09-28 2022-02-25 超讯通信股份有限公司 Phase noise suppression method, phase noise suppression device and storage medium
CN114095322B (en) * 2021-09-28 2023-07-14 超讯通信股份有限公司 Phase noise suppression method, device and storage medium
CN114465864A (en) * 2022-02-17 2022-05-10 哲库科技(北京)有限公司 Phase noise compensation method, phase noise compensation device, terminal, storage medium, and program product
CN114465864B (en) * 2022-02-17 2024-01-12 哲库科技(北京)有限公司 Phase noise compensation method, apparatus, terminal, storage medium, and program product
CN115778319A (en) * 2022-11-09 2023-03-14 山东大学 Visible light OCT system based on double-spectrometer light source noise compensation

Also Published As

Publication number Publication date
CN109347562B (en) 2020-05-05

Similar Documents

Publication Publication Date Title
CN109347562A (en) A kind of CO-OFDM system phase noise optimization compensation method
Zhong et al. Digital signal processing for short-reach optical communications: A review of current technologies and future trends
CN107395282B (en) A kind of big line width CO-OFDM system phase noise compensation method
CN107171735B (en) A kind of big line width CO-OFDM phase noise compensation method of time-frequency domain Kalman filtering
US9112608B2 (en) Resource-efficient digital chromatic dispersion compensation in fiber optical communication using spectral shaping subcarrier modulation
Horlin et al. Dual-polarization OFDM-OQAM for communications over optical fibers with coherent detection
Chen et al. Zero-guard-interval coherent optical OFDM with overlapped frequency-domain CD and PMD equalization
Yu et al. Polarization demultiplexing in stokes space for coherent optical PDM-OFDM
CN107078982A (en) The dispersive estimates of digital coherent optical receiver
CN112036543B (en) Time domain equalizer combining neural network equalization and linear equalization and equalization method
Bruno et al. Real-time 20.37 Gb/s optical OFDM receiver for PON IM/DD systems
Nhan et al. Sparse preamble design for polarization division multiplexed CO-OFDM/OQAM channel estimation
Hong et al. Linearly interpolated sub-symbol optical phase noise suppression in CO-OFDM system
CN107949992A (en) Frequency domain optical channel is estimated
CN114844749B (en) Fiber channel estimation method based on neural network
WO2023245828A1 (en) Compensation method for distorted signal of multi-carrier access network, and nonlinear equalizer
da Silva et al. Widely linear blind adaptive equalization for transmitter IQ-imbalance/skew compensation in multicarrier systems
Yin et al. Multi-band DFT-S 100 Gb/s 32 QAM-DMT transmission in intra-DCI using 10 G-class EML and low-resolution DAC
Xi et al. Simplified SVM equalization algorithm based on single hyperplane training enabled 50Gb/s PAM-4/8 with 10-G optics in NG-PON system
Yang et al. A modified convolutional neural network-based signal demodulation method for direct detection OFDM/OQAM-PON
Hong et al. Low-complexity optical phase noise suppression in CO-OFDM system using recursive principal components elimination
Zhang et al. Channel estimation based on linear interpolation algorithm in DDO-OFDM system
Bernhard et al. Multicarrier transmission using Hadamard transform for optical communications
Bai et al. Development of discrete Fourier transform‐based channel estimation algorithms for a coherent optical orthogonal frequency division multiplexing transmission system
Lan et al. A LMMSE channel estimator for coherent optical OFDM system

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant