CN108964731B - A fast convolution-free hybrid carrier continuous stream transmission method without cyclic prefix filtering - Google Patents
A fast convolution-free hybrid carrier continuous stream transmission method without cyclic prefix filtering Download PDFInfo
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Abstract
基于快速卷积的无循环前缀滤波混合载波连续流传输方法,用于无线通信领域。本发明可解决现有方法带外泄露高,以及现有方法每个混合载波符号前加入循环前缀导致频谱效率低的问题。本发明多个子带同时并行传输,每个子带进行加权傅里叶变换预处理后,经由滤波映射到不同的频段上。本发明使用快速卷积完成滤波,能有效地抑制子带的带外泄露。各子带可灵活的选取子带宽度,并设定不同的加权傅里叶变换调制阶数。不同子带的调制阶数既可以相同,也可以根据均衡法则设定各自最优的调制阶数,以使得误比特率性能最优。并且,调整调制阶数也能抑制峰均比。可结合系统对峰均比的要求,兼顾接收机采用的均衡法则,选取合适的调制阶数。
A fast convolution-free mixed-carrier continuous stream transmission method without cyclic prefix filtering is used in the field of wireless communication. The invention can solve the problems of high out-of-band leakage in the existing method and low spectral efficiency caused by adding a cyclic prefix before each mixed carrier symbol in the existing method. In the present invention, multiple subbands are transmitted in parallel at the same time, and each subband is preprocessed by weighted Fourier transform, and then mapped to different frequency bands through filtering. The invention uses fast convolution to complete filtering, and can effectively suppress out-of-band leakage of sub-bands. Each sub-band can flexibly select the sub-band width and set different weighted Fourier transform modulation orders. The modulation orders of different subbands may be the same, or their optimal modulation orders may be set according to the equalization rule, so as to optimize the bit error rate performance. In addition, adjusting the modulation order can also suppress the peak-to-average ratio. The appropriate modulation order can be selected according to the requirements of the system on the peak-to-average ratio and the equalization rule adopted by the receiver.
Description
技术领域technical field
本发明涉及无线通信领域,具体涉及基于快速卷积的无循环前缀滤波混合载波连续流传输方法。The present invention relates to the field of wireless communication, in particular to a method for continuous stream transmission of mixed carriers without cyclic prefix filtering based on fast convolution.
背景技术Background technique
在现代通信系统中,正交频分复用(Orthogonal Frequency DivisionMultiplexing,OFDM)技术由于具备高频谱效率和强抗多径衰落能力而得到广泛应用。但其存在较高的旁瓣功率,因而对传输的同步有严格的要求。为了抑制带外功率以降低系统对同步的要求,学者们提出了多种技术,如滤波器组多载波(Filter bank multi-carrier,FBMC)、滤波OFDM(Filtered-OFDM)、广义频分复用(Generalized frequency divisionmultiplexing,GFDM)和通用滤波多载波(Universal filtered multi-carrier,UFMC)(Vakilian,V.;Wild,T.;Schaich,F.;ten Brink,S.&Frigon,J.F.Universal-filteredmulti-carrier technique for wireless systems beyond LTE 2013 IEEE GlobecomWorkshops(GC Wkshps),2013,223-228)等。这些技术都使用了滤波来抑制带外功率泄露。同时,近年来,一种基于加权分数傅里叶变换(Weighted-type fractional Fouriertransform,WFRFT)的混合载波设计理念(Mei L,Sha X J,Ran Q W,et al.Research onthe application of 4-weighted fractional Fourier transform in communicationsystem[J].Science China(Information Sciences),2010,53(6):1251-1260.)由于能整合传统的单载波与多载波体制得到了广泛的关注。在《加权类分数傅里叶变换及其在通信系统中的应用》(梅林.加权类分数傅里叶变换及其在通信系统中的应用[D].哈尔滨:哈尔滨工业大学,2010:51-60)中,基于WFRFT的混合载波体系成功整合了传统的单载波与多载波体制,以加权组合的方式融合两种不同的载波分量。混合载波信号能量在时频平面能均匀分布,可提升系统的抗干扰性能(Wang K,Sha X,Mei L,et al.Performance Analysisof Hybrid Carrier System with MMSE Equalization over Doubly-DispersiveChannels[J].IEEE Communications Letters,2012,16(7):1048-1051)。原有WFRFT混合载波传输方法在载波频率偏移(Mei L,Zhang Q,Sha X,et al.WFRFT Precoding forNarrowband Interference Suppression in DFT-Based Block Transmission Systems[J].IEEE Communications Letters,2013,17(10):1916-1919)以及符号间干扰和载波间干扰抑制(Li Y,Sha X,Zheng F C,et al.Low Complexity Equalization of HCMSystems with DPFFT Demodulation over Doubly-Selective Channels[J].IEEE SignalProcessing Letters,2014,21(7):862-865)等方向具有良好性能,得到了学界与产业界的重视。In modern communication systems, Orthogonal Frequency Division Multiplexing (OFDM) technology is widely used due to its high spectral efficiency and strong anti-multipath fading capability. However, it has high side lobe power, so it has strict requirements on the synchronization of transmission. In order to suppress the out-of-band power and reduce the synchronization requirements of the system, scholars have proposed a variety of techniques, such as Filter bank multi-carrier (FBMC), Filtered-OFDM (Filtered-OFDM), Generalized Frequency Division Multiplexing (Generalized frequency division multiplexing, GFDM) and universal filtered multi-carrier (Universal filtered multi-carrier, UFMC) (Vakilian, V.; Wild, T.; Schaich, F.; ten Brink, S. & Frigon, J.F. Universal-filteredmulti-carrier technique for wireless systems beyond LTE 2013 IEEE Globecom Workshops (GC Wkshps), 2013, 223-228) et al. These techniques all use filtering to suppress out-of-band power leakage. Meanwhile, in recent years, a hybrid carrier design concept based on Weighted-type fractional Fourier transform (WFRFT) (Mei L, Sha X J, Ran Q W, et al. Research on the application of 4-weighted fractional Fourier transform in communication system [J]. Science China (Information Sciences), 2010, 53(6): 1251-1260.) has received extensive attention due to its ability to integrate traditional single-carrier and multi-carrier systems. In "Weighted Fractional Fourier Transform and Its Application in Communication Systems" (Merlin. Weighted Fractional Fourier Transform and Its Application in Communication Systems [D]. Harbin: Harbin Institute of Technology, 2010: 51- 60), the WFRFT-based hybrid carrier system successfully integrates the traditional single-carrier and multi-carrier systems, and fuses two different carrier components in a weighted combination. The energy of the hybrid carrier signal can be evenly distributed in the time-frequency plane, which can improve the anti-interference performance of the system (Wang K, Sha X, Mei L, et al. Performance Analysis of Hybrid Carrier System with MMSE Equalization over Doubly-Dispersive Channels [J]. IEEE Communications Letters, 2012, 16(7):1048-1051). The original WFRFT hybrid carrier transmission method is in the carrier frequency offset (Mei L, Zhang Q, Sha X, et al. WFRFT Precoding for Narrowband Interference Suppression in DFT-Based Block Transmission Systems [J]. IEEE Communications Letters, 2013, 17 (10 ): 1916-1919) and Inter-Symbol Interference and Inter-Carrier Interference Suppression (Li Y, Sha X, Zheng F C, et al. Low Complexity Equalization of HCMSystems with DPFFT Demodulation over Doubly-Selective Channels [J]. IEEE SignalProcessing Letters, 2014 , 21(7):862-865) and other directions have good performance and have been paid attention by the academic and industrial circles.
发明内容SUMMARY OF THE INVENTION
本发明的目的是为了解决现有方法带外泄露高,以及现有方法是按“块”传输,每个混合载波符号前加入循环前缀,导致频谱效率低的缺点,而提出基于快速卷积的无循环前缀滤波混合载波连续流传输方法。The purpose of the present invention is to solve the shortcomings of high out-of-band leakage of the existing method, and the existing method is to transmit in "block", adding a cyclic prefix before each mixed carrier symbol, resulting in low spectral efficiency, and proposes a fast convolution based method. A method for continuous stream transmission of hybrid carriers without cyclic prefix filtering.
基于快速卷积的无循环前缀滤波混合载波连续流传输方法包括以下步骤:The fast convolution-free cyclic prefix filtering hybrid carrier continuous stream transmission method includes the following steps:
步骤一:每一子带的连续数据流Dk独立生成后,k=1,2,…,K,K为子带的个数,将Dk按给定长度Q分成P段,将P段数据流采用预编码矩阵进行预编码后,得到与编码后的数据p=1,2,…,P,再将P段预编码后的数据拼接成连续的数据流sk(n),n=0,1,…,m-1,m为数据流的总点数;Step 1: After the continuous data stream Dk of each subband is independently generated, k =1, 2,...,K, where K is the number of subbands, divide Dk into P segments according to a given length Q, and divide the P segments into P segments. After the data stream is precoded by the precoding matrix, the coded data is obtained p=1,2,...,P, and then splicing the precoded data of P segments into a continuous data stream sk (n), n=0,1,...,m-1, m is the total number of points in the data stream ;
步骤二:对预编码后的连续数据流sk(n)进行长度为Lk,b的不重叠分块,对于第q个分块的信号,表示为sk,q,取出三个连续不重叠分块信号sk,q-1,sk,q,sk,q+1的中间的Nk,b个符号,得到重叠分块信号其中,Nk,b是Lk,b的2倍;当q=1时,在两个连续不重叠分块信号sk,1,sk,2前补Lk,b个零后取出中间的Nk,b个符号,完成前述的重叠分块操作。Step 2: Perform non-overlapping blocks of length L k,b on the precoded continuous data stream sk (n), for the signal of the qth block, it is expressed as sk,q , Take out N k,b symbols in the middle of three consecutive non-overlapping block signals sk,q-1 , sk,q , sk,q+1 to obtain overlapping block signals Among them, N k,b is 2 times of L k,b ; when q=1, after two consecutive non-overlapping block signals sk,1 , sk,2 are filled with L k,b zeros and the middle is taken out The N k,b symbols of , complete the aforementioned overlapping block operation.
步骤三:将重叠分块信号进行Nk,b点的傅里叶变换;Step 3: Block the overlapping signal Carry out the Fourier transform of the N k,b point;
步骤四:将傅里叶变换结果直接复制Mk次并首尾拼接形成N点的数据,与该子带对应的N点的频域滤波器的系数ek进行点乘,完成频域滤波,得到各子带频域滤波后的结果Pk;Step 4: Copy the Fourier transform result directly M k times and splicing it end to end to form the data of N points, perform dot multiplication with the coefficient ek of the frequency domain filter of N points corresponding to the subband, complete the frequency domain filtering, and obtain The result P k after each subband is filtered in the frequency domain;
步骤五:将步骤四得到的各子带频域滤波的结果加和后进行N点的傅里叶逆变换,得到 Step 5: After adding the results of the frequency domain filtering of each subband obtained in
步骤六:取出的中间L个符号,完成重叠保留操作,得到输出信号y(n)的不重叠分块信号将得到的个分块信号yq拼接得到y(n),并经过上变频处理后发出;其中,2L=N;Step 6: Take out The middle L symbols of , complete the overlapping reservation operation, and obtain the non-overlapping block signal of the output signal y(n). will get A block signal y q is spliced to obtain y(n), and sent out after up-conversion processing; wherein, 2L=N;
步骤七:对接收到的步骤六发出并经过信道后的信号进行下变频处理得到混合载波基带信号r(n),对混合载波基带信号r(n)进行长度为L的不重叠分块;对于第q个分块的信号,表示为rq,取出三个连续不重叠分块信号rq-1,rq,rq+1的中间N个符号,得到重叠分块信号其中,L是2的幂,N是L的2倍;Step 7: Perform down-conversion processing on the received signal sent in
步骤八:将重叠分块信号进行N点的傅里叶变换;Step 8: Block the overlapping signal Perform Fourier transform of N points;
步骤九:将步骤八进行傅里叶变换后的分块信号进行均衡和子带滤波;Step 9: performing equalization and subband filtering on the block signal after the Fourier transform in
步骤十:将步骤九进行均衡和子带滤波后的数据进行Nk,b点傅里叶逆变换,得到其中,Nk,b=2Lk,b,Lk,b=L/Mk,Mk为第k个子带对应的插值倍数;Step 10: Perform inverse Fourier transform on the data after equalization and sub-band filtering in
步骤十一:保留各分块的中间Lk,b个数据xk,q,组成数据流xk;再按长度Q分成P段,每段记为xk,p,各段分别进行逆预编码变换,得到再将逆预编码后的P个结果组合得到各子带的接收信号 Step 11: Keep each block The middle L k,b pieces of data x k,q form the data stream x k ; then it is divided into P segments according to the length Q, and each segment is denoted as x k,p , and each segment is reversely precoded transform, get Then combine the P results after inverse precoding to obtain the received signal of each subband
本发明的有益效果为:The beneficial effects of the present invention are:
本发明将子带滤波引入混合载波系统来抑制带外泄露,拓展了传统混合载波系统的结构。同时,原有混合载波传输方法是按“块”传输,即每个混合载波符号前加入循环前缀(Cyclic Prefix,CP),而本发明提出的是连续流传输方法,混合载波符号不加CP,提高了频谱效率。The invention introduces subband filtering into the mixed carrier system to suppress out-of-band leakage, and expands the structure of the traditional mixed carrier system. At the same time, the original hybrid carrier transmission method is based on "block" transmission, that is, a cyclic prefix (Cyclic Prefix, CP) is added before each hybrid carrier symbol, while the present invention proposes a continuous stream transmission method, and the hybrid carrier symbol does not add CP, Increased spectral efficiency.
已有的子带滤波通信系统中,各子带传输波形是单载波或者是多载波信号,本发明提出各子带可灵活的选取所传输的波形,传输的信号为混合载波信号,其中,单载波与多载波信号为两个特例。本发明发明了一种无循环前缀滤波混合载波连续流传输方法,多个子带同时并行传输,每个子带进行加权傅里叶变换预处理后,经由滤波映射到不同的频段上。而快速卷积能完成时域滤波的线性卷积过程,并具有低复杂度与高灵活性的好处。本发明使用快速卷积完成滤波,能有效地抑制子带的带外泄露。各子带可灵活的选取子带宽度,并设定不同的加权傅里叶变换调制阶数。不同子带的加权傅里叶变换调制阶数既可以相同,也可以根据均衡法则设定各自最优的调制阶数,以使得误比特率性能最优。并且,调整调制阶数也能抑制峰均比(Peak to Average Power Ratio,PAPR)。在实施例给定的参数下,子带间的带外抑制泄露相比原有的混合载波系统下降了20dB左右。实施例中也可看出本发明的单一子带的PAPR得到了有效的抑制。同时,本发明所提系统不再依赖CP的使用,提高了系统的频谱效率。本发明将所提系统命名为无循环前缀滤波混合载波系统,其是对传统混合载波系统的拓展,进一步灵活了波形设计,抑制了子带的带外泄露,并不使用CP,提高了系统的频谱效率。In the existing sub-band filtering communication system, the transmission waveform of each sub-band is a single carrier or a multi-carrier signal. The present invention proposes that each sub-band can flexibly select the transmitted waveform, and the transmitted signal is a mixed carrier signal, wherein the single Carrier and multi-carrier signals are two special cases. The invention provides a continuous stream transmission method of mixed carrier without cyclic prefix filtering. Multiple sub-bands are transmitted in parallel at the same time. After each sub-band is preprocessed by weighted Fourier transform, it is mapped to different frequency bands through filtering. The fast convolution can complete the linear convolution process of time domain filtering, and has the advantages of low complexity and high flexibility. The invention uses fast convolution to complete filtering, and can effectively suppress out-of-band leakage of sub-bands. Each sub-band can flexibly select the sub-band width and set different weighted Fourier transform modulation orders. The weighted Fourier transform modulation orders of different sub-bands can be the same, or the optimal modulation orders can be set according to the equalization rule, so as to optimize the bit error rate performance. In addition, adjusting the modulation order can also suppress the Peak to Average Power Ratio (PAPR). Under the parameters given in the embodiment, the out-of-band leakage suppression between subbands is reduced by about 20dB compared with the original hybrid carrier system. It can also be seen from the examples that the PAPR of a single subband of the present invention is effectively suppressed. At the same time, the system proposed in the present invention no longer relies on the use of CP, which improves the spectral efficiency of the system. The present invention names the proposed system as a cyclic prefix-free filtering hybrid carrier system, which is an extension of the traditional hybrid carrier system, further flexes the waveform design, suppresses the out-of-band leakage of the sub-band, does not use CP, and improves the system reliability. Spectral efficiency.
附图说明Description of drawings
图1为本发明框图;Fig. 1 is a block diagram of the present invention;
图2为本发明系统接收机模块示意图;2 is a schematic diagram of a system receiver module of the present invention;
图3为本发明与传统混合载波系统的单子带功率谱对比图;Fig. 3 is the single subband power spectrum comparison diagram of the present invention and the traditional hybrid carrier system;
图4为本发明与传统混合载波系统的功率谱对比图;4 is a power spectrum comparison diagram of the present invention and a traditional hybrid carrier system;
图5为本发明与传统混合滤波系统的单一子带的峰均功率比对比图;5 is a comparison diagram of the peak-to-average power ratio of a single subband of the present invention and a traditional hybrid filter system;
图6为ZF均衡准则下,本发明在固定频选信道下的误比特率性能图;Fig. 6 is under the ZF equalization criterion, the bit error rate performance diagram of the present invention under the fixed frequency selection channel;
图7为MMSE均衡准则下,本发明在固定频选信道下的误比特率性能图;7 is a bit error rate performance diagram of the present invention under a fixed frequency selection channel under the MMSE equalization criterion;
图8为ZF均衡准则下,本发明随机频选信道下误比特率性能图;FIG. 8 is a bit error rate performance diagram under the random frequency selection channel of the present invention under the ZF equalization criterion;
图9为MMSE均衡准则下,本发明随机频选信道下误比特率性能图;FIG. 9 is a bit error rate performance diagram under the random frequency selection channel of the present invention under the MMSE equalization criterion;
图10为数据流重叠分块示意图。FIG. 10 is a schematic diagram of overlapping blocks of data streams.
具体实施方式Detailed ways
具体实施方式一:基于快速卷积的无循环前缀滤波混合载波连续流传输方法包括以下步骤:Embodiment 1: The method for continuous stream transmission of mixed carriers without cyclic prefix filtering based on fast convolution includes the following steps:
本发明所提出的系统结构如图1所示。发送机部分由一组插值滤波器构成,接收机部分由一组相应的采样滤波器构成。专利的核心在于将发射端的每个子带可选择不同的调制阶数进行WFRFT预编码,在接收端每个子带经过匹配滤波与均衡处理后,相应的子带作逆预编码处理,从而把符号判决位置从时域变换到分数域。预编码的操作使得波形的选择变得灵活,针对不同的检测算法,选取最优的调制阶数α,可使得误比特性能最优。并可兼顾系统对PAPR的要求,调整α。由文献(Mei L,Sha X J,Ran Q W,et al.Research on theapplication of 4-weighted fractional Fourier transform in communicationsystem[J].Science China(Information Sciences),2010,53(6):1251-1260),经由WFRFT预编码的多载波调制过程可视为混合载波系统。随着α的变化,其单载波分量和多载波分量的比重在发生变化。在本发明提出的系统中,每个用户占据一个子带,不同的子带可选取相同的调制阶数,也可选取不同的调制阶数,因此本发明提出的传输方法属于灵活的混合载波传输。同时,本发明的另一个核心是,利用快速卷积完成无CP混合载波系统各子带的滤波操作,并针对发射机的快速卷积滤波结构,在接收端基于快速卷积算法完成了均衡处理与相应的匹配滤波。该结构不仅能有效抑制子带的带外泄露,同时,不同于传统的混合载波系统,该结构不再依赖CP的使用,即发射端传输的是无CP的混合载波连续流。The system structure proposed by the present invention is shown in FIG. 1 . The transmitter part consists of a set of interpolation filters, and the receiver part consists of a set of corresponding sampling filters. The core of the patent is to select different modulation orders for WFRFT precoding for each subband at the transmitting end. After each subband at the receiving end is subjected to matched filtering and equalization processing, the corresponding subband is subjected to inverse precoding processing, so as to determine the symbol. The position is transformed from the time domain to the fractional domain. The operation of precoding makes the selection of waveform flexible. For different detection algorithms, selecting the optimal modulation order α can make the bit error performance optimal. And can take into account the system's requirements for PAPR, adjust α. From the literature (Mei L, Sha X J, Ran Q W, et al. Research on the application of 4-weighted fractional Fourier transform in communication system [J]. Science China (Information Sciences), 2010, 53(6): 1251-1260), The multi-carrier modulation process via WFRFT precoding can be regarded as a mixed-carrier system. With the change of α, the proportion of the single-carrier component and the multi-carrier component is changing. In the system proposed by the present invention, each user occupies one subband, and different subbands can select the same modulation order or different modulation orders, so the transmission method proposed by the present invention belongs to flexible mixed carrier transmission . At the same time, another core of the present invention is to use fast convolution to complete the filtering operation of each subband of the non-CP hybrid carrier system, and for the fast convolution filtering structure of the transmitter, the equalization processing is completed at the receiving end based on the fast convolution algorithm with the corresponding matched filter. This structure can not only effectively suppress out-of-band leakage of sub-bands, but also, unlike the traditional hybrid carrier system, this structure no longer relies on the use of CP, that is, the transmitter transmits a continuous stream of hybrid carriers without CP.
如图1所示,对于本发明系统,调制过程可以映射为若干路多载波基带信号先经由WFRFT预编码处理生成混合载波信号后,再通过合成滤波器组,解调过程可以映射为基带接收信号通过分析滤波器组,再经过逆预编码处理。具体来说,每一路预编码后信号通过相应的插值滤波器进行上采样和滤波后,信号被调制至某个载频并限制在相应的频段内,再经过叠加形成合成信号。载波的个数等于插值滤波器的个数。在接收端,合成信号经过一组采样滤波器进行滤波和下采样后,分别得到各路的原始预编码信号,再经由逆预编码处理,得到原始信号。As shown in Fig. 1, for the system of the present invention, the modulation process can be mapped to several multi-carrier baseband signals. After the mixed carrier signal is generated through WFRFT precoding processing, the demodulation process can be mapped to the baseband received signal through the synthesis filter bank. By analyzing the filter bank, it is processed by inverse precoding. Specifically, after each precoded signal is up-sampled and filtered by a corresponding interpolation filter, the signal is modulated to a certain carrier frequency and limited to a corresponding frequency band, and then superimposed to form a composite signal. The number of carriers is equal to the number of interpolation filters. At the receiving end, after the synthesized signal is filtered and down-sampled by a set of sampling filters, the original precoded signals of each channel are obtained respectively, and then the original signals are obtained through inverse precoding processing.
本发明所提系统是以快速卷积的方式完成了插值滤波器的实现。对于每一子带,都可认为是以插值滤波的方式完成了基带信号的映射。而每一子带传送的数据都经由相应的预编码处理。这里可假设每一子带连续流信号是-α阶的加权分数域信号,经过α阶的WFRFT变换后,原始信号就由-α阶的加权分数域信号变换成了时域信号。本发明使用了重叠保留法在频域完成了时域滤波的过程,类似的,可使用重叠相加法在频域完成时域滤波的过程。The system proposed in the present invention completes the realization of the interpolation filter by means of fast convolution. For each subband, it can be considered that the mapping of the baseband signal is completed by means of interpolation filtering. The data transmitted in each subband is processed by corresponding precoding. It can be assumed here that each subband continuous flow signal is a weighted fractional domain signal of order -α. After the WFRFT transformation of order α, the original signal is transformed from a weighted fractional domain signal of order -α to a time domain signal. The present invention uses the overlap-preserving method to complete the time-domain filtering process in the frequency domain, and similarly, the overlap-add method can be used to complete the time-domain filtering process in the frequency domain.
在本发明所提系统中,对于每一子带,都可认为是以基于快速卷积的插值滤波的方式完成了基带信号的映射。每一子带输入的为连续的数据流,以k标识不同的子带。记Lk,b与L分别为插值前与插值后的不重叠分段的长度,Nk,b与N为相应的重叠分段长度,下标b指明Lk,b与Nk,b为第k个子带的插值前的不重叠分段长度和重叠分段长度。其中,Nk,b=N/Mk为整数,Mk为第k子带的插值倍数及采样倍数,N是2的幂。In the system proposed in the present invention, for each subband, it can be considered that the mapping of the baseband signal is completed by means of interpolation filtering based on fast convolution. The input of each subband is a continuous data stream, and different subbands are identified by k. Let L k,b and L be the lengths of non-overlapping segments before and after interpolation, respectively, N k,b and N are the corresponding overlapping segment lengths, and the subscript b indicates that L k,b and N k,b are Non-overlapping segment length and overlapping segment length before interpolation for the kth subband. Wherein, N k,b =N/M k is an integer, M k is an interpolation multiple and a sampling multiple of the kth subband, and N is a power of 2.
步骤一:每一子带的连续数据流Dk独立生成后,k=1,2,…,K,K为子带的个数,将Dk按给定长度Q分成P段,将P段数据流采用预编码矩阵进行预编码后,得到预编码后的数据p=1,2,…,P,再将P段预编码后的数据拼接成连续的数据流sk(n),n=0,1,…,m-1;Step 1: After the continuous data stream Dk of each subband is independently generated, k =1, 2,...,K, where K is the number of subbands, divide Dk into P segments according to a given length Q, and divide the P segments into P segments. The data stream uses a precoding matrix After precoding, the precoded data is obtained p=1,2,...,P, and then splicing the precoded data of P segments into a continuous data stream sk (n), n=0,1,...,m-1;
步骤二:对预编码后的连续数据流sk(n)进行长度为Lk,b的不重叠分块,对于第q个分块的信号,表示为sk,q,sk,q=[sk,q(0) sk,q(1)…sk,q(Lk,b-1)]T;取出三个连续不重叠分块信号sk,q-1,sk,q,sk,q+1的中间的Nk,b个符号,得到重叠分块信号 其中,Nk,b是Lk,b的2倍;当q=1时,在两个连续不重叠分块信号sk,1,sk,2前补Lk,b个零后取出中间的Nk,b个符号,完成前述的重叠分块操作。Step 2: Perform non-overlapping blocks of length L k,b on the precoded continuous data stream sk (n), for the signal of the qth block, it is expressed as sk,q , sk,q = [s k,q (0) s k,q (1)…s k,q (L k,b -1)] T ; take out three consecutive non-overlapping block signals s k,q-1 ,s k, N k, b symbols in the middle of q , s k, q+1 to obtain overlapping block signals Among them, N k,b is 2 times of L k,b ; when q=1, after two consecutive non-overlapping block signals sk,1 , sk,2 are filled with L k,b zeros and the middle is taken out The N k,b symbols of , complete the aforementioned overlapping block operation.
步骤三:将重叠分块信号进行Nk,b点的傅里叶变换;Step 3: Block the overlapping signal Carry out the Fourier transform of the N k,b point;
步骤四:将傅里叶变换结果直接复制Mk次并首尾拼接形成N点的数据,与该子带对应的N点的频域滤波器的系数ek进行点乘,完成频域滤波,得到各子带频域滤波后的结果Pk;Step 4: Copy the Fourier transform result directly M k times and splicing it end to end to form the data of N points, perform dot multiplication with the coefficient ek of the frequency domain filter of N points corresponding to the subband, complete the frequency domain filtering, and obtain The result P k after each subband is filtered in the frequency domain;
步骤五:将步骤四得到的各子带频域滤波的结果加和后进行N点的傅里叶逆变换,得到 Step 5: After adding the results of the frequency domain filtering of each subband obtained in
步骤六:取出的中间L个符号,完成重叠保留操作,得到输出信号y(n)的不重叠分块信号yq,yq=[yq(0) yq(1)…yq(L-1)]T;将得到的个分块信号yq拼接得到y(n),并经过上变频处理后发出;其中,2L=N;Step 6: Take out The middle L symbols of , complete the overlapping reservation operation, and obtain the non-overlapping block signal y q of the output signal y(n), y q =[y q (0) y q (1)...y q (L-1)] T ; will get A block signal y q is spliced to obtain y(n), and sent out after up-conversion processing; wherein, 2L=N;
步骤七:对接收到的步骤六发出并经过信道后的信号进行下变频处理得到混合载波基带信号r(n),对混合载波基带信号r(n)进行长度为L的不重叠分块;对于第q个分块的信号,表示为rq,rq=[rq(0) rq(1)…rq(L-1)]T;取出三个连续不重叠分块信号rq-1,rq,rq+1的中间N个符号,得到重叠分块信号 其中,L是2的幂,N是L的2倍;Step 7: Perform down-conversion processing on the received signal sent in
步骤八:将重叠分块信号进行N点的傅里叶变换;Step 8: Block the overlapping signal Perform Fourier transform of N points;
步骤九:将步骤八进行傅里叶变换后的分块信号进行均衡和子带滤波;Step 9: performing equalization and subband filtering on the block signal after the Fourier transform in
步骤十:将步骤九进行均衡和子带滤波后的数据进行Nk,b点傅里叶逆变换,得到 其中,Nk,b=2Lk,b,Lk,b=L/Mk,Mk为第k个子带对应的插值倍数;Step 10: Perform inverse Fourier transform on the data after equalization and sub-band filtering in
步骤十一:保留各分块的中间Lk,b个数据xk,q,组成数据流xk;再按长度Q分成P段,每段记为p=1,2,…,P,各段分别进行逆预编码变换,得到再将逆预编码后的P个结果组合得到各子带的接收信号 Step 11: Keep each block The middle L k,b data x k,q form the data stream x k ; then it is divided into P segments according to the length Q, and each segment is denoted as p=1,2,...,P, each segment is reversely precoded transform, get Then combine the P results after inverse precoding to obtain the received signal of each subband
缩略语定义:Abbreviation Definitions:
主要变量的说明:Description of the main variables:
滤波器说明及复杂度分析:Filter description and complexity analysis:
本发明以根升余弦滤波器为例,并使用快速卷积的方式在频域完成了时域滤波的等效操作。本发明的仿真实例中,以滚降系数β=0.125的根升余弦滤波器进行说明。实际系统,可结合需要,进行滚降系数调整。但本发明所提系统不限于使用根升余弦滤波器。有相关文献对子带滤波器的优化设计方法进行了研究。在文献(Yli-Kaakinen J,RenforsM.Optimization of flexible filter banks based on fast-convolution[C]//IEEEInternational Conference on Acoustics,Speech and Signal Processing.IEEE,2014:1-11)中,提出了以最小化滤波器的通频带和阻带波动为准则的最优化设计方法;在文献(Yli-Kaakinen J,Renfors M.Optimized burst truncation in fast-convolutionfilter bank based waveform generation[C]//IEEE,International Workshop onSignal Processing Advances in Wireless Communications.IEEE,2015:71-75)中,提出了以时频局部化特性为第一目标的滤波器设计方法。技术人员可参考有关资料进行滤波器设计。The present invention takes the root raised cosine filter as an example, and uses the fast convolution method to complete the equivalent operation of time domain filtering in the frequency domain. In the simulation example of the present invention, a root raised cosine filter with roll-off coefficient β=0.125 is used for description. For the actual system, the roll-off coefficient can be adjusted according to the needs. However, the system proposed in the present invention is not limited to using a root raised cosine filter. There are relevant literatures on the optimization design method of subband filter. In the literature (Yli-Kaakinen J, Renfors M. Optimization of flexible filter banks based on fast-convolution [C]//IEEE International Conference on Acoustics, Speech and Signal Processing. IEEE, 2014: 1-11), it is proposed to minimize The optimal design method based on the filter passband and stopband fluctuations; in the literature (Yli-Kaakinen J, Renfors M. Optimized burst truncation in fast-convolution filter bank based waveform generation [C]//IEEE, International Workshop on Signal Processing In Advances in Wireless Communications. IEEE, 2015:71-75), a filter design method with time-frequency localization characteristics as the first goal is proposed. Technicians can refer to relevant materials for filter design.
本发明的一大特色是,针对混合载波系统,采用快速卷积的方式完成滤波,相比传统的时域滤波,极大地降低了算法复杂度。同时,在实现上,快速卷积的操作可使用并行处理,加快处理速度。A major feature of the present invention is that, for the hybrid carrier system, the filtering is completed by means of fast convolution, which greatly reduces the complexity of the algorithm compared with the traditional time domain filtering. At the same time, in terms of implementation, the operation of fast convolution can use parallel processing to speed up the processing speed.
与传统混合载波系统相比,本发明系统增加了滤波的操作,提高了算法复杂度。但基于快速卷积的操作,相比传统的时域滤波,又显著地减小了复杂度。Compared with the traditional hybrid carrier system, the system of the present invention increases the filtering operation and improves the algorithm complexity. However, the operation based on fast convolution significantly reduces the complexity compared with the traditional temporal filtering.
为在统一的标准下进行比较,设混合载波系统载波总数为N,实际使用L个子载波,复杂度的衡量标准为发送LU个符号需要的乘法运算次数。为了对比,表1中也列出了传统的OFDM系统。针对本发明系统,对于某一个归一化带宽为Nk,b/N并采用滚降系数为β的RRC滤波器的子带k,发送ULk,b个符号所需的计算量可近似为(ULk,blogQ+4ULk,b)+(UNlogN)+UNk,b(β+logNk,b)。其中,Q为ULk,b个符号组成的连续流的分段长度。For comparison under a unified standard, the total number of carriers in the mixed-carrier system is set to N, L sub-carriers are actually used, and the measure of complexity is the number of multiplication operations required to transmit LU symbols. For comparison, traditional OFDM systems are also listed in Table 1. For the system of the present invention, for a subband k of an RRC filter with a normalized bandwidth of N k,b /N and a roll-off coefficient of β, the calculation amount required to transmit UL k,b symbols can be approximated as (UL k,b logQ+4UL k,b )+(UNlogN)+UN k,b (β+logN k,b ). Among them, Q is the segment length of the continuous stream composed of UL k,b symbols.
表1几种多载波技术的发射机计算复杂度Table 1 Transmitter computational complexity of several multi-carrier technologies
如表1所示,列出了4个系统的近似发射机复杂度。对于本发明系统,由于频域滤波系数在阻带上为0,在大部分通带上为1,可节省运算量。不过,快速卷积在进行快速傅里叶变换/快速傅里叶逆变换计算前后,重叠保留的操作在一定程度上降低了计算效率。而对于时域滤波混合载波系统,时域线性卷积的滤波操作极大提高了算法复杂度。基于快速卷积在频域完成滤波,与在时域完成滤波本质上是等效的,都能抑制带外泄露,但极大地减小了算法复杂度。值得指出的是,本发明不是简单地使用快速卷积替代时域滤波,而是相应地对传统混合载波接收机结构进行设计,使得接收端的匹配滤波与均衡操作能进行有机融合。从而,让混合载波信号不再依赖CP的使用,并提高了频谱效率。但是各子带采用根升余弦滤波器,再映射到相应频带,由于为了抑制带外引入了过渡带,也导致各子带所占子载波数增加,亦会带来频谱效率的下降。故本发明所提系统相比在时域滤波的混合载波系统,能有效提高频谱效率。但与未滤波的混合载波系统,其频谱效率无明显提高。As shown in Table 1, the approximate transmitter complexity of the four systems is listed. For the system of the present invention, since the frequency domain filter coefficient is 0 in the stopband and 1 in most of the passband, the computational complexity can be saved. However, before and after the fast Fourier transform/inverse fast Fourier transform calculation of fast convolution, the operation of overlapping preservation reduces the computational efficiency to a certain extent. For time-domain filtering mixed-carrier systems, the filtering operation of time-domain linear convolution greatly increases the complexity of the algorithm. Filtering in the frequency domain based on fast convolution is essentially equivalent to filtering in the time domain, which can suppress out-of-band leakage, but greatly reduces the algorithm complexity. It is worth noting that the present invention does not simply use fast convolution to replace time domain filtering, but designs the structure of the traditional hybrid carrier receiver accordingly, so that matched filtering and equalization operations at the receiving end can be organically integrated. Therefore, the mixed carrier signal is no longer dependent on the use of the CP, and the spectral efficiency is improved. However, each subband adopts a root raised cosine filter and maps it to the corresponding frequency band. Since a transition band is introduced to suppress the out-of-band, the number of subcarriers occupied by each subband also increases, which will also bring about a decrease in spectral efficiency. Therefore, compared with the hybrid carrier system filtered in the time domain, the system proposed in the present invention can effectively improve the spectral efficiency. However, compared with the unfiltered hybrid carrier system, the spectral efficiency is not significantly improved.
具体实施方式二:本实施方式与具体实施方式一不同的是:所述步骤一中将Dk按给定长度Q分成P段,将P段数据流采用预编码矩阵进行预编码后,得到与编码后的数据p=1,2,…,P,再将P段预编码后的数据拼接成连续的数据流sk(n)的具体过程为:Embodiment 2: This embodiment differs from
Dk=[DT k,1 DT k,2…DT k,P]T D k =[D T k,1 D T k,2 ...D T k,P ] T
Dk,p=[Dk,p(0) Dk,p(1)…Dk,p(Q-1)]T D k,p = [D k,p (0) D k,p (1)…D k,p (Q-1)] T
其中sk为sk(n)的矢量表示;DT k,1表示第k个子带第1个分段的数据信号的转置;DT k,2表示第k个子带第1个分段的数据信号的转置;DT k,P表示第k个子带第P个分段的数据信号的转置;Dk,p(0)表示第k个子带第p个分段的数据信号里的第1个数据信号点;Dk,p(1)表示第k个子带第p个分段的数据信号里的第2个数据信号点;Dk,p(Q-1)表示第k个子带第p个分段的数据信号里的第Q个数据信号点;表示第k个子带第1个分段的数据信号进行预编码后得到的数据的转置;表示第k个子带第1个分段的数据信号进行预编码后得到的数据的转置;表示第k个子带第P个分段的数据信号进行预编码后得到的数据的转置;where sk is the vector representation of sk (n); D T k,1 represents the transposition of the data signal of the first segment of the kth subband; D T k,2 represents the first segment of the kth subband The transposition of the data signal; D T k,P represents the transposition of the data signal of the p-th segment of the k-th subband; D k,p (0) represents the data signal of the p-th segment of the k-th subband. the first data signal point of the Qth data signal point in the data signal with the pth segment; represents the transposition of the data obtained after precoding the data signal of the first segment of the kth subband; represents the transposition of the data obtained after precoding the data signal of the first segment of the kth subband; represents the transposition of the data obtained after precoding the data signal of the p-th segment of the k-th subband;
预编码矩阵为Q阶的加权分数傅立叶变换矩阵αk为子带k的加权分数傅里叶变换的调制阶数;The precoding matrix is a weighted fractional Fourier transform matrix of order Q α k is the modulation order of the weighted fractional Fourier transform of subband k;
所述矩阵的表达式具体为:the matrix The expression is specifically:
其中是加权系数,定义如下:in is the weighting coefficient, defined as follows:
其中IQ是Q×Q单位矩阵,FQ是Q×Q离散傅里叶变换矩阵;TQ是置换矩阵,每一行每一列只有一个元素非零,具体表示如下:where I Q is a Q×Q unit matrix, F Q is a Q×Q discrete Fourier transform matrix; T Q is a permutation matrix, each row and each column has only one non-zero element, which is specifically expressed as follows:
另外,加权分数傅里叶逆变换可以表示为即为对应的逆预编码矩阵。αk的小标k用于区分不同子带的加权傅里叶变换调制阶数α。并且,根据文献(Mei L,ShaX J,Ran Q W,et al.Research on the application of 4-weighted fractionalFourier transform in communication system[J].Science China(InformationSciences),2010,53(6):1251-1260),经由WFRFT预编码的多载波调制过程可视为混合载波系统。由上述的公式可知,WFTFR变换是由原函数和其傅里叶变换后的函数加权求和得到的。同时,随着α在[0,1]内变化,其单载波分量和多载波分量的比重在发生变化。当α=0时,只有单载波分量,混合载波系统变成单载波系统;当α=1时,只有多载波分量,混合载波系统变成OFDM系统。可将调制阶数α限定在[0,1]内,通过自由选择α,来调整单载波和多载波的比重,以发挥其对抗信道畸变的优势,这便是混合载波调制的机理。In addition, the weighted inverse fractional Fourier transform can be expressed as which is for The corresponding inverse precoding matrix. The subscript k of α k is used to distinguish the weighted Fourier transform modulation order α of different subbands. And, according to the literature (Mei L, ShaX J, Ran QW, et al. Research on the application of 4-weighted fractional Fourier transform in communication system [J]. Science China (Information Sciences), 2010, 53(6): 1251-1260 ), the multi-carrier modulation process via WFRFT precoding can be regarded as a mixed-carrier system. It can be seen from the above formula that the WFTFR transform is obtained by the weighted summation of the original function and its Fourier transformed function. At the same time, as α changes within [0,1], the proportions of its single-carrier components and multi-carrier components are changing. When α=0, there is only a single-carrier component, and the mixed-carrier system becomes a single-carrier system; when α=1, there is only a multi-carrier component, and the mixed-carrier system becomes an OFDM system. The modulation order α can be limited within [0,1], and the proportion of single carrier and multi-carrier can be adjusted by freely selecting α, so as to exert its advantages against channel distortion, which is the mechanism of hybrid carrier modulation.
其它步骤及参数与具体实施方式一相同。Other steps and parameters are the same as in the first embodiment.
具体实施方式三:本实施方式与具体实施方式一或二不同的是:如图10所示,所述步骤二中取出三个连续不重叠分块信号sk,q-1,sk,q,sk,q+1的中间的Nk,b个符号,得到重叠分块信号的表达式为:Embodiment 3: The difference between this embodiment and
其中,Rb为发送机重叠分块矩阵, 为Nk,b阶的单位矩阵,记2Lk,s为相邻重叠分块信号间的重叠样本数,则Nk,b=Lk,b+2Lk,s;本发明取Lk,s=Lk,b/2,即Nk,b=2Lk,b;值得指出的是,2Lk,s的长度可结合滤波器的时域响应长度进行设计,非本发明重点,未予讨论。Among them, R b is the overlapping block matrix of the transmitter, is a unit matrix of order N k,b , and 2L k,s is the number of overlapping samples between adjacent overlapping block signals, then N k,b =L k,b +2L k,s ; the present invention takes L k, s =L k,b /2, that is, N k,b =2L k,b ; it is worth noting that the length of 2L k,s can be designed in combination with the time domain response length of the filter, which is not the focus of the present invention and is not given. discuss.
其它步骤及参数与具体实施方式一或二相同。Other steps and parameters are the same as in the first or second embodiment.
具体实施方式四:本实施方式与具体实施方式一至三之一不同的是:所述步骤四中将傅里叶变换结果直接复制Mk次并首尾拼接形成N点的数据,与该子带对应的N点的频域滤波器的系数ek进行点乘,完成频域滤波,得到各子带频域滤波后的结果Pk的具体过程为:Embodiment 4: This embodiment is different from one of
其中,为Nk,b点傅里叶变换矩阵, 表示增益修正后的滤波器系数矩阵,Bk是第k个子带滤波器的3dB带宽;Mk表示插值倍数,对于所有子带,有Lk,bMk=N。Λk表示以第k个子带对应的插值滤波器频域响应为对角线元素的N×N矩阵,Λk=diag(ek),diag(·)表示以括号内的数作为对角线元素生成对角矩阵; 表示Mk行1列的全1矩阵,表示Nk,b阶的单位阵,表示克罗内可(Kronecker)积。in, is the Fourier transform matrix of N k,b points, represents the gain-modified filter coefficient matrix, B k is the 3dB bandwidth of the kth subband filter; M k represents the interpolation multiple, and for all subbands, L k,b M k =N. Λ k represents an N×N matrix with the frequency domain response of the interpolation filter corresponding to the kth subband as the diagonal elements, Λ k =diag(e k ), diag( ) represents the number in the brackets as the diagonal Elements generate a diagonal matrix; represents an all-one matrix with M k rows and 1 column, represents the identity matrix of order N k,b , Represents the Kronecker product.
各子带使用快速卷积在频域实现了滤波。因此,每个子带滤波器只需在频域对应的抽头上设计不同的权重系数。滤波器可选用根升余弦滤波器频域系数进行设计,通带内的最大权重设为1。若要调整该子带的中心频率,可将(1+β)Nk,b的非零值在N点的FFT单元上移动,最小的移动距离对应的频率变化(也即频率分辨率)Δf=B/N,B为系统总带宽。相应的,接收端对应的匹配滤波器做同样的移动即可。Each subband implements filtering in the frequency domain using fast convolution. Therefore, each subband filter only needs to design different weight coefficients on the corresponding taps in the frequency domain. The filter can be designed with the frequency domain coefficients of the root raised cosine filter, and the maximum weight in the passband is set to 1. To adjust the center frequency of the subband, move the non-zero value of (1+β)N k,b on the FFT unit at point N, and the minimum moving distance corresponds to the frequency change (ie frequency resolution) Δf =B/N, where B is the total bandwidth of the system. Correspondingly, the matched filter corresponding to the receiving end can do the same movement.
所述频域滤波器的系数ek的非零部分根据根升余弦滤波器的归一化的频域公式H(f)直接采样生成:The non-zero part of the coefficient ek of the frequency domain filter is directly sampled and generated according to the normalized frequency domain formula H(f) of the root raised cosine filter:
所述频域滤波器的3dB带宽为Nk,b点,ek的总点数为N点,非零点数为(1+β)Nk,b点。其中β为滚降系数,f0=Δf·Nk,b/2为截止频率,Δf为频率分辨率,f为频率,对f进行离散采样,得到滤波器系数的非零值。The 3dB bandwidth of the frequency domain filter is N k,b point, the total number of e k points is N point, and the number of non-zero points is (1+β)N k,b point. where β is the roll-off coefficient, f 0 =Δf·N k, b /2 is the cutoff frequency, Δf is the frequency resolution, f is the frequency, and f is discretely sampled, Get the non-zero values of the filter coefficients.
其它步骤及参数与具体实施方式一至三之一相同。Other steps and parameters are the same as one of the first to third embodiments.
具体实施方式五:本实施方式与具体实施方式一至四之一不同的是:所述步骤五中将步骤四得到的各子带频域滤波的结果加和后进行N点的傅里叶反变换,得到具体为:Embodiment 5: This embodiment differs from one of
其中,为FN共轭转置矩阵,FN为N点傅里叶变换矩阵,[FN]p,d=N-1/2e-j2π(p-1)(d-1)/N。in, is F N conjugate transpose matrix, F N is N-point Fourier transform matrix, [F N ] p,d =N -1/2 e -j2π(p-1)(d-1)/N .
其它步骤及参数与具体实施方式一至四之一相同。Other steps and parameters are the same as one of the first to fourth embodiments.
具体实施方式六:本实施方式与具体实施方式一至五之一不同的是:所述步骤六中分块信号yq的表达式为:Embodiment 6: This embodiment differs from one of
其中,O为发送机重叠保留矩阵,O=[0L×L/2 IL 0L×L/2],IL为L阶的单位矩阵。Wherein, O is the overlapping reservation matrix of the transmitter, O=[0 L×L/2 I L 0 L×L/2 ], and I L is the L-order identity matrix.
其它步骤及参数与具体实施方式一至五之一相同。Other steps and parameters are the same as one of the specific embodiments one to five.
具体实施方式七:本实施方式与具体实施方式一至六之一不同的是:所述步骤七中重叠分块信号的表达式为:Embodiment 7: The difference between this embodiment and one of
其中,R为接收机重叠分块矩阵,R=[0N×(L-L/2) IN 0N×(L-L/2)],IN为N阶的单位矩阵。Wherein, R is the receiver overlapping block matrix, R=[0 N×(LL/2) I N 0 N×(LL/2) ], and I N is an N-order identity matrix.
其它步骤及参数与具体实施方式一至六之一相同。Other steps and parameters are the same as one of
具体实施方式八:本实施方式与具体实施方式一至七之一不同的是:所述步骤九中将步骤八进行傅里叶变换后的分块信号进行均衡和子带滤波的具体过程为:Embodiment 8: This embodiment differs from
先根据不同的均衡法则得到均衡系数矩阵Φ后,将均衡系数矩阵Φ与不同子带的滤波器系数矩阵Πk进行相乘合并后,再与N点的傅里叶变换结果进行点乘;将结果按长度Nk,b均分成Mk段,进行叠加,完成降采样滤波,得到第k子带第q个分段的降采样滤波后的数据Qk,q;所述不同的均衡准则为迫零准则和最小均方误差准则;After obtaining the equalization coefficient matrix Φ according to different equalization rules, the equalization coefficient matrix Φ and the filter coefficient matrix Π k of different subbands are multiplied and merged, and then point multiplied with the Fourier transform result of N points; The result is divided into Mk sections by length Nk,b , superimposes, completes downsampling filtering, obtains the data Qk,q after the downsampling filtering of the kth subband qth subsection; Described different equalization criteria are Zero forcing criterion and minimum mean square error criterion;
这里,ET为E的转置,由于本发明采用成对的采样滤波器和插值滤波器,故接收端的增益修正后的滤波器系数矩阵Πk与发射机相同。Φ为频域均衡系数矩阵,其可根据ZF准则或MMSE准则得到;Here, E T is the transposition of E, because the present invention adopts paired sampling filters and interpolation filters, the filter coefficient matrix Π k after the gain correction of the receiving end is the same as that of the transmitter. Φ is the frequency domain equalization coefficient matrix, which can be obtained according to the ZF criterion or the MMSE criterion;
所述均衡系数矩阵Φ具体为:The equalization coefficient matrix Φ is specifically:
当采用最小均方误差准则时,均衡系数矩阵Φ为:When the minimum mean square error criterion is adopted, the equalization coefficient matrix Φ is:
当采用迫零准则时,均衡系数矩阵Φ为:When the zero-forcing criterion is adopted, the equalization coefficient matrix Φ is:
Φ=diag(1/hi),i=1,2,…,NΦ=diag(1/h i ), i=1,2,...,N
其中,hi为的主对角线元素,HN×N为多径信道矩阵,FN为N维的傅里叶变换矩阵,为FN的共轭转置矩阵,σ2为噪声方差,为hi的共轭。Among them, hi is The main diagonal elements of , H N×N is the multipath channel matrix, F N is the N-dimensional Fourier transform matrix, is the conjugate transpose matrix of F N , σ 2 is the noise variance, is the conjugate of hi .
在接收端,本分明采用了快速卷积的方式完成采样滤波。将均衡过程加入,其实现框图如图2所示。图2中,滤波器的阻带部分值为0,与均衡系数进行相乘合并后,依旧为0,图中没有标注。At the receiving end, the present invention adopts the fast convolution method to complete the sampling filtering. The equalization process is added, and its implementation block diagram is shown in Figure 2. In Figure 2, the value of the stopband part of the filter is 0, and after multiplication and merging with the equalization coefficient, it is still 0, which is not marked in the figure.
可以发现,这里使用的是简单的单点抽头均衡,均衡器系数可与滤波器系数相乘合并。这一简化的单抽头均衡器的设计理念的可行性在文献(Zhao J,Wang W,GaoX.Transceiver design for fast-convolution multicarrier systems in multipathfading channels[C]//International Conference on Wireless Communications&Signal Processing.IEEE,2015:1-5)进行了说明。本发明结合不同的加权变换调制阶数,在不同均衡准则下,予以了仿真验证。为了尽量减小段间干扰,各分段之间的重叠部分应显著大于信道长度。同时,多径信道矩阵HN×N会被傅里叶变换矩阵渐进对角化,对角化程度随着傅里叶变换点数N的增大而增大。也即,N越大,取主角线元素hi生成的均衡系数对信道畸变的补偿效果更佳。It can be found that a simple one-point tap equalization is used here, and the equalizer coefficients can be multiplied and combined with the filter coefficients. The feasibility of this simplified single-tap equalizer design concept is presented in the literature (Zhao J, Wang W, GaoX. Transceiver design for fast-convolution multicarrier systems in multipathfading channels [C]//International Conference on Wireless Communications&Signal Processing. IEEE, 2015:1-5) were described. The present invention has been simulated and verified under different equalization criteria in combination with different weighted transformation modulation orders. To minimize inter-segment interference, the overlap between segments should be significantly larger than the channel length. At the same time, the multipath channel matrix H N×N will be progressively diagonalized by the Fourier transform matrix, and the degree of diagonalization increases as the number of Fourier transform points N increases. That is, the larger N is, the better the compensation effect of channel distortion is by the equalization coefficient generated by taking the main line element hi.
其它步骤及参数与具体实施方式一至七之一相同。Other steps and parameters are the same as one of the first to seventh embodiments.
具体实施方式九:本实施方式与具体实施方式一至八之一不同的是:所述步骤十中将步骤九进行均衡和子带滤波后的数据进行Nk,b点傅里叶逆变换,得到的表达式为:Embodiment 9: The difference between this embodiment and one of
其中为Nk,b点傅里叶逆变换矩阵。in is the inverse Fourier transform matrix of N k,b points.
其它步骤及参数与具体实施方式一至八之一相同。Other steps and parameters are the same as one of
具体实施方式十:本实施方式与具体实施方式一至九之一不同的是:所述步骤十一中保留各分块的中间Lk,b个数据xk,q,组成数据流xk;再按长度Q分成P段,每段记为各段分别进行逆预编码变换,得到再将逆预编码后的P个结果组合得到各子带的接收信号具体为:Embodiment 10: The difference between this embodiment and one of
其中,Ob为接收机重叠保留矩阵, 为Lk,b阶的单位矩阵。xT k,1表示第k个子带第1个不重叠分块数据信号的转置;xT k,2表示第k个子带第2个不重叠分块数据信号的转置;表示第k个子带第个不重叠分块数据信号的转置;数据流xk由个不重叠分块数据xk,q拼接而成,再将其按长度Q分成成P段,每段数据信号记为 表示第k个子带第1个分段的数据信号的转置;表示第k个子带第2个分段的数据信号的转置;表示第k个子带第P个分段的数据信号的转置;表示第k个子带第1个分段的数据逆预编码变换后得到的结果的转置;表示第k个子带第2个分段的数据逆预编码变换后得到的结果的转置;表示第k个子带第P个分段的数据逆预编码变换后得到的结果的转置;where O b is the receiver overlap retention matrix, is the identity matrix of order L k,b . x T k,1 represents the transposition of the first non-overlapping block data signal of the kth subband; x T k,2 represents the transposition of the second non-overlapping block data signal of the kth subband; represents the kth subband The transpose of a non-overlapping block data signal; the data stream x k is given by The non-overlapping block data x k, q are spliced together, and then divided into P segments according to the length Q, and each segment of data signal is recorded as represents the transposition of the data signal of the first segment of the kth subband; represents the transposition of the data signal of the second segment of the kth subband; represents the transpose of the data signal of the p-th segment of the k-th subband; represents the transposition of the result obtained after the inverse precoding transformation of the data of the first segment of the kth subband; represents the transposition of the result obtained after the data inverse precoding transformation of the second segment of the kth subband; represents the transposition of the result obtained after the inverse precoding transformation of the data of the p-th segment of the k-th subband;
其它步骤及参数与具体实施方式一至九之一相同。Other steps and parameters are the same as one of
实施例一:Example 1:
本发明提出的系统相对于传统的混合载波系统最显著的优点便是陡峭的带外衰减。带外泄露会带来子带间的信号干扰,因而本发明提出的系统带外衰减快,能更好地利用碎片化频谱资源间的“空隙”。The most significant advantage of the system proposed by the present invention over the traditional hybrid carrier system is the steep out-of-band attenuation. Out-of-band leakage will bring about signal interference between sub-bands, so the system proposed in the present invention has fast out-of-band attenuation, and can better utilize the "gap" between fragmented spectrum resources.
基本仿真参数设置为:加权变换的调制阶数α=0.5,N=2048,L=1024,根升余弦滤波器滚降系数β=0.125,连续流的分段进行加权变换预编码的长度Q=64,重叠分块长度Nk,b=128,仿真中初始数据信号均采用4QAM调制生成,均未使用信道编码。仿真中各子带使用了相同的重叠分块长度与相同的加权变换调制阶数。图3比较了本发明提出的系统的某个子带使用根升余弦滤波器后与传统混合载波系统不使用根升余弦滤波器进行滤波两种情况下的功率谱。仿真中,使用的根升余弦滤波器的滚降因子β=0.125。从图3中可以看出,本发明提出的系统能较好地抑制旁瓣。同时,在相同的频谱效率下,将本发明所提系统与传统混合载波系统已调信号的功率谱进行比较。两系统实际使用的载波数均为1024。结果如图4所示,相邻子带间的干扰在-30dB以下,子带间干扰较小。需要指出的是,本发明系统中,各子带带宽和中心频率可自由调整。并且,可以改变滤波器的滚降系数改变带外抑制性能。The basic simulation parameters are set as: the modulation order of the weighted transform α=0.5, N=2048, L=1024, the root raised cosine filter roll-off coefficient β=0.125, the length of the weighted transform precoding Q= 64. The overlapping block length N k,b =128. In the simulation, the initial data signals are all generated by 4QAM modulation, and channel coding is not used. In the simulation, each subband uses the same overlapping block length and the same weighted transform modulation order. FIG. 3 compares the power spectrum of a certain subband of the system proposed by the present invention using the root raised cosine filter and the traditional hybrid carrier system not using the root raised cosine filter for filtering. In the simulation, a root raised cosine filter was used with a roll-off factor β=0.125. It can be seen from FIG. 3 that the system proposed by the present invention can better suppress the side lobes. Meanwhile, under the same spectral efficiency, the power spectrum of the modulated signal of the system proposed by the present invention is compared with that of the traditional hybrid carrier system. The actual number of carriers used by the two systems is 1024. The results are shown in Figure 4, the interference between adjacent subbands is below -30dB, and the interference between subbands is small. It should be pointed out that, in the system of the present invention, the bandwidth and center frequency of each subband can be adjusted freely. Also, the roll-off factor of the filter can be changed to change the out-of-band rejection performance.
峰值-平均功率比(PAPR)定义为信号的最大瞬时功率与平均功率之比:Peak-to-average power ratio (PAPR) is defined as the ratio of the maximum instantaneous power to the average power of a signal:
其中,s(n)表示发射的时域信号,E[·]表示期望值。无线通信系统发射机的功率放大器存在着最大功率限制。为了确保信号经过功率放大器之后不发生非线性失真,要求功率放大器工作在线性工作区内,即发射机信号的最大瞬时功率一般不能超过功率放大器的最大输出功率。信号的高PAPR会损害系统的通信性能。互补累计分布函数(CCDF)被用来评估系统的PAPR性能,其定义为信号实际峰均功率比超过门限峰均功率比PAPR0的概率:Among them, s(n) represents the transmitted time-domain signal, and E[·] represents the expected value. The power amplifier of the wireless communication system transmitter has a maximum power limit. In order to ensure that no nonlinear distortion occurs after the signal passes through the power amplifier, the power amplifier is required to work in the linear working region, that is, the maximum instantaneous power of the transmitter signal generally cannot exceed the maximum output power of the power amplifier. The high PAPR of the signal can impair the communication performance of the system. The complementary cumulative distribution function (CCDF) is used to evaluate the PAPR performance of the system, which is defined as the probability that the actual peak-to-average power ratio of the signal exceeds the threshold peak-to-average power ratio PAPR 0 :
CCDF=Pr[PAPR>PAPR0]CCDF=Pr[PAPR>PAPR 0 ]
其中,Pr[·]表示概率。我们知道,对于混合载波系统,其PAPR值介于OFDM系统与DFT-S-OFDM之间(沙学军,梅林,张钦宇.加权分数傅里叶变换及其在通信系统中的应用[M].人民邮电出版社,2016:81-83)。而OFDM系统与DFT-S-OFDM系统是特定加权变换调制阶数下的混合载波系统。而本发明所提的基于快速卷积的滤波混合载波系统不仅保留了传统混合载波系统根据不同调制阶数对PAPR进行调整的灵活性,同时,滤波的操作可在一定程度上进一步减小PAPR。Among them, Pr[·] represents the probability. We know that for mixed carrier system, its PAPR value is between OFDM system and DFT-S-OFDM (Sha Xuejun, Mei Lin, Zhang Qinyu. Weighted Fractional Fourier Transform and Its Application in Communication System [M]. People's Posts and Telecommunications Press, 2016: 81-83). The OFDM system and the DFT-S-OFDM system are hybrid carrier systems under a specific weighted transform modulation order. The fast convolution-based filtering hybrid carrier system proposed in the present invention not only retains the flexibility of the traditional hybrid carrier system to adjust the PAPR according to different modulation orders, but also the filtering operation can further reduce the PAPR to a certain extent.
图5以单一子带为例,对本发明系统的PAPR特性与传统混合载波系统进行了对比。仿真中,基本仿真参数同上,采用4QAM调制方式,本发明系统子带3dB带宽对应的子载波占用数为128(Nk,b=128),其余系统单一子带均占用128个子载波。由图可见,当单个子带用于一个用户传输数据时,本发明系统发射端信号的PAPR明显小于传统混合载波系统的PAPR。其中,OFDM可视为α=1的混合载波系统,DFT-S-OFDM可视为α=0的混合载波系统。在相同的调制阶数下,本发明系统PAPR性能优于传统混合载波系统。并且,对于本发明系统,当发送的原始数据为时域信号,即α=0时,其具有最佳的PAPR性能。FIG. 5 takes a single subband as an example to compare the PAPR characteristics of the system of the present invention with the traditional hybrid carrier system. In the simulation, the basic simulation parameters are the same as above, and the 4QAM modulation mode is adopted. The occupied number of subcarriers corresponding to the 3dB bandwidth of the subband of the system of the present invention is 128 (N k,b =128), and the other single subbands of the system occupy 128 subcarriers. As can be seen from the figure, when a single subband is used for data transmission by one user, the PAPR of the signal at the transmitter of the system of the present invention is significantly smaller than that of the traditional hybrid carrier system. Among them, OFDM can be regarded as a mixed carrier system with α=1, and DFT-S-OFDM can be regarded as a mixed carrier system with α=0. Under the same modulation order, the PAPR performance of the system of the present invention is better than that of the traditional hybrid carrier system. Moreover, for the system of the present invention, when the transmitted original data is a time domain signal, that is, when α=0, it has the best PAPR performance.
实施例二:Embodiment 2:
该实施例对本发明方法的误比特率性能进行了仿真。This embodiment simulates the bit error rate performance of the method of the present invention.
基于前述对本发明系统接收机的介绍,结合系统特点,在接收端采用基于快速卷积的均衡器设计。该接收机设计,不再要求所发送信号带有循环前缀CP。本发明系统不依赖CP结构,提高了频谱效率,同时,在频选信道下,可达到和原混合载波系统接近的误比特率性能。本发明提出,可根据均衡所采用的基于ZF或MMSE均衡准则,与系统对PAPR的要求,在发端选取合适的调制阶数。需要注意的是,每一子带可根据各自不同的需求选取不同的调制阶数α。而本发明的仿真以单一子带进行说明。如果是多子带系统,可将各子带的误码率进行平均处理。同时,下述仿真中,连续流分段进行预编码的分段长度Q都设为64。Based on the foregoing introduction to the system receiver of the present invention, combined with the characteristics of the system, an equalizer design based on fast convolution is adopted at the receiving end. This receiver design no longer requires the transmitted signal to carry the cyclic prefix CP. The system of the present invention does not depend on the CP structure, thereby improving the spectrum efficiency, and at the same time, under the frequency selection channel, the bit error rate performance close to that of the original mixed carrier system can be achieved. The present invention proposes that an appropriate modulation order can be selected at the transmitting end according to the ZF or MMSE-based equalization criterion adopted for equalization and the requirements of the system for PAPR. It should be noted that, each subband can select different modulation order α according to different requirements. In contrast, the simulation of the present invention is illustrated with a single subband. If it is a multi-subband system, the bit error rate of each subband can be averaged. Meanwhile, in the following simulation, the segment length Q for precoding of continuous stream segments is set to 64.
在固定频选信道下,对本发明系统进行了仿真。仿真使用三径信道,信道延迟[0,5,10]个码片,信道冲激响应为[1,-0.5,0.3]并做能量归一化处理。基本参数设置为:加权变换的调制阶数α=0,0.5,1,N=2048,L=1024,RRC滤波器滚降系数β=0.125,单子带占用的3dB带宽为128个子载波,即Nk,b=128。初始连续流信号均采用4QAM调制生成,均未使用信道编码。接收端均假设具有理想的信道状态信息。Under the fixed frequency selection channel, the system of the present invention is simulated. The simulation uses a three-path channel, the channel delay is [0, 5, 10] chips, the channel impulse response is [1, -0.5, 0.3] and the energy is normalized. The basic parameters are set as: the modulation order of the weighted transformation α=0, 0.5, 1, N=2048, L=1024, the RRC filter roll-off coefficient β=0.125, and the 3dB bandwidth occupied by a single subband is 128 subcarriers, namely N k,b =128. The initial continuous stream signals are all generated using 4QAM modulation, and none of them use channel coding. The receivers are assumed to have ideal channel state information.
从图6与图7可见,对于本发明所提系统,当子带所发送数据所含的单载波成分越高,该子带越类似宽带单载波系统,能更好的利用频率分集,误比特率性能更佳。It can be seen from Fig. 6 and Fig. 7 that for the system proposed in the present invention, when the single-carrier component contained in the data transmitted by the sub-band is higher, the sub-band is more similar to the broadband single-carrier system, and the frequency diversity can be better utilized, and the bit error rate is reduced. better rate performance.
在随机频选信道下,对本发明系统的性能进行了蒙特卡罗仿真。各径的相对时延为[0,10,20,30,50,70]个码片,各径的平均功率为[0,-3.6,-7.2,-10.8,-18,-25.2]dB。基本参数设置为:加权变换的调制阶数α=0,0.5,1,N=2048,L=1024,RRC滤波器滚降系数β=0.125,单子带占用的3dB带宽为128个子载波,即Nk,b=128。初始连续流信号均采用4QAM调制生成,均未使用信道编码。为了体现衰落信道的随机性,生成了1000次随机信道,每次信道仿真1000次,并取平均。Under the random frequency selection channel, Monte Carlo simulation of the performance of the system of the present invention is carried out. The relative time delay of each path is [0, 10, 20, 30, 50, 70] chips, and the average power of each path is [0, -3.6, -7.2, -10.8, -18, -25.2] dB. The basic parameters are set as: the modulation order of the weighted transformation α=0, 0.5, 1, N=2048, L=1024, the RRC filter roll-off coefficient β=0.125, and the 3dB bandwidth occupied by a single subband is 128 subcarriers, namely N k,b =128. The initial continuous stream signals are all generated using 4QAM modulation, and none of them use channel coding. In order to reflect the randomness of the fading channel, 1000 random channels are generated, each channel is simulated 1000 times, and the average is taken.
由图8与图9可见,在随机频选信道下,本发明提出的基于快速卷积滤波混合载波系统在不同的均衡法则下,不同的加权调制阶数α预编码下,呈现不同的误比特率性能。在ZF准则下,子带发送多载波数据(即α=1)时,具有最好的误比特率性能。在MMSE准则下,子带发送单载波数据(即α=0)时,具有最好的误比特率性能。故而,针对本发明系统,如果只考虑误比特率性能时,接收机采用ZF准则进行均衡时,可选取预编码矩阵的调制阶数α=1;如果接收机采用MMSE准则进行均衡时,选取预编码矩阵的调制阶数α=0。同时,调制阶数也会影响PAPR的大小。当系统单载波成分越多,即α接近0时,PAPR越小。故,可结合系统对PAPR的要求,兼顾接收机采用的均衡法则,选取合适的调制阶数α。It can be seen from FIG. 8 and FIG. 9 that under the random frequency selection channel, the hybrid carrier system based on fast convolution filtering proposed by the present invention presents different bit errors under different equalization rules and different weighted modulation order α precoding. rate performance. Under the ZF criterion, when the subband transmits multi-carrier data (ie α=1), it has the best bit error rate performance. Under the MMSE criterion, when the subband transmits single-carrier data (ie, α=0), it has the best bit error rate performance. Therefore, for the system of the present invention, if only the bit error rate performance is considered, when the receiver adopts the ZF criterion for equalization, the modulation order α of the precoding matrix can be selected as α=1; if the receiver adopts the MMSE criterion for equalization, the precoding matrix is selected. The modulation order α=0 of the coding matrix. At the same time, the modulation order will also affect the size of the PAPR. When there are more single-carrier components in the system, that is, when α is close to 0, the PAPR is smaller. Therefore, the appropriate modulation order α can be selected according to the requirements of the system for PAPR and the equalization rule adopted by the receiver.
本发明还可有其它多种实施例,在不背离本发明精神及其实质的情况下,本领域技术人员当可根据本发明作出各种相应的改变和变形,但这些相应的改变和变形都应属于本发明所附的权利要求的保护范围。The present invention can also have other various embodiments. Without departing from the spirit and essence of the present invention, those skilled in the art can make various corresponding changes and deformations according to the present invention, but these corresponding changes and deformations are all It should belong to the protection scope of the appended claims of the present invention.
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Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
CN103188053A (en) * | 2013-04-03 | 2013-07-03 | 电子科技大学 | Signal detection method for space time block code (STBC)-orthogonal frequency division multiplexing (OFDM) system under condition of lack of cyclic prefix |
WO2013148076A1 (en) * | 2012-03-28 | 2013-10-03 | Qualcomm Incorporated | Extending cyclic prefix length in lte with mixed unicast broadcast subframe |
CN104092641A (en) * | 2014-07-17 | 2014-10-08 | 哈尔滨工业大学 | A Mixed Carrier Communication Method Based on Signal Probability Density Selection of ADC Optimal Scale Factor |
CN106301691A (en) * | 2016-11-04 | 2017-01-04 | 中国电子科技集团公司第五十四研究所 | Low density parity check code disturbance restraining method based on transform domain |
CN107707501A (en) * | 2017-10-13 | 2018-02-16 | 哈尔滨工业大学 | Based on time-interleaved more vector WFRFT mixed carrier parallel transmission methods |
-
2018
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Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
WO2013148076A1 (en) * | 2012-03-28 | 2013-10-03 | Qualcomm Incorporated | Extending cyclic prefix length in lte with mixed unicast broadcast subframe |
CN103188053A (en) * | 2013-04-03 | 2013-07-03 | 电子科技大学 | Signal detection method for space time block code (STBC)-orthogonal frequency division multiplexing (OFDM) system under condition of lack of cyclic prefix |
CN104092641A (en) * | 2014-07-17 | 2014-10-08 | 哈尔滨工业大学 | A Mixed Carrier Communication Method Based on Signal Probability Density Selection of ADC Optimal Scale Factor |
CN106301691A (en) * | 2016-11-04 | 2017-01-04 | 中国电子科技集团公司第五十四研究所 | Low density parity check code disturbance restraining method based on transform domain |
CN107707501A (en) * | 2017-10-13 | 2018-02-16 | 哈尔滨工业大学 | Based on time-interleaved more vector WFRFT mixed carrier parallel transmission methods |
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