CN108781193B - Method for generating pulse waveforms with adjustable length, orthogonality and localization properties - Google Patents

Abstract

An embodiment of the present invention provides a method for generating a pulse waveform, comprising the steps of: obtaining a pulse waveform g (t); performing optimization on the obtained pulse waveform g (t); generating an optimized pulse waveform by truncating the optimized pulse waveform; determining whether the generated pulse waveform meets at least one predetermined requirement, outputting the generated pulse waveform if it meets the at least one predetermined requirement, otherwise repeating the optimizing method steps and the truncating method steps by using the generated pulse waveform as the obtained pulse waveform. Another embodiment relates to a pulse waveform generated by the disclosed method. Another embodiment relates to a multi-carrier modulation system configured to perform the disclosed method. Another embodiment relates to the use of the generated pulse waveform. Another embodiment relates to a computer program performing the disclosed method. Another embodiment relates to a system comprising a multi-carrier modulation system and a communication device, wherein the multi-carrier modulation system is configured to update an obtained pulse shape g (t) by performing communication between the multi-carrier modulation system and the communication device according to a communication protocol.

Description

Method for generating pulse waveforms with adjustable length, orthogonality and localization properties

Technical Field

The present invention relates to a method for generating a pulse waveform, a pulse waveform generated by performing the method, a multi-carrier modulation system configured to perform the method, the use of a pulse waveform in a multi-carrier modulation system, a computer program for performing the method and a system comprising a multi-carrier modulation system and a communication device. In particular, aspects of the present invention relate to wireless communications based on a Filter Bank Multi Carrier (FBMC) system, particularly for QAM symbol transmission. Furthermore, the present invention is applicable to forward compatible 5G air interfaces that focus on supporting enhanced Mobile broadband (eMBB), Massive Machine-type Communication (MMC), and Ultra-Reliable and low-latency Communication (URC). The present invention can be used for Multiple Input Multiple Output (MIMO) channels.

Background

In the real world, a physical channel is exposed to two kinds of dispersion, namely, time dispersion due to multipath propagation and frequency dispersion due to doppler shift. With respect to time dispersion, due to multipath propagation, when a signal, whether it has the form of an electromagnetic wave or an acoustic wave, is transmitted from a transmitter to a receiver in a wireless communication system, it may interact with objects in the environment. Due to such interactions, the receiver may receive multiple copies of the original signal, each with its own propagation delay and attenuation factor. This situation is known as multipath propagation. Multipath propagation may result in Inter-Symbol Interference (ISI) in the time domain, which is related to frequency selectivity in the frequency domain. Further, regarding frequency dispersion, due to doppler shift, when the relative distance between the transmitter, the receiver, and the interfering object changes, the frequency of the wave may be shifted. This effect is known as doppler shift, which can lead to Inter-Carrier Interference (ICI).

In fact, in the real world, if a signal propagates within a linear time-variable (LTV) channel h (t, τ), the transmitted signal s (t) suffers from both multipath propagation and doppler shift. The generalized operators are denoted as generalized operators throughout this patent application

And neglecting the additive noise, the received signal r (t) is

If it is not

Modeled as a broad-sense stationary with undercorrelated scattering (WSSUS), the corresponding channel scattering function C_H(τ, ν) can be given by:

in this patent application, system causality is ignored in the rest of the application in order to simplify the analysis.

Furthermore, the basic idea of multi-carrier (MC) modulation is shown in fig. 1, where the main idea is shown to divide a wideband channel with high frequency selectivity into M narrowband subchannels with almost flat frequency response. Further, g (t) and γ (t) represent a transmission prototype filter and a reception prototype filter, respectively. To model the transmitted signal s (T), the QAM symbol sampling grid is first defined as (T, F) with a density condition TF ≧ 1, where T denotes the symbol duration of the signal and F denotes the intercarrier spacing. Then, s (t) is given by:

wherein, g_m，n(t) is defined as g_m，n(t)＝g(t-nT)e^j2πmF(t-nT)The transmit filter bank of (1).

At the receiver side, by combining the received signal r (t) with the receive filter bank γ_m，n(t)＝γ(t-nT)e^j2πmF(t-nT)Performing correlation to obtain demodulated symbols

Namely, it is

g (t) and γ (t) should satisfy two basic properties. The first is orthogonality. For a full reconstruction at the receiver side, the multi-carrier transmission requires g_m，n(t) and γ_m，nOrthogonality between (t) to overcome ISI and ICI. Furthermore, T-F localization should be presented to avoid symbol energy "smearing out" on the channel and spreading over adjacent symbols during double dispersion transmission, so good localization of g (T) and γ (T) is also needed.

In the prior art, two types of MC modulation are used, namely Orthogonal Frequency Division Multiplexing (OFDM) and filterbank Multi-Carrier (FBMC), the main difference being the choice of g (t) and γ (t). Of the OFDM, it is undoubtedly the current most popular MC architecture in which both g (t) and γ (t) are chosen as rectangular pulses. By applying Inverse Fast Fourier Transform (IFFT) and Fast Fourier Transform (FFT), modulation and demodulation can be smoothly performed. In addition, Cyclic Prefix (CP, e.g., CP-OFDM) helps to effectively avoid ISI. Although OFDM provides several advantages, it results in having large sidelobes caused by rectangular pulses in the time domain, which results in poor performance in certain applications, e.g., uplink transmission in a multi-carrier system where subsets of subcarriers are allocated to each user.

Regarding FBMC, to avoid ICI, FBMC is also an active search region. Most design algorithms in the prior art consider a symmetric prototype function, i.e., g (t) ═ γ (t). In FBMC, the duration of g (T) is typically an integer multiple of T, which leads to better T-F localization properties as in OFDM. There are two different kinds of FBMC, FBMC/OQAM and FBMC/QAM, which are designed to transmit real-valued and complex-valued data symbols, respectively. In this patent application, only FBMC/QAM is considered. Furthermore, based on framework theory, "gaussian orthogonalization" is one of the most popular methods. However, for any bandwidth efficiency requirement, such as LTE, the length of the generated pulse may be very long.

As mentioned above, MC modulation schemes as OFDM or FBMC face their own problems. To date, much effort has been expended in both areas. For OFDM, for example, windowed OFDM has been proposed as a variant to optimize the frequency response. However, when the transition periods at the beginning and end are short, the prototype filter still suffers from poor frequency response in order to achieve high bandwidth efficiency. In order to find a satisfactory alternative, it is an object of the invention to design a numerical algorithm that can obtain pulse waveforms with adjustable length, orthogonality and TF localization properties, in particular for FBMC/QAM systems.

In the prior art, within CP-OFDM, a transmitter and a receiver perform 'Add CP' and 'remove CP' operations, respectively, wherein, moreover

This is true. As shown in fig. 2, the transmission prototype filter g_cpofdm(t) and a receiving prototype filter gamma_cpofdm(t) is selected as a rectangular pulse and is given by:

furthermore, CP-OFDM can be modeled as "half-prefix and half-suffix OFDM," but is still referred to as "CP-OFDM," when system causality is ignored. Due to the poor spectral confinement of CP-OFDM, although successful in preventing temporal dispersion up to T_cpBut it is not resistant to ICI.

Furthermore, to handle large sidelobes in CP-OFDM, windowed OFDM was introduced, where the rectangular pulse is replaced by a window function with soft transitions at both ends. In practice, g (t) and γ (t) are typically chosen as root-raised cosine (RRC) pulses as shown in fig. 3. In this context, it should be noted that such an improvement of the frequency response is only possible if the soft transition is relatively long. However, if bandwidth efficiency is considered, a long transition period cannot be selected (see e.g. the left diagram of fig. 3). Thus, the RRC pulse shape may still suffer from poor spectral constraints.

Furthermore, the approach of "orthogonalizing gaussian" as opposed to OFDM and its variants is directly focused on constructing TF localized good pulses as their prototype filters. From having good TF localization properties and linearity independence

Gauss g of_gauss(t) starting, the corresponding quadrature system may be constructed

Here, the number of the first and second electrodes,

is and

dual frame of

An associated frame operator.

For simplicity, it is written as

g_orth＝orth{g_gauss，T，F}。

A numerical solution can be efficiently obtained by matrix factorization. The resulting pulses for TF 1.07 and TF 1.25 are shown in fig. 4. For g_orthOne problem is that for arbitrary bandwidth efficiency requirements, e.g., for LTE with TF-1.07, the length of the generated pulse may be long (see left diagram of fig. 4). Those skilled in the art can directly truncate the pulse to the desired length, but may loseIts orthogonality and TF localization.

Furthermore, a short PR FMT can be used to better balance between TF localization and filter length, where the short pulse waveform satisfies the full reconstruction (PR) condition with relatively good TF localization properties. In this context, two analysis methods for calculating an optimal PR Filtered polyphonic (FMT) prototype filter have been proposed in the prior art. One to minimize Out-of-Band (OOB) energy and the other to maximize TF localization. However, both algorithms are limited by

In which the time is shifted

And the number of sub-channels

Wherein, F_SRepresenting the sampling frequency. In addition, fig. 5 shows g when TF is 1.25_OOB(t) and g_TFL(t)。

Thus, in the prior art, there is no general method for generating orthogonal prototype filters of arbitrary length and good TF localization properties.

Disclosure of Invention

The invention therefore relates in particular to a method for generating pulse waveforms which are well localized in the T-F domain, at least approximately orthogonal and of arbitrary length. It is a further problem to provide a corresponding pulse shape generated by the generation method described herein, a corresponding multi-carrier modulation system configured to perform the claimed method, use of said pulse shape in a multi-carrier modulation system, a computer program for performing a process for implementing the claimed method and a corresponding system comprising a multi-carrier modulation system and a communication device.

These problems are solved by the subject matter of the independent claims. Advantageous implementations of the invention are further defined in the respective dependent claims.

A first aspect of the invention relates to a method for generating a pulse waveform, comprising the steps of: obtaining a pulse waveform g (t); performing optimization on the obtained pulse waveform g (t); generating an optimized pulse waveform by truncating the optimized pulse waveform; determining whether the generated pulse waveform meets at least one predetermined requirement, outputting the generated pulse waveform if it meets the at least one predetermined requirement, otherwise repeating the optimizing method steps and the truncating method steps by using the generated pulse waveform as the obtained pulse waveform.

Thus, according to a first aspect, an iterative method for designing pulses with adjustable length, orthogonality and localization properties is provided. Thus, an at least approximately orthogonal transmit prototype filter and receive prototype filter may be provided that have good T-F localization properties at the same time.

In a first implementation form of the first aspect, the optimization is an orthogonalization and is performed by computing

To perform the orthogonalization of the signal to be processed,

is a dual frame with filter banks

An associated dual Gabor frame operator, where T is the symbol duration of the input signal and F is the inter-carrier spacing.

Furthermore, a specific implementation form for performing the optimization is provided, which is similar to a very easy and efficient method for performing the orthogonalization of a certain obtained pulse waveform.

In a second implementation form of the first aspect, the truncation of the optimized pulse waveform is performed by multiplying the optimized pulse waveform with a truncation window.

This helps to provide an efficient method for generating at least approximately orthogonal pulse waveforms.

In a third implementation form of the first aspect, the truncation window is fixed for the entire method.

In a fourth implementation form of the first aspect, the truncation window is a rectangular window RECT, a raised cosine window RC (β) or a root raised cosine window RRC (β), where β is the roll-off factor and β ≧ 0.

In a fifth implementation form of the first aspect, the truncation window is varied each time the truncation step is performed.

In a sixth implementation form of the first aspect, the obtained pulse shape g (t) is polynomial localized or sub-exponential localized.

In a seventh implementation form of the first aspect, the polynomial localized pulse shape g (t) is a spline-type pulse shape and the sub-exponential localized pulse shape g (t) is a gaussian pulse shape.

In an eighth implementation form of the first aspect, the series of method steps of optimizing and truncating form an iteration, and in a first alternative the at least one predetermined requirement comprises a difference between a pulse waveform generated by the iteration and a pulse waveform generated by a previous iteration being below a threshold, or in a second alternative the at least one predetermined requirement comprises exceeding a maximum number of iterations. Thus, a very efficient and easy criterion for stopping the iterative method is provided.

In a second aspect of the invention, there is provided a pulse waveform generated by performing any of the above methods.

In a third aspect of the invention, a multicarrier modulation system configured to perform any of the above-described methods is provided.

In an implementation form of the third aspect, the multicarrier system is a filter bank.

In a fourth aspect of the invention, the use of a pulse shape according to the above method in a multi-carrier modulation system, in particular in a filter bank.

In a fifth aspect of the invention, there is provided a computer program for executing a process according to any one of the above-mentioned methods.

In a sixth aspect of the present invention, there is provided a system comprising a communication device and a multi-carrier modulation system according to the third aspect or an implementation form of the third aspect, wherein the multi-carrier modulation system is configured to update the obtained pulse waveform g (t) by performing communication between the multi-carrier modulation system and the communication device according to a communication protocol.

In general, it has to be noted that all arrangements, devices, modules, components, models, elements, units, means, etc. described in the present application can be realized by software or hardware elements or any type of combination thereof. All steps performed by the various entities described in the present application, as well as the functions described as being performed by the various entities, are intended to mean that the respective entity is adapted or configured to perform the respective steps and functions. Even if in the following description of specific embodiments specific functions or steps to be performed by a general entity are not reflected in the description of specific detailed elements of the entity performing the specific steps or functions, it should be clear to a person skilled in the art that these methods and functions can be implemented in corresponding hardware or software elements or any type of combination thereof. Furthermore, the method of the present invention and its various steps are embodied in the functionality of the various described apparatus components.

Drawings

The above aspects and implementations of the invention will be explained in the following description of specific embodiments with respect to the drawings, in which:

fig. 1 shows a unified block diagram of an OFDM transceiver and an FBMC transceiver;

fig. 2 shows CP-OFDM prototype filters at the transmitter (solid line) and receiver (dashed line);

fig. 3 shows the function g for TF 1.07 (left diagram) and TF 1.25 (right diagram)_RRc；

Fig. 4 shows g at TF 1.07 (left panel) and TF 1.25 (right panel)_orth；

Fig. 5 shows g when TF 1.25_OOB(left panel) and g_TFL(right panel);

FIG. 6 illustrates an embodiment of an iterative method according to the present invention;

FIG. 7 illustrates an implementation of the iterative method of FIG. 6 of the present invention;

fig. 8 shows g when TF is 1.07 (upper half) and TF is 1.25 (lower half)_cpofdmAnd g_iterPulse and frequency response of;

fig. 9 shows g when TF is 1.07 (upper half) and TF is 1.25 (lower half)_cpofdmThe ambiguity surface of (left panel) and giter (right panel);

fig. 10 shows g at TF 1.07 (left panel) and TF 1.25 (right panel)_cpofdm(solid curve) and g_iterSIR contours (dashed curves);

fig. 11 shows g when TF is 1.25 and K is 1_iterConverge to g_TFL(left panel) and g_OOB(right panel);

fig. 12 shows g when TF is 1.07 and K is 1_iter；

Fig. 13 shows g at TF 1.07 (left panel) and TF 1.25 (right panel)_cpofdm(solid curve) and g_iterSIR contours (dashed curves);

fig. 14 shows a scenario of uplink TA-less access;

fig. 15 shows a scenario of HST-V2V with high mobility; and

fig. 16 shows the case of one symbol per TTI.

Detailed Description

Fig. 6 shows an embodiment of an iterative method according to the invention for designing a pulse shape with adjustable length, orthogonality and localization properties.

In step 600, a well-localized initial pulse waveform g is provided⁽⁰⁾(t)。g⁽⁰⁾(T) may be well localized in the T-F domain and may be, for example, a sub-exponential localized pulse shape such as a Gaussian pulse or a polynomial localized pulse shape such as a spline-type pulse. Here, in the present invention, Gaussian pulses may be selected

As g⁽⁰⁾(t) wherein g⁽⁰⁾(t) is predetermined and σ is the attenuation factor. In the context of this background, it is,

may be contracted or expanded to match the channel double dispersion properties. In the context of this background, it is,

at a given symbol duration T and frequency interval F, α > 1 is such that

The attenuation is faster in the time domain, and alpha < 1 makes

The attenuation is fast in the frequency domain. Given T and F, α > 1 attenuates faster in the time domain, where α < 1 attenuates faster in the frequency domain.

In particular, g⁽⁰⁾(T) may be localized in the time domain (e.g., rectangular pulses) or localized in the T-F domain (e.g., Gaussian pulses). For the

It may be shrunk or expanded to match the channel double dispersion characteristics.

At the input of the initial pulse waveform g⁽⁰⁾After (t), as can be seen in step 610 of FIG. 6, for the initial pulse waveform g⁽⁰⁾(t) performing an orthogonalization process and setting an iteration index to n-1. In this context, the orthogonalization process performed in step 610 may be the same as discussed and mentioned above with respect to the orthogonalized gaussian. This means that the optimized pulse shape g is generated by an optimization process⁽¹⁾(t) of (d). This orthogonalization may not be a complete orthogonalization, but some regularization may still remain after step 610 is performed. Moreover, orthogonalization may be further performed on the original or dual time-frequency grids. In this context, the optimized pulse wave is calculated by using the following general formulaShape g⁽¹⁾(t)：g⁽ⁿ⁾＝orth{g^(n-1)T, F, where n is 1, and an optimized pulse shape g⁽¹⁾(t) may be, for example, the Weyl-Heisenberg Riesz sequence configuration.

Subsequently, as can be seen in fig. 6, in step 620, a truncation window g of n ═ 1 is used_W ⁽ⁿ⁾By mixing g⁽¹⁾(t) and g_W ⁽¹⁾Multiply to truncate g⁽¹⁾(t) to obtain a truncated pulse waveform g_t ⁽¹⁾(t) of (d). Thereafter, through g⁽¹⁾(t)<--g_t ⁽¹⁾(t) resetting. In this context, the truncation window g may be changed each time truncation is performed_W ⁽ⁿ⁾Or the window g may be truncated for the entire iterative method_W ⁽ⁿ⁾Are the same. Commonly used windows include Rectangular (RECT) windows, raised cosine (RC (β)) windows or root raised cosine (RRC (β)) windows, where β is the roll-off factor. For β → 0, RC (β) and RRC (β) converge to REC. For example, in FIG. 6, only having a non-zero duration D may be considered_reqFixed window g of KT ═_WWherein D is_reqKT is the desired time interval and K.gtoreq.1. For K-4, the numerical results show that RC (0.25) may be a good choice. In this context, one iteration (loop) in fig. 6 is defined as a series of method steps 610 and 620, and the iteration index n counts the number of iterations (iteration loops) performed in the method of fig. 6.

Furthermore, in step 630, the generated pulse waveform g generated in method step 620 is checked⁽¹⁾(t) whether at least one predetermined requirement is met. In this context, | (g) when n ═ 1 is examined⁽ⁿ⁾-g^(n-1))||/||g^(n-1)||<Whether epsilon holds, where epsilon is a predetermined requirement as a threshold, in particular epsilon>0, convergence factor epsilon. Here, the convergence coefficient ∈ controls the orthogonality and TF localization properties of the output pulse waveform output in step 640. Large epsilon may focus more on TF localization, while small epsilon focuses on orthogonality. Alternatively, the predetermined requirement may also be that the number of iterations performed exceeds a certain maximum number of iterations. Furthermore, several predefined requirements are also conceivableA combination, e.g. of the convergence coefficient epsilon and the maximum number of iterations.

If the above condition of step 630 is true, then g is output⁽¹⁾(t), otherwise the iteration index is incremented by 1 in step 635 and the process returns to step 610, at which time the pair g is shown in step 610 of FIG. 6⁽¹⁾(t) not for g⁽⁰⁾(t) performing an orthogonalization process so as to obtain a value by calculating g when n is 2⁽ⁿ⁾＝orth{g^(n-1)T, F) to calculate g⁽²⁾(t) of (d). Thereafter, method steps 620 and 630 are performed again. If the condition of step 630 holds, output g⁽²⁾(t), otherwise increment the iteration index by 1 and perform method steps 610, 620 and 630 again until the condition of method step 630 is true. Then, the finally generated pulse waveform is output.

Fig. 7 shows an implementation of the method of fig. 6, wherein the method steps of the orthogonalization of step 610 are explained in further detail by calculating the following equation in step 710:

wherein the content of the first and second substances,

is defined above and frame g^(n-1)The associated frame operator and n is the iteration index. Furthermore, all other steps 700, 720, 730, 735 and 740 in the flowchart of fig. 7 correspond to steps 600, 620, 630, 635 and 640, respectively, in the flowchart of fig. 6.

Further, algorithm 1 below shows an iterative method for designing pulses with adjustable length, orthogonality, and localization properties that can be implemented in a computer program.

Algorithm 1

Since several parameters in the iterative methods of fig. 6 and 7 can be adjusted, the algorithm of the present invention remains flexible and feasible for any requirements set forth.

Furthermore, the methods of fig. 6 and 7 may be performed by a multi-carrier modulation system, in particular by a filter bank. Furthermore, the multi-carrier modulation system may be a transmitter, a receiver or other devices with computing capabilities, which may perform the methods of fig. 6 and 7 online or offline, which means connected to (online) or not connected to (offline) another system, e.g. via the internet. In another embodiment, the pulse waveforms generated by the methods of fig. 6 and 7 may also be used in a multicarrier modulation system, in particular in a filter bank, after generation. Of course, the computer program may also store instructions for execution on a computer that performs the methods shown in fig. 6 and 7. Further, the computer program may also be configured to subsequently use the pulse waveform after generating the pulse waveform by the method of fig. 6 and 7. In another embodiment, there may also be provided a system including a multicarrier modulation system configured to perform the methods of fig. 6 and 7 and a communication apparatus, wherein the multicarrier modulation system is configured to update the obtained pulse waveform g by performing communication between the multicarrier modulation system and the communication apparatus according to a communication protocol⁽⁰⁾(t)。

In the following, a performance analysis according to the invention presented in fig. 6 and 7 is shown. In addition to the usual impulse and frequency responses, the performance analysis hereinafter includes orthogonality and self-interference.

g_m，n(t) and γ_m，nThe orthogonality between (t) is described by a cross-ambiguity function defined as:

A_g，γ(τ，υ)＝∫g(t)γ^*(t-τ)e^-2j2πυtdt

wherein τ and ν represent time delay and frequency shift, respectively. Furthermore, the ambiguity surface is selected to visually observe the ambiguity function. For complete reconstruction, it is necessary

In addition, the channel is dispersed at TF

Introduces self-interference, which is determined by a cross-ambiguity function A_g，γIs expressed as (τ, υ) and is calculated as

Wherein the content of the first and second substances,

for a vector with normalized support (τ)₀，υ₀) Given scattering function of

Resulting SIR contour

Is very important because of the resulting SIR contour

Indicating the self-interference of a pulse when the signal carried on it experiences a proportional misalignment in the time and frequency domains.

Furthermore, the performance analysis in the present invention can be performed by evaluating the pulse g_iterAnd g is carried out for TF 1.07 and TF 1.25, respectively_iterCompared to CP-OFDM. All other basic simulation parameters are listed in table 1.

Table 1

Number of subcarriers: m	256
		Symbol overlap factor: k	4
Types of truncated windows	RC
		Roll-off factor of truncated window	0.25
Convergence coefficient	10-4

Furthermore, the impulse response and the frequency response can also be clearly seen in fig. 8 of the present patent application. In addition, orthogonality comparisons were also made in terms of SIR (dB, see table 2) and ambiguity surface (see fig. 9), respectively. For both transmission modes, the ambiguity function A_g，γ(τ, ν) regularly crosses zero at grid points τ -nT and ν -mF for non-zero integers n and m, thus guaranteeing ISI and ICI free transmission on an ideal channel. In addition, with g_cpofdmComparison with g_iterThe spread along the frequency axis is small, especially 1.25 for TF.

Table 2: g when K is 4_cpofdmAnd g_iterSIR (dB) comparison between

	TF＝1.07	TF＝1.25
			gcpofdm	∞	∞
giter	48.95	56.81

Furthermore, mismatch losses were also examined, where CP-OFDM has a mismatch loss ξ for TF ═ 1.07_dB0.3dB, 1.25 for TF, CP-OFDM has mismatch loss ξ_dB≈1dB。g_iterThere is no such loss.

Furthermore, by using the method of fig. 6 or 7, the method can converge to g for TF 1.25_TFLAnd g_OOB(see in particular fig. 11). In addition, table 3 below lists the basic parameters for achieving such a generalization. Furthermore, since there is no restriction on the relationship between n and m, a pulse waveform when TF is 1.07 and K is 1 is first obtained in the literature (see fig. 12).

Table 3: for convergence to g_TFLAnd g_OOBG of_iterSetting of parameters of

In this context, note that K-1 is for a special case called one symbol per Transmission Time Interval (TTI), where a guard period is inserted between uplink and downlink transmissions, so there is no ISI in this case. To analyze about g_iterShows the self-interference prototype at TF 1.07 and TF 1.07, respectivelySIR contour at 1.25

Both of which are in agreement with g_cpofdmA comparison is made.

In addition, the present application can be applied to various scenarios. For example, the invention may be applied to uplink timing unadjusted access. In uplink transmission, signals arrive at the base station with timing misalignment due to propagation delay. For example, for a cell with a radius of 800m, such a timing offset is up to 5.4 μ s. To address this problem, closed loop Timing Adjustment (TA) is employed. However, if each user equipment is used only to transmit a small data packet for a relatively long period of time, TA becomes a large overhead. For the purpose of alleviating the burden, uplink TA-free access or relaxed TA transmission is enabled as shown in fig. 14. As presented above, the proposed pulses with good orthogonality are more robust to temporal dispersion and can therefore be used in such scenarios.

Further, fig. 15 shows a scenario of a High Speed Train (HST) and a vehicle-to-vehicle (V2V) with high mobility. In this scenario, the objects move at high speed, which results in a more "double-dispersed" channel than in other scenarios. For example, when a car moves towards BS at a speed of 20m/s, for carrier frequency f_cThe corresponding doppler shift is 173Hz at 2.6 GHz. In conventional CP-OFDM, a longer cyclic prefix is used to handle the double dispersion, but inevitably increases the mismatch loss, e.g. 1.25 for TF _dB1 dB. However, as shown in fig. 15, using the pulses proposed in the present invention in such challenging scenarios can reasonably maintain reliable performance without incurring any mismatch loss.

Further, fig. 16 shows a case of one symbol per TTI. As shown in FIG. 16, the pulse of the present invention may be g_cpofdmEspecially for TF 1.25, because in this case its TF-supported contours are much larger than CP-OFDM.

Furthermore, in table 4 below, some applicable parameter settings for the aforementioned scenarios are presented.

Table 4: example parameter settings

The invention has been described herein in connection with various embodiments. However, other variations to the appended embodiments and practicing the claimed invention will be readily understood and effected by those skilled in the art upon a study of the drawings, the disclosure and the appended claims. In the claims, the word "comprising" does not exclude other elements or steps, and the indefinite article "a" or "an" does not exclude a plurality. A single processor of an entity may fulfill the functions of several items recited in the claims. The mere fact that certain measures are recited in mutually different dependent claims does not indicate that a combination of these measures cannot be used to advantage. A computer program may be stored/distributed on a suitable medium, such as an optical storage medium or a solid-state medium supplied together with or as part of other hardware, but may also be distributed in other forms, such as via the internet or other wired or wireless telecommunication systems.

Claims (7)

1. A method for generating a pulse waveform, comprising the steps of:

obtaining a pulse waveform g (t),

optimization is performed on the obtained pulse waveform g (t),

generating an optimized pulse waveform by truncating the optimized pulse waveform, including truncating the optimized pulse waveform by multiplying the optimized pulse waveform with a truncation window, wherein a series of the optimization method steps and the truncation method steps form an iteration, the truncation window being varied each time the truncation step in the iteration is performed,

determining whether the generated pulse waveform meets at least one predetermined requirement;

outputting the generated pulse waveform if the generated pulse waveform satisfies at least one predetermined requirement;

otherwise repeating the optimizing method step and the truncating method step by using the generated pulse waveform as the obtained pulse waveform.

2. The method of claim 1, wherein the optimization is orthogonalization and is by calculation

To perform said orthogonalization in such a way,

is a dual frame with filter banks

An associated dual Gabor frame operator, where T is the symbol duration of the input signal and F is the inter-carrier spacing.

3. The method according to claim 1, wherein the truncation window is a rectangular window RECT, a raised cosine window RC (β), or a root raised cosine window RRC (β), wherein β is a roll-off factor and β ≧ 0.

4. The method of claim 1, wherein the obtained pulse shape g (t) is polynomial localized or sub-exponential localized.

5. The method of claim 4, wherein the polynomial localized pulse shape g (t) is a spline-type pulse shape and the sub-exponential localized pulse shape g (t) is a Gaussian pulse shape.

6. The method of claim 1, wherein,

in a first alternative, the at least one predetermined requirement comprises a difference between the pulse waveform generated by the iteration and the pulse waveform generated by a previous iteration being below a threshold, or

In a second alternative, the at least one predetermined requirement comprises exceeding a maximum number of iterations.

7. A storage medium, wherein the storage medium stores a computer program which, when executed by a computer, causes the computer to perform the method for generating a pulse waveform of any one of claims 1 to 6.

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