CN108398669B - Space-time broadband self-adaptive monopulse angle measurement method based on non-pre-delay processing - Google Patents

Space-time broadband self-adaptive monopulse angle measurement method based on non-pre-delay processing Download PDF

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CN108398669B
CN108398669B CN201810072852.XA CN201810072852A CN108398669B CN 108398669 B CN108398669 B CN 108398669B CN 201810072852 A CN201810072852 A CN 201810072852A CN 108398669 B CN108398669 B CN 108398669B
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CN108398669A (en
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李会勇
谢明
陈敲月
胡君丽
谢菊兰
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University of Electronic Science and Technology of China
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
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    • G01S7/4052—Means for monitoring or calibrating by simulation of echoes

Abstract

The invention discloses a space-time broadband self-adaptive monopulse angle measurement method based on non-pre-delay processing, which comprises the following steps: calculating a space-time autocorrelation matrix of a received interference signal under a physical uniform linear array; calculating to obtain a sum beam weight vector by using a space-time adaptive beam forming method without pre-delay; correcting the sum beam weight vector to obtain a difference beam weight vector; and carrying out sum-difference amplitude comparison processing on each sample to obtain a sum-difference ratio value of each sample: and averaging the sum-difference ratio to obtain a signal deviation angle, and adding the signal deviation angle and the beam pointing angle to obtain a single-pulse angle measurement angle. The method is used for the broadband self-adaptive single-pulse angle measurement, can effectively inhibit broadband interference and simultaneously carry out the single-pulse angle measurement, and can still carry out the effective single-pulse angle measurement under the conditions of main lobe interference and low signal-to-noise ratio; compared with the traditional space frequency processing method, the method has the advantages of small required signal-to-noise ratio, small calculation amount and no need of signal detection on each frequency band.

Description

Space-time broadband self-adaptive monopulse angle measurement method based on non-pre-delay processing
Technical Field
The invention relates to a monopulse angle measurement technology in the field of array signal processing, in particular to a space-time processing method without pre-delay aiming at broadband monopulse to realize broadband self-adaptive monopulse angle measurement.
Background
Monopulse angle measurement is a commonly used angle measurement method in radar, which uses multiple antennas to simultaneously receive echo signals, and obtains angular position information of a target by comparing amplitudes or phases of the echo signals. The monopulse angle measurement method can determine the angle error information of the target only by one echo pulse, is simple to operate, has strong real-time performance and high angle measurement precision, and therefore, is widely applied to modern phased array radar systems and has important application value in the fields of radar, sonar, wireless communication and the like.
The single-pulse angle measurement is divided into amplitude comparison method and phase comparison method according to the angle information mode of extracting the target from the echo signal, and the amplitude comparison method is generally adopted for angle measurement. The traditional monopulse angle measurement algorithm is based on a single target of white noise, and when interference exists, particularly when the interference is close to the target, a large angle estimation error is generated.
In order to accurately measure the single-pulse angle while interference exists, a modern Digital-beam-forming (DBF) technology is introduced to inhibit the interference, so that a current common self-adaptive single-pulse angle measuring technology appears. The literature, "the mathematical principle and the engineering implementation of the traditional single pulse method", the system engineering and electronic technology 2004,26(4):467 and 473 deduces the basic principle of single pulse angle measurement, and provides theoretical support for the engineering implementation of single pulse angle measurement. The document "Synthesis of adaptive monopulse patterns," IEEE Transactions on Antennas and Propagation, 1999,47(5): 773-. When it is a wideband signal and there is wideband interference, the algorithm cannot measure the angle accurately. At present, the research on the broadband monopulse angle measurement in the fast sampling process is still few, and documents ' old and bright, prosperous satellite, korea jade soldier, and the like, ' broadband adaptive monopulse algorithm for improving bayesian focusing, ' science of electric waves, 2013, 28 (2): 0237-12 to incorporated herein by reference, discloses a wideband single-pulse goniometry algorithm, but it is not universal and requires distribution of desired and interfering signals. The conventional self-adaptive monopulse algorithm can work well on narrow-band signals, and the principle of the method is as follows:
in a linear array, the unit vector of the signal incidence direction is
r＝-sinθ (1)
By theta0Indicating beam pointing when the signal direction (theta ═ theta)0+ Δ θ) bitWithin the range of the main lobe beam, the linear relation of the monopulse ratio with respect to the deviation angle can be obtained as
Figure BDA0001558642250000021
Where k is a fixed constant, a (θ) is a steering vector, and the symbol "(.)H"denotes a matrix conjugation operation, and a (θ) is
Figure BDA0001558642250000022
Where M is the number of linear array elements, where the symbol "(-)T"denotes the matrix transposition operation, e is a natural base, j is an imaginary unit, phim(M-1, …, M) is
Figure BDA0001558642250000023
Where λ is c/f, c is the speed of light and f is the frequency.
A in formula (2)Σ(θ0) For static and beam weights, aΣ(θ0)＝a(θ0)。aΔ(θ0) Is static difference weight, which can be expressed as
Figure BDA0001558642250000024
In the formula,' indicates a Hadamard product.
When there is interference, ADBF (adaptive beamforming) is usually adopted to suppress the interference, improve the signal-to-interference-and-noise ratio of the output, and improve the detection performance of the target. The self-adaptive monopulse technology is that after the sum channel is calculated to obtain the self-adaptive weight, the sum beam direction vector is corrected to obtain the weight of the difference channel. According to the MVDR (minimum Variance relationship response) criterion, and the beam adaptation weight is
Figure BDA0001558642250000025
In practical application, R-1Usually obtained by inverse estimation using a sampling matrix, i.e.
Figure BDA0001558642250000026
Wherein z (N) represents the nth sampled interference signal sample, and N represents the number of samples.
Two constraints are provided for the difference beam weight: firstly, the self-adaptive monopulse ratio after the restraint approaches the conventional static monopulse ratio; the second is to minimize the output power of interference and noise. Since it is impossible to constrain each point on the angle identifying curve, only discrete points are selected for constraint, for example, 3 points of 0, ± Δ θ 'are selected as constraint points, where (Δ θ') is a constraint interval, and the following constraints are applied to the monopulse ratio:
Figure BDA0001558642250000031
and k is a linear ratio constant, the size can be defined by itself during implementation, and is set to be 1 in the implementation process for convenience. Σ (θ) is an adaptive sum beam, Δ (θ) is an adaptive difference beam, and its expression is:
Figure BDA0001558642250000032
wΔthe weights are adapted for the difference beam. The constraints of the formula (8) can be written in the form of a matrix
Figure BDA0001558642250000033
Wherein the constraint matrix H and the response matrix rho are respectively
H＝[a(θ0+Δθ′)a(θ0)a(θ0-Δθ′)]∈CM×3 (11)
Figure BDA0001558642250000034
The difference beam weight wΔThe interference power of the poor beam output must be minimized, the extremum problem can be described as,
Figure BDA0001558642250000035
the self-adaptive difference beam weight under the constraint can be obtained by applying the Lagrange multiplier method
wΔ＝R-1H(HHR-1H)-1ρH∈CM×1 (14)
When processing a broadband signal, the conventional method mainly includes two methods, namely a space-frequency method and a space-time method. The space-frequency method can be divided into a non-coherent processing method and a coherent processing method. The wideband signal can be regarded as the result of adding a plurality of narrowband signals, as follows
Figure BDA0001558642250000036
Wherein P is the number of narrow bands, A (f)m) For narrow-band signals equal in frequency to fmA matrix of steering vectors of the time of day,
Figure BDA0001558642250000037
for narrow-band signals equal in frequency to fmThe envelope signal of the time of flight or flight,
Figure BDA0001558642250000038
for narrow-band signals equal in frequency to fmThe noise signal of time.
The basic idea of space-frequency processing is to transform a broadband time domain signal into a frequency domain signal through Fourier transform, then perform correlation operation to convert the frequency domain signal into a narrowband signal corresponding to a plurality of frequency points, and then adopt a narrowband methodEach narrow-band signal is processed, and the space-frequency processing structure is shown as figure 1, wherein x ism(n) representing array elementsmWherein M is 0, …, M-1, M represents the number of array elements, xm,i(n) representing array elementsmOf the ith group of input signals, wm,jRepresenting array elementsmWherein J is 0, …, J-1, J denotes the number of subbands, Xm,i(j) Denotes xm,i(n) Discrete Fourier Transform (DFT) results; y isi(j) Denotes the beamforming result, y, of the j-th sub-band frequency domaini(j) Represents Yi(j) The inverse fourier transform (IDFT) result of (a). If the space-frequency method is adopted to carry out the broadband self-adaptive monopulse processing, for the space-frequency non-coherent processing, the broadband signal needs to be divided into narrow-band signals of a plurality of frequency points, and then the narrow-band of each frequency point needs to carry out the self-adaptive monopulse angle measurement, but the method needs to carry out the detection of an expected signal on the data of each frequency point, the detection of the expected signal is easy to make mistakes when the signal-to-noise ratio is low, and the calculated amount is larger because the monopulse direction measurement when the interference exists on the plurality of frequency points needs to be carried out; for the space-frequency coherent processing, a broadband signal needs to be projected to a certain frequency point in a focusing manner, and then adaptive monopulse angle measurement is performed on the basis of the frequency point, however, huge calculation amount is introduced by calculation of a focusing matrix and focusing transformation. For the broadband signal processing of the space-frequency structure, the problem that the signal gain is lost due to Fourier transformation exists, so that the angle measurement result is poor, and the angle measurement is not good under the condition of low signal-to-noise ratio.
Now comparing the classical space-time beam adaptive forming algorithm proposed by Frost, the processing structure diagram is shown in fig. 2, wherein xm(n) denotes a received signal, M is 0, …, M-1, M denotes the number of array elements, w* m,kThe kth time tap weight vector representing array element m, k being 0, …, N-1, N representing the time tap order. The space-time method can carry out pre-delay processing on the received signals to compensate the phase difference and the time delay difference of the expected signals among the array elements, and the inaccurate time delay difference compensation can cause the compensated expected signals not to meet the basic condition of single-pulse amplitude measurement angle, thereby further leading to the conventional Frost space-time self-adaptive beam formingThe algorithm cannot be used to perform wideband single pulse goniometry.
Disclosure of Invention
The invention provides a space-time broadband self-adaptive monopulse angle measurement method based on non-pre-delay processing, aiming at the technical problem that the monopulse angle measurement effect based on space-frequency processing is poor under low signal-to-noise ratio when interference exists in a broadband signal environment. Compared with the traditional self-adaptive monopulse angle measurement algorithm, the method has the advantages of effectively utilizing the broadband signal angle measurement, and compared with an empty frequency processing structure, the method has the advantages of low signal-to-noise ratio angle measurement, high angle measurement accuracy and no need of detecting signals of each frequency band.
In order to achieve the purpose, the technical scheme adopted by the invention is as follows:
a space-time broadband self-adaptive monopulse angle measurement method based on no need of pre-delay processing comprises the following steps:
calculating a space-time autocorrelation matrix of a received interference signal under a physical uniform linear array; carrying out space-time tapping on the limited-time snapshot data of the received interference signal, and then carrying out time average estimation on the tapped data to be used as an estimation value of a space-time autocorrelation matrix of the received interference signal, namely using an interference signal data covariance matrix as estimation of the interference signal autocorrelation matrix;
calculating to obtain a sum beam weight vector by using a space-time adaptive beam forming method without pre-delay; the preferred mode is as follows: based on the constraint of a space-time autocorrelation matrix, under the condition that a space-time two-dimensional vector on a given beam direction is kept unchanged, the minimum output power is calculated to obtain a sum beam weight vector;
correcting the sum beam weight vector to obtain a difference beam weight vector; the preferred mode is as follows: uniformly taking out a plurality of frequency points in the broadband signal bandwidth, and correcting the space-time two-dimensional vector of the taken frequency points and the beam weight vector by adopting a method similar to a narrow band to obtain a difference beam weight vector;
performing sum-difference amplitude comparison on each sample, namely performing time domain tap processing on a received interference signal under a physical uniform linear array, performing sum-difference beam forming on a time domain tap result and a difference beam weight vector, and performing sum-difference amplitude comparison on each snapshot after the sum-difference beam forming to obtain a sum-difference value;
and calculating the single-pulse angle measurement angle, namely averaging the sum-difference ratio to obtain a signal deviation angle, and adding the signal deviation angle and the beam pointing angle to obtain the single-pulse angle measurement angle.
In summary, due to the adoption of the technical scheme, the invention has the beneficial effects that:
1. when the pulse signal is a broadband signal and broadband interference exists, the single-pulse angle measurement algorithm provided by the invention can effectively inhibit the broadband interference and simultaneously carry out single-pulse angle measurement;
2. the method provided by the invention can still carry out effective angle measurement under the condition of main lobe interference;
3. the method provided by the invention can still carry out effective single-pulse angle measurement under the condition of low signal-to-noise ratio;
4. the method provided by the invention fully utilizes the characteristic that the space-time method is used for carrying out integral processing on the broadband signal, and has the advantages of small required signal-to-noise ratio, small calculation amount and no need of carrying out signal detection on each frequency band compared with the traditional space-frequency processing method.
Drawings
FIG. 1 is a diagram of a wideband space-frequency architecture;
fig. 2 is a block diagram of Frost space-time adaptive beamforming;
FIG. 3 is a flow chart of the method of the present invention;
fig. 4 is a space-time processing normalization and beam pattern, wherein fig. 4- (a) is a front view of the space-time processing normalization and beam pattern, and fig. 4- (b) is a side view of the space-time processing normalization and beam pattern;
fig. 5 is a space-time processing normalized difference beam pattern, wherein fig. 5- (a) is a front view of the space-time processing normalized difference beam pattern and fig. 5- (b) is a side view of the space-time processing normalized difference beam pattern;
FIG. 6 is a schematic view of a space-time processing wideband adaptive single-pulse angle-measuring curved surface, wherein FIG. 6- (a) is a front view of the space-time processing wideband adaptive single-pulse angle-measuring curved surface, and FIG. 6- (b) is a side view of the space-time processing wideband adaptive single-pulse angle-measuring curved surface;
FIG. 7 is a narrow-band adaptive monopulse angular discrimination curve;
fig. 8 is a space-time wideband adaptive monopulse angular surface with main lobe interference, where fig. 8- (a) is a front view of the space-time wideband adaptive monopulse angular surface with main lobe interference, and fig. 8- (b) is a side view of the space-time wideband adaptive monopulse angular surface with main lobe interference;
FIG. 9 is a diagram illustrating a space-time wideband adaptive single-pulse estimation angle RMS error with mainlobe interference;
fig. 10 is a schematic diagram of a wideband adaptive monopulse angle discriminating surface based on space-frequency non-coherent processing, where fig. 10- (a) is a front view of the wideband adaptive monopulse angle discriminating surface based on space-frequency non-coherent processing, and fig. 10- (b) is a side view of the wideband adaptive monopulse angle discriminating surface based on space-frequency non-coherent processing;
fig. 11 is a schematic diagram of a wideband adaptive monopulse angle detection curved surface based on a space-frequency coherent process, where fig. 11- (a) is a front view of the wideband adaptive monopulse angle detection curved surface based on the space-frequency coherent process, and fig. 11- (b) is a side view of the wideband adaptive monopulse angle detection curved surface based on the space-frequency coherent process;
FIG. 12 is a diagram of the root mean square error of the estimated angle within + -4 degrees of the beam pointing direction of the signal;
FIG. 13 shows the estimated angular root mean square error for different signal-to-noise ratios when the signal is pointing at-4 from the beam.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention will be described in further detail with reference to the following embodiments and accompanying drawings.
Referring to fig. 3, the method of the present invention is implemented as follows:
1. and calculating a space-time autocorrelation matrix of the received interference signals under the physical uniform linear array.
The incidence number of the interference signal is represented by Kr, and then the interference signal received by the M uniform linear arrays is modeled as
z(n)＝As(n)+v(n),n＝1,2,…,N (16)
Wherein N is fast beat number, v (N) is independent and equally distributed additive white Gaussian noise vector. The interference signal vector s (n) and the direction matrix A are defined as
s(n)＝[s1(n),s2(n),…,sKr(n)]T∈CKr×1 (17)
A＝[a(θ1),a(θ2),…,a(θKr)]∈CM×Kr (18)
Then the multi-snapshot received data of the array interference signal model is written into a matrix form
Z＝AS+V (19)
Wherein Z ∈ CM×N，A∈CM×Kr，S∈CKr×N，V∈CM×N。
Because the invention is realized based on a space-time processing structure, space-time tap processing is needed to be adopted for received interference data, wherein the tap realization mode is that
Figure BDA0001558642250000071
In equation (20), K is the time tap order.
Theoretically, the space-time autocorrelation matrix of the interfering signal is
RST＝E[zST(n)zST H(n)](21)
Wherein z isSTThe interference signal after the space-time tap. The samples required for the calculation of the correlation matrix in equation (21) are infinite and cannot be realized in actual engineering. Therefore, in practical cases, the correlation matrix R is often calculated using a finite number of samplesSTTime-averaged estimation of
Figure BDA0001558642250000072
Using data covariance matrices
Figure BDA0001558642250000073
Instead of the theoretical correlation matrix RST. Data covariance matrix
Figure BDA0001558642250000074
Can be calculated by
Figure BDA0001558642250000075
2. And calculating a sum beam weight vector by using a space-time self-adaptive beam forming method without pre-delay.
By theta0Representing the beam pointing direction, f representing the frequency, the space-time two-dimensional steering vector can be represented as
Figure BDA0001558642250000077
Wherein
Figure BDA0001558642250000078
Is the Kroncker product, as(θ0And f) is a space-domain steering vector with frequency f, which is defined as formula (3), at(f) Is a time-oriented vector expressed as
Figure BDA0001558642250000076
Wherein T issIs the sampling period. In the frequency range of broadband signals f0-B/2,f0+B/2]Evenly get L frequency points [ f1,f2,…,fL]Then the space-time two-dimensional steering matrix formed by them is
AST(θ0)＝[ast(θ0,f1)ast(θ0,f2)…ast(θ0,fL)]∈CMK×L (25)
The space-time response is
Figure BDA0001558642250000081
Keeping the constraint condition on the condition that the sum beam output power is minimum, and keeping the sum beam weight value woptDo the followingConstraining
Figure BDA0001558642250000082
I.e. when the constraint condition is satisfied
Figure BDA0001558642250000083
On the premise of solving the sum beam weight w which minimizes the sum beam output poweropt。
If Lagrange multiplier method is applied to solve the formula (27), the sum beam weight value which satisfies the constraint and has the minimum sum beam output power can be obtained
Figure BDA0001558642250000084
3. And correcting the sum beam weight vector to obtain a difference beam weight vector.
The sum beam weight woptAdopting a method similar to a narrow band to carry out alignment on L frequency points [ f1,f2,…,fL]Correcting the angle-identifying curved surface to obtain a difference beam weight wdiffThe specific implementation process is as follows:
the adaptive sum beam and the difference beam of each frequency point are respectively sigma (theta)0,f)、Δ(θ0,f)
Figure BDA0001558642250000085
The constraint matrix is
Figure BDA0001558642250000086
Wherein the constraint matrix
Figure BDA0001558642250000087
And a response matrix
Figure BDA0001558642250000088
Is composed of
Figure BDA0001558642250000089
Figure BDA0001558642250000091
Where (Δ θ') is the constraint interval, the difference beam weight wdiffThe interference power of the poor beam output must be minimized and this extremum problem can be described as
Figure BDA0001558642250000092
The self-adaptive difference beam weight under the constraint can be obtained by applying the Lagrange multiplier method
Figure BDA0001558642250000093
4. And comparing the sum and difference of each group of samples.
Performing time domain tap processing on the received signal with the same order K as the previous one, and assuming that the received signal matrix is X belongs to CM×NWith a time domain tap result of
Figure BDA0001558642250000094
The formed sum beam is
ysum＝wopt H*XST∈C1×(N-K+1) (36)
The difference beam formed is
ydiff＝wdiff H*XST∈C1×(N-K+1) (37)
So that the sum-to-difference ratio of each sample is
Figure BDA0001558642250000101
Where real () denotes taking the real part of the complex number in brackets,/dot division denotes that each number in the vector corresponds to a division.
5. And calculating the single-pulse angle measurement angle.
The sum-difference ratio of each sample calculated in the formula (38) is close to the real value through simulation observation, and the final result is calculated by averaging, so that the finally obtained single-pulse angle measurement angle is
Figure BDA0001558642250000102
The feasibility and the superiority of the method are described by comparing the method provided by the invention with the traditional self-adaptive narrow-band monopulse method and the traditional space-frequency processing method through simulation:
simulation test 1: in simulation, a uniform linear array of 16 array elements is assumed, the spacing of the array elements is half wavelength of the highest frequency of a signal, a wave beam points to 10 degrees, interference incoming directions are-30 degrees and 40 degrees respectively, wherein the signal-to-noise ratio (SNR) is 0dB, and the center frequency f of the signal is01GHz, bandwidth B300 MHz, sampling frequency f s2 × B, sample fast beat number snap 5000, two interference-to-noise ratio JNR 40dB, and interference center frequency f1＝1.2GHz、f20.9GHz, interference bandwidth B1＝280MHz、B2＝350MHz。
The constraint angle interval delta theta' selected in the simulation is 4 degrees, the distribution and the number of the frequency points are reasonably selected, the more the frequency division points are, the smaller the final angle measurement error is, but the more the frequency points are, the more the consumed degrees of freedom are, the sampling time order must be increased, so that K is larger than L, and meanwhile, the distribution of the frequency points is as uniform as possible to cover the whole frequency band. The order of the time tap selected in the simulation process is uniformly K-15, and the number of frequency points is uniformly L-10, and the frequency points are uniformly selected in the signal bandwidth range. Fig. 4 is a normalized sum beam diagram, fig. 5 is a normalized difference beam diagram, fig. 6 is a graph of an adaptive single-pulse angle-identifying curved surface in a signal bandwidth, and fig. 7 is a narrow-band single-pulse angle-identifying curve under the same conditions.
Fig. 4 shows that the sum beam weight vector can hold the signal and suppress the interference, and fig. 5 shows that the difference beam weight vector can make the signal form a deep null in the beam pointing direction, which is consistent with the principle of adaptive single pulse sum difference amplitude measurement. It can be seen from the simulation fig. 6 that the single pulse to difference ratio has good focality in the whole frequency band within the range of ± 4 ° of beam pointing, and although the angle identification curve has a certain amplitude fluctuation, which is related to only selecting L frequency points when the difference beam weight is obtained by correction, the single pulse ratio can be approximately regarded as a linear function with the deviation of the central angle Δ θ slope of 1. Comparing fig. 6- (b) and fig. 7, it can be seen that the curves are approximately the same, which shows that the constrained wide-band angle discriminating curve is very close to the conventional narrow-band adaptive single-pulse angle discriminating curve.
Simulation test 2: the first interference direction was changed to 7 ° while keeping the other experimental conditions in simulation experiment 1 unchanged. The signal direction is changed within the range of +/-4 degrees of the beam pointing direction, the method is used for measuring the angle of the single pulse, and the Monte-Carlo test times are 100. The measurement accuracy of the space-time broadband self-adaptive monopulse angular measurement error is verified by adopting a Root Mean Square Error (RMSE) which has the expression as follows,
Figure BDA0001558642250000111
wherein, M is the number of times of the test,
Figure BDA0001558642250000112
and theta is the estimated value of the angle at the m time, and theta is the real value of the signal angle. Under the condition, the angle curve of the invention is shown in figure 8, and the root mean square error of the simulated angle measurement is shown in figure 9.
Fig. 8 shows that when the interference is within 3dB of the main lobe beam, the angle-identifying curved surface of the present invention will have a deviation at the interference position, and the correction angle position ± 4 ° and 0 ° are identical to the theoretical value, and the linear relationship with a slope of 1 can still be maintained in other angle regions, so that when the main lobe interference exists, the angle-identifying curved surface of the present invention can still maintain a good linear relationship in the angle regions other than the interference. Fig. 9 shows that when the interference is within 3dB of the main lobe beam, the method of the present invention can not effectively measure the angle near the interference, and can still accurately measure the angle in other angle area ranges. This demonstrates the advantage of the method of the invention that accurate angle measurement is still possible in the presence of mainlobe interference.
Simulation test 3: dividing a broadband signal into a plurality of equivalent narrowband signals on a frequency domain by adopting a traditional broadband signal processing method, then carrying out self-adaptive monopulse angle measurement by using a narrowband processing mode, keeping the signal condition in simulation 1 unchanged, wherein the number J of the selected subband divisions is 100, increasing the fast beat number of samples in the simulation 1 to snap 20000 in order to ensure that each subband has signals, and using f to process the space frequency in order to ensure that each subband has signalss1The signal is sampled B. The simulation is based on the broadband self-adaptive monopulse angular profile of the space-frequency non-coherent processing as shown in figure 10, and the simulation is based on the broadband self-adaptive monopulse angular profile of the space-frequency non-coherent processing as shown in figure 11.
FIG. 10 shows that the angle-identifying curved surface for realizing the broadband adaptive single-pulse angle measurement based on the space-frequency non-coherent processing is close to the angle-identifying curved surface FIG. 6 of the algorithm provided by the invention, which shows that the method has the feasibility of realizing the broadband adaptive single-pulse angle measurement; fig. 11 shows that the angle-identifying curved surface for realizing the broadband adaptive single-pulse angle measurement based on the space-frequency coherent processing is close to the angle-identifying curved surface fig. 6 of the algorithm provided by the invention, which shows that the method has the feasibility of realizing the broadband adaptive single-pulse angle measurement. The performance of the method of the present invention is compared with that of the conventional space-frequency processing in the following simulation experiment.
Simulation test 4: the conditions in the simulation 1 are kept unchanged, samples are increased to 20000 when the broadband single-pulse angle measurement is realized by using a space-frequency processing method, the signal direction is changed within the range of +/-4 degrees of the beam pointing direction, and the Monte-Carlo test frequency is 100. And verifying the measurement precision of the space-time broadband self-adaptive monopulse angle error by adopting a Root Mean Square Error (RMSE), wherein the expression of the root mean square error is shown as a formula (40). The simulated root mean square error is shown in figure 12.
Fig. 12 shows that the algorithm of the present invention can still measure the angle accurately when the SNR is 0dB, while at the same time, the traditional space-frequency processing is adopted to realize the adaptive single-pulse angle measurement, and the method of the present invention approaches to 0 at the constrained angle, which is consistent with the theory. This illustrates the advantage of the invention with accurate goniometry within 3dB of beam pointing at low signal-to-noise ratios.
Simulation test 5: keeping the conditions in simulation 4 unchanged, fixing the signal direction to deviate from the beam pointing direction by-4 degrees, namely, the signal direction is 6 degrees, changing the signal-to-noise ratio, and comparing the angle measurement performance of the broadband space-time and space-frequency processing under the same conditions as shown in fig. 13.
FIG. 13 shows that the algorithm of the invention can realize accurate angle measurement under a very low signal-to-noise ratio of-10 dB, and the estimation angle root mean square error can approach to be stable after-5 dB, which is far superior to the broadband single pulse angle measurement based on space-frequency processing under the same condition. Looking at the final convergence of each curve in the graph, it was found that the root mean square error of the method of the present invention and the two methods of space-frequency processing are nearly the same and close to 0, which is consistent with the correction interval Δ θ' being 4 °. This again illustrates the advantage of the algorithm of the present invention in accurately measuring angles at low signal-to-noise ratios under the same conditions.
While the invention has been described with reference to specific embodiments, any feature disclosed in this specification may be replaced by alternative features serving the same, equivalent or similar purpose, unless expressly stated otherwise; all of the disclosed features, or all of the method or process steps, may be combined in any combination, except mutually exclusive features and/or steps.

Claims (4)

1. A space-time broadband self-adaptive monopulse angle measurement method based on non-pre-delay processing is characterized by comprising the following steps:
calculating a space-time autocorrelation matrix of a received interference signal under a physical uniform linear array;
calculating to obtain a sum beam weight vector by using a space-time adaptive beam forming method without pre-delay;
correcting the sum beam weight vector to obtain a difference beam weight vector;
and (3) carrying out sum and difference amplitude comparison processing on each sample: carrying out time domain tap processing on a received interference signal under a physical uniform linear array, carrying out sum and difference beam forming on a time domain tap result and a difference beam weight vector, and carrying out sum and difference amplitude comparison on each snapshot after the sum and difference beam forming to obtain a sum and difference value;
averaging and calculating the sum-difference ratio to obtain a signal deviation angle, and adding the signal deviation angle and the beam pointing angle to obtain a single-pulse angle measurement angle;
the sum beam weight vector calculated by the space-time adaptive beam forming method without pre-delay is as follows:
by theta0Representing the beam pointing direction, and f representing the frequency, the space-time two-dimensional steering vector is:
Figure FDA0003118485430000011
wherein the content of the first and second substances,
Figure FDA0003118485430000012
is a Kroncker product, M is the number of uniform linear arrays, K is the time tap order, as(θ0And f) is the space domain steering vector at frequency f, at(f) Is a time-oriented vector whose expression is:
Figure FDA0003118485430000016
wherein T issIs a sampling period;
in the frequency range of broadband signals f0-B/2,f0+B/2]Evenly get L frequency points [ f1,f2,…,fL]And then the space-time two-dimensional steering matrix formed by the two steering matrices is: a. theST(θ0)＝[ast(θ0,f1)ast(θ0,f2)…ast(θ0,fL)]∈CMK×LThe space-time response is
Figure FDA0003118485430000013
Wherein f is0Is the signal center frequency, B is the bandwidth;
when the constraint condition is satisfied
Figure FDA0003118485430000014
On the premise of solving the sum beam weight w which minimizes the sum beam output poweroptWherein the sum beam output power is:
Figure FDA0003118485430000015
representing a data covariance matrix.
2. The method of claim 1, wherein the sum beam weight vector is calculated by: and (3) enabling the output power to be calculated to be minimum under the condition that the space-time two-dimensional vector on the direction of a given beam is kept unchanged based on the constraint of the space-time autocorrelation matrix to obtain a sum beam weight vector.
3. The method of claim 1, wherein modifying the sum beam weight vector to obtain the difference beam weight vector comprises:
uniformly taking out a plurality of frequency points in the broadband signal bandwidth;
calculating the adaptive sum beam of each frequency point
Figure FDA0003118485430000021
Differential beam
Figure FDA0003118485430000022
Wherein woptRepresentation and beam weight, wdiffRepresenting the weight of the difference beam, the sign' (.)H"represents a matrix conjugation operation;
when the constraint condition is satisfied
Figure FDA0003118485430000023
On the premise of (1) solving the difference beam weight w with the minimum interference power output by the difference beamdiffObtaining a difference beam weight vector, wherein
Figure FDA0003118485430000024
A constraint matrix is represented that is,
Figure FDA0003118485430000025
representing a response matrix.
4. The method of claim 1, wherein the space-time autocorrelation matrix is calculated in the manner: the finite snapshot data of the received interference signals under the physical uniform linear array are firstly subjected to space-time tap processing, and then the tapped data are subjected to time average estimation to obtain the space-time autocorrelation matrix.
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