CN108108573A - A kind of IGBT power module junction temperature dynamic prediction method - Google Patents

Abstract

The invention provides a method for dynamically predicting the junction temperature of an IGBT power module, which solves the current problems of excessive derating and unreasonable thermal design to avoid failure caused by excessive junction temperature and excessive fluctuation of the IGBT power module. It performs dynamic analysis of circuit parameters including modulation ratio, output current, output voltage, output frequency, etc. according to the operating state of the motor, and inputs the analytical values into the junction temperature calculation model considering electrothermal coupling to realize dynamic junction temperature prediction under working conditions .

Description

A Dynamic Prediction Method of IGBT Power Module Junction Temperature

technical field

The invention relates to the field of junction temperature prediction of IGBT modules, in particular to a dynamic junction temperature prediction method of an IGBT power module in a three-phase inverter system under working conditions.

Background technique

The kHz-level switching frequency characteristics of the IGBT (insulated gate transistor) power module will cause a large heat loss during operation, resulting in the rise and fluctuation of the temperature (junction temperature) at the PN junction of the chip, which will lead to module failure in severe cases . The existing countermeasures are to derate the IGBT module, or match the inverter system where it is located with a radiator with a large mass and volume to fully ensure heat dissipation. However, in the above measures, excessive derating will reduce the application range of the power module, and unreasonable heat dissipation design will also increase the weight of the system and waste the occupied space. Therefore, accurate junction temperature prediction is of great significance for determining the safety limit of IGBT power modules, improving application range and reliability, and making reasonable thermal design.

There is mutual coupling between the electric field and the temperature field in the IGBT power module. At the same time, the operating conditions of related circuit parameters in different applications also change in real time. The existing IGBT junction temperature prediction models often ignore the influence of electrothermal coupling, or can only be used for specific The junction temperature prediction at the working point is still insufficient in the application of working conditions. Therefore, it is necessary to predict the dynamic junction temperature of the IGBT power module under working conditions.

Contents of the invention

Aiming at the above-mentioned technical problems in this field, the present invention provides a method for dynamically predicting the junction temperature of an IGBT power module, which specifically includes the following steps:

Step 1. Obtain the torque and speed at the first working point of the motor according to the running state of the motor;

Step 2. Establish an analytical model of the motor operating point; input the torque and rotational speed obtained in the step 1 into the established analytical model of the motor operating point to obtain dq axis current and voltage values, and then obtain the inverter output three-phase current Voltage, switching signal. Store the grid resistance of the driving end, the switching frequency and the DC bus voltage information of the DC end;

Step 3. Establish a loss calculation model of the IGBT power module; set the initial temperature of the IGBT in the power module and the anti-parallel diode FWD, and input the parameters described in step 2 into the established loss calculation Model, obtain the loss value of described IGBT and anti-parallel diode FWD;

Step 4, establishing a thermal resistance network model of the IGBT power module; inputting the loss value obtained in the step 3 into the thermal resistance network model to obtain the junction temperature corresponding to the current motor operating point;

Step 5. Feedback the junction temperature obtained in step 4 into the loss calculation model of the IGBT power module to realize dynamic junction temperature prediction under working conditions.

Further, the establishment of the motor operating point analytical model described in step 2 specifically includes:

For the surface-mounted motor, the inductance of the d and q axes is the same as that of the phase inductance, so its electromagnetic torque T _em is expressed as follows

Among them, p is the number of pole pairs, ψ _f is the flux linkage of the permanent magnet, and i _q is the q-axis current;

Or, in terms of expressing the electromagnetic torque by the motor constants k _t and i _q :

Judging whether the motor is in the non-weakening field area or the field-weakening area: use id=0 in the non-field-weakening area, and the amplitude of the output phase voltage at this time is expressed as:

In the field weakening area, the amplitude of the output phase voltage is expressed as follows:

After obtaining id and iq, the output of the three phases A, B, and C at the corresponding torque speed is obtained through constant amplitude transformation. Among them, the change of dq axial αβ rotating coordinate system applies Park transformation, which has the following conversion relationship:

Among them, θ is the phase angle;

The transformation from the αβ rotating coordinate system to the three-phase A, B, and C is Clark transformation, and the transformation relationship is as follows:

Among them, i _A , i _B , and i _C are the currents of each phase respectively.

This model can realize the state analysis of circuit parameters such as output voltage, current, frequency, switching signal, etc. of the module under different load conditions.

Further, the establishment of the loss calculation model of the IGBT power module described in step 3 specifically includes:

The loss P _Module of the IGBT power module includes: the on-state loss P _{IGBT_con generated during the IGBT operation, the turn-on loss P IGBT_on and} the turn-off loss P _{IGBT_off} during the switching transient state; the on-state loss P _{FWD_con} and the reverse recovery loss during the FWD operation P _{FWD_re} :

P _Module ＝P _{IGBT_con} +P _{IGBT_on} +P _{IGBT_off} +P _{FWD_con} +P _{FWD_re}

The on-state voltage drops V _CE and V _D when the IGBT and FWD are turned on are composed of two parts: the respective threshold voltages V _CEO , V _DO and the voltage drops generated by the on-state resistances R _ch and R _d , and are related to the actual temperature T , the relationship is expressed as follows, T ₀ is the reference temperature, I _C and I _D are the currents passing through the IGBT and FWD respectively, among which, b _T , m and b _D are temperature-related items that can be obtained by curve fitting.

V _D (T)＝V _DO (T ₀ )+b _D ·(TT ₀ )+R _d (T)·I _D ²

Calculate the on-state loss, where D _T and D _D are the duty cycle of IGBT and FWD in the unit switching period:

P _{IGBT_con} ＝(V _CEO (T)·I _C +R _ch (T)I _C ² )·D _T

P _{FWD_con} ＝(V _DO (T)I _D +R _d (T)I _D ² )·D _D

When the switching frequency is f _sw , the turn-on power loss P _{IGBT_on} of the IGBT, the turn-off power loss P _{IGBT_off} and the reverse recovery power loss P _{FWD_re} of the FWD can be expressed as follows:

P _{IGBT_on} = f _sw E _{IGBT_on}

P _{IGBT_off} = f _sw E _{IGBT_off}

P _{FWD_re} = f _sw · E _{FWD_re} ;

in,

Among them, E _{IGBT_on} is the turn-on energy loss of IGBT, E _{IGBT_off} is the turn-off energy loss of IGBT, E _{FWD_re} is the reverse recovery energy loss of FWD, a _on , b _on , c _on , a _off , b _off , c _off , a _re , b _re , _cre are fitting constants, k _on , k _off , k _re are temperature-related items, E _on (R _g ), E _off (R _g ), E _off (R _g ) and E _on ( R _rated ), E _off (R _rated ), E _off (R _rated ) are the turn-on and turn-off energy consumption of the corresponding IGBT under the actual gate resistance and the reference gate resistance, and the reverse recovery energy loss of FWD, V _{DC_rated} is the reference DC bus value.

Further, the establishment of the thermal resistance network model of the IGBT power module described in step 4 specifically includes: the model is based on the following assumptions:

(1) Neglecting the influence of heat radiation and heat convection, the form of heat transfer in the module is heat conduction:

(2) Due to the filling of insulating silica gel, the heat transfer path is from the chip to the substrate:

(3) The influence of thermal coupling between chips is negligible:

(4) Neglecting the local temperature difference of a single chip, using the lumped parameter method;

Based on the above assumptions, a fourth-order Foster thermal resistance network model is established.

Further, the junction temperature corresponding to the current motor working point is calculated in the following manner:

Assuming that the bottom surface temperature of the copper substrate is constant as T _C , the thermal resistances of the IGBT and FWD from their respective chips to the bottom case are R _{jc_IGBT} and R _{jc_FWD} , and the junction temperatures T _{j_IGBT} and T _{j_FWD} of the IGBT and the antiparallel diode FWD can be expressed as follows:

T _{j_IGBT} ＝T _c +R _{jc_IGBT} P _IGBT

T _{j_FWD} ＝T _c +R _{jc_FWD} ·P _FWD

Wherein, P _IGBT includes P _{IGBT_con} , P _{IGBT_on} and P _{IGBT_off} , and P _FWD includes P _{FWD_con} and P _{FWD_rr} .

The above-mentioned method provided by the present invention solves the current problems of excessive derating and unreasonable thermal design to avoid failure caused by excessively high junction temperature and excessive fluctuation of the IGBT power module. It performs dynamic analysis of circuit parameters including modulation ratio, output current, output voltage, output frequency, etc. according to the operating state of the motor, and inputs the analytical values into the junction temperature calculation model considering electrothermal coupling to realize dynamic junction temperature prediction under working conditions . Compared with the prior art, it has many non-obvious beneficial effects.

Description of drawings

Fig. 1 is a schematic flow chart of the method provided according to the present invention

Figure 2 is a schematic diagram of the half-bridge structure of the IGBT power module

Figure 3 is a schematic diagram of the switching transient process of the IGBT

Figure 4 is an equivalent schematic diagram of the internal package of the IGBT power module

Figure 5 is a schematic diagram of the principle of SPWM bipolar modulation (regular sampling method)

Figure 6 is Foster's equivalent thermal resistance network model

Figure 7 is the judgment logic of the motor's field-weakening area and non-field-weakening area

Figure 8 is a schematic diagram of the switching signal under SWPM modulation

Detailed ways

The technical solutions of the present invention will be further explained in detail below in conjunction with the accompanying drawings.

As shown in Figure 1, the present invention provides a method for dynamically predicting the junction temperature of an IGBT power module, which specifically includes the following steps:

Step 1. Obtain the torque and speed at the first working point of the motor according to the running state of the motor;

Step 2. Establish an analytical model of the motor operating point; input the torque and rotational speed obtained in the step 1 into the established analytical model of the motor operating point to obtain dq axis current and voltage values, and then obtain the inverter output three-phase current Voltage, switching signal. Store the grid resistance of the driving end, the switching frequency and the DC bus voltage information of the DC end;

Step 3. Establish a loss calculation model of the IGBT power module; set the initial temperature of the IGBT in the power module and the anti-parallel diode FWD, and input the parameters described in step 2 into the established loss calculation Model, obtain the loss value of described IGBT and anti-parallel diode FWD;

Step 4, establishing a thermal resistance network model of the IGBT power module; inputting the loss value obtained in the step 3 into the thermal resistance network model to obtain the junction temperature corresponding to the current motor operating point;

Step 5. Feedback the junction temperature obtained in step 4 into the loss calculation model of the IGBT power module to realize dynamic junction temperature prediction under working conditions.

In a preferred embodiment of the present application, the establishment of the motor operating point analysis model described in step 2 specifically includes:

For the surface-mounted motor, the inductance of the d and q axes is the same as that of the phase inductance, so its electromagnetic torque T _em is expressed as follows

Among them, p is the number of pole pairs, ψ _f is the flux linkage of the permanent magnet, and i _q is the q-axis current;

Or, in terms of expressing the electromagnetic torque by the motor constants k _t and i _q :

It is judged whether the motor is in the non-field-weakening area or the field-weakening area, as shown in Figure 7: In the non-field-weakening area, id=0 is generally used to meet the control requirements. But when the voltage reaches the limit u _lim , if the motor wants to run at a higher speed, it is necessary to reduce the excitation current, that is, to perform field-weakening control. Even if id becomes smaller and negative, the inverter can output under SPWM modulation The amplitude of the maximum phase voltage is 0.5U _DC .

Therefore, if the motor is in the non-field weakening area, then id=0, the amplitude of the output phase voltage at this time can be expressed as:

And if it has entered the field weakening area, the amplitude of the output phase voltage is expressed as follows:

Therefore, when performing field weakening, it needs to be judged based on certain logic. At any given speed and torque, the id can be calculated when it has entered the field weakening zone. Then combined with the above-mentioned field-weakening discrimination, if it is not field-weakening, it means id=0, and the calculated value is reset to 0. If it is in the field-weakening area, it remains unchanged according to the calculation results of the above test.

After obtaining id and iq, the output of three-phase A, B, and C at the corresponding torque speed can be obtained through constant amplitude conversion. Among them, the change of the dq axial αβ rotating coordinate system is Park transformation, and its conversion formula is as follows:

The conversion of αβ to three-phase A, B, C is Clark transformation, and the conversion formula is as follows.

Taking phase A as an example, assuming its phase voltage U _A and triangular carrier wave U _triangle , the signal output 1 means that the upper bridge arm is on and the lower bridge arm is off; 0 means that the lower bridge arm is on and the upper bridge arm is off. Taking phase A as an example, the output rule of the switching signal is as follows: When U _A > U _triangle , the switching signal outputs 1, the upper bridge arm is turned on, and the IGBT or the same bridge arm FWD works in combination with the positive and negative phase voltage i _A Discrimination; when U _A < U _triangle , the switch signal outputs 0, the lower bridge arm is turned on, and the corresponding IGBT and FWD work discrimination is also based on the positive and negative judgment of i _A and is opposite to the upper bridge arm. The switch signal output under this discrimination is shown in the figure:

This model can realize the state analysis of circuit parameters such as output voltage, current, frequency, switching signal, etc. of the module under different load conditions.

In a preferred embodiment of the present application, the loss calculation model of establishing the IGBT power module described in step 3 specifically includes:

The IGBT power module usually includes an IGBT and a diode FWD connected in antiparallel, thereby forming a power module with a typical half-bridge structure as shown in Figure 2, including two upper and lower bridge arms. The loss P _Module of the IGBT power module includes: the transient process of the IGBT is shown in Figure 3, the on-state loss P IGBT_con generated during its operation, the turn-on loss P _{IGBT_on} and the turn-off loss P _{IGBT_off} during the switching transient _state ; when the FWD is working On-state loss P _{FWD_con} and reverse recovery loss P _{FWD_rr} :

P _Module ＝P _{IGBT_con} +P _{IGBT_on} +P _{IGBT_off} +P _{FWD_con} +P _{FWD_re}

The on-state voltage drops V _CE and V _D when the IGBT and FWD are turned on are composed of two parts: the voltage drops generated by their respective threshold voltages V _CEO , V _DO and the on-state resistances R _ch and R _d , and are related to the temperature T. The relationship is expressed as follows, T0 is the reference temperature, I _C and I _D are the currents passing through the IGBT and FWD respectively, where b _T , m and b _D are all temperature-related items that can be obtained through curve fitting.

V _D (T)＝V _DO (T ₀ )+b _D ·(TT ₀ )+R _d (T)·I _D ²

Calculate the on-state loss, where D _T and D _D are the duty cycle of IGBT and FWD in the unit switching period:

P _{IGBT_con} ＝(V _CEO (T)·I _C +R _ch (T)I _C ² )·D _T

P _{FWD_con} ＝(V _DO (T)I _D +R _d (T)I _D ² )·D _D

The turn-on process of the IGBT includes three stages: turn-on delay t _d(on) , current rise t _ri , and voltage drop t _fv . The turn-on energy E _{IGBT_on} generated by each turn-on of the IGBT is expressed as follows:

Among them, V _DC is the DC bus voltage, I _RM is the diode reverse recovery peak current

The turn-off process of the IGBT is from the conduction state to the forward blocking state. This process includes three stages: voltage rising t _rv , current falling t _fi and tailing t _tail . The turn-off energy E _{IGBT_off} generated every time the IGBT is turned off is expressed as follows, and I _tail is the tail current.

Among them, △V is the additional voltage spike.

Since the time of each stage in the switching process is difficult to determine, the linear approximation of the voltage and current change rate can be obtained. The IGBT switching energy loss has a linear relationship with the DC bus voltage _{V DC} , and a quadratic relationship with the collector current IC. Considering the influence of gate resistance and temperature, the above formula can be transformed into:

Similarly, the reverse recovery loss of a diode can be expressed as follows

In the SPWM bipolar modulation shown in Figures 5 and 8, the IGBT works in the positive half cycle of the current, and the FWD works in the negative half cycle of the current. Through the geometric similarity relationship, the pulse width δ of the IGBT and FWD devices of the upper bridge arm in the kth modulation wave period can be expressed as follows, where T ₀ is the modulation wave period, and T _C is the switching period.

Therefore, under SPWM modulation, the conduction loss P _{IGBT_con} and P _{FWD_con} of the upper bridge arm IGBT and FWD in the kth modulation wave cycle can be obtained as

When the switching frequency is f _sw , the turn-on power loss P _{IGBT_on} of the IGBT, the turn-off power loss P _{IGBT_off} and the reverse recovery power loss P _{FWD_re} of the FWD can be expressed as follows:

P _{IGBT_on} = f _sw E _{IGBT_on}

P _{IGBT_off} = f _sw E _{IGBT_off}

P _{FWD_re} = f _sw · E _{FWD_re} .

In a preferred embodiment of the present application, the establishment of the thermal resistance network model of the IGBT power module described in step 4 specifically includes: as shown in Figure 4, the model is based on the following assumptions:

(1) Neglecting the influence of heat radiation and heat convection, the form of heat transfer in the module is heat conduction:

(2) Due to the filling of insulating silica gel, the heat transfer path is from the chip to the substrate:

(3) The influence of thermal coupling between chips is negligible:

(4) Neglecting the local temperature difference of a single chip, the lumped parameter method is adopted.

Each layer of material in the IGBT module can be regarded as an isotropic thin flat wall. And its Bi number is very small, and the lumped parameter method can be used for unsteady analysis. Its one-dimensional unsteady heat conduction equation can be expressed as follows:

In the formula, ρ is the material density, and C _P is its heat capacity. Differential equations and electrical equations describing the physical process of one-dimensional heat conduction

The equations for conduction have the same form. Therefore, through the electrothermal analogy, the problems about heat can be converted into electrical

The problem is that thermal resistance is compared to resistance, thermal capacity is compared to capacitor, and power is similar to current. And the system over time

The changing thermal impedance value can be expressed as a simple analytical formula as follows, where τ is the thermal time constant,

It can be seen from the formula that the value of thermal resistance R and thermal capacity C determines the response of the system to the step function of power loss.

τ _i =R _i ·C _i

The parameters matching the model can be obtained by fitting the transient thermal impedance curve, and the specific calculation

The calculation formula is as follows, where n is the order of fitting. The fourth-order Foster thermal resistance network model is thus obtained, as shown in Figure 6.

In a preferred embodiment of the present application, the junction temperature corresponding to the current motor operating point is calculated in the following manner:

For a single IGBT power module, assume that the bottom surface temperature of the copper substrate is constant as T _C , the thermal resistances of the IGBT and FWD from their respective chips to the case are R _{jc_IGBT} and R _{jc_FWD} , and the junction temperature T _{j_IGBT} and T _{j_FWD} of the IGBT and the antiparallel diode FWD can be They are expressed as follows:

T _{j_IGBT} ＝T _c +R _{jc_IGBT} P _IGBT

T _{j_FWD} ＝T _c +R _{jc_FWD} ·P _FWD

Wherein, P _IGBT includes P _{IGBT_con} , P _{IGBT_on} and P _{IGBT_off} , and P _FWD includes P _{FWD_con} and P _{FWD_re} .

Although the embodiments of the present invention have been shown and described, those skilled in the art can understand that various changes, modifications and substitutions can be made to these embodiments without departing from the principle and spirit of the present invention. and modifications, the scope of the invention is defined by the appended claims and their equivalents.

Claims (5)

1. A method for dynamic prediction of IGBT power module junction temperature, characterized in that: specifically comprise the following steps: Step 1. Obtain the torque and speed at the first working point of the motor according to the running state of the motor; Step 2. Establish an analytical model of the motor operating point; input the torque and rotational speed obtained in the step 1 into the established analytical model of the motor operating point to obtain dq axis current and voltage values, and then obtain the inverter output three-phase current Voltage, switching signal. Store the grid resistance of the driving end, the switching frequency and the DC bus voltage information of the DC end; Step 3. Establish a loss calculation model of the IGBT power module; set the initial temperature of the IGBT in the power module and the anti-parallel diode FWD, and input the parameters described in step 2 into the established loss calculation Model, obtain the loss value of described IGBT and anti-parallel diode FWD; Step 4, establishing a thermal resistance network model of the IGBT power module; inputting the loss value obtained in the step 3 into the thermal resistance network model to obtain the junction temperature corresponding to the current motor operating point; Step 5. Feedback the junction temperature obtained in step 4 into the loss calculation model of the IGBT power module to realize dynamic junction temperature prediction under working conditions. 2. The method according to claim 1, characterized in that: the establishment of the motor operating point analysis model described in the step 2 specifically includes: For surface-mounted motors, the d-axis inductance L _d is the same as the q-axis inductance L _q and is equal to the phase inductance, so its electromagnetic torque T _em is expressed as follows: <mrow><msub><mi>T</mi><mrow><mi>e</mi><mi>m</mi></mrow></msub><mo>=</mo><mfrac><mn>3</mn><mn>2</mn></mfrac><mi>p</mi><mo>&amp;CenterDot;</mo><msub><mi>&amp;psi;</mi><mi>f</mi></msub><mo>&amp;CenterDot;</mo><msub><mi>i</mi><mi>q</mi></msub></mrow> Among them, p is the number of pole pairs, ψ _f is the flux linkage of the permanent magnet, and i _q is the q-axis current; Or, in the form of expressing the electromagnetic torque by the motor constants K _t and i _q : <mrow><msub><mi>T</mi><mrow><mi>e</mi><mi>m</mi></mrow></msub><mo>=</mo><msub><mi>K</mi><mi>t</mi></msub><mo>&amp;CenterDot;</mo><mfrac><msub><mi>i</mi><mi>q</mi></msub><msqrt><mn>2</mn></msqrt></mfrac><mo>;</mo></mrow> Judging whether the motor is in the non-field-weakening area or the field-weakening area: use _id = 0 in the non-field-weakening area, and the amplitude of the output phase voltage at this time is expressed as follows, where ω is the electrical angular velocity; <mrow><mi>u</mi><mo>=</mo><mfrac><msqrt><mrow><msup><mrow><mo>(</mo><msub><mi>L</mi><mi>q</mi></msub><msub><mi>i</mi><mi>q</mi></msub><mo>)</mo></mrow><mn>2</mn></msup><mo>+</mo><msup><msub><mi>&amp;psi;</mi><mi>f</mi></msub><mn>2</mn></msup></mrow></msqrt><mi>&amp;omega;</mi></mfrac></mrow> In the field weakening area, the amplitude of the output phase voltage is expressed as follows, u _lim is the voltage limit, and V _DC is the DC bus voltage: <mrow><mi>u</mi><mo>=</mo><msub><mi>u</mi><mi>lim</mi></msub><mo>=</mo><mfrac><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi></mrow></msub><mn>2</mn></mfrac><mo>=</mo><mfrac><msqrt><mrow><msup><mrow><mo>(</mo><msub><mi>L</mi><mi>q</mi></msub><msub><mi>i</mi><mi>q</mi></msub><mo>)</mo></mrow><mn>2</mn></msup><mo>+</mo><msup><mrow><mo>(</mo><msub><mi>L</mi><mi>d</mi></msub><msub><mi>i</mi><mi>d</mi></msub><mo>+</mo><msub><mi>&amp;psi;</mi><mi>f</mi>mi></msub><mo>)</mo></mrow><mn>2</mn></msup></mrow></msqrt><mi>&amp;omega;</mi></mfrac><mo>.</mo></mrow> After obtaining id and iq, the output of the three phases A, B, and C at the corresponding torque speed is obtained through constant amplitude transformation. Among them, the change of dq axial αβ rotating coordinate system applies Park transformation, which has the following conversion relationship: <mrow><mfenced open = "[" close = "]"><mtable><mtr><mtd><msub><mi>i</mi><mi>&amp;alpha;</mi></msub></mtd></mtr><mtr><mtd><msub><mi>i</mi><mi>&amp;beta;</mi></msub></mtd></mtr></mtable></mfenced><mo>=</mo><mfenced open = "[" close = "]"><mtable><mtr><mtd><mrow><mi>c</mi><mi>o</mi><mi>s</mi><mi>&amp;theta;</mi></mrow></mtd><mtd><mrow><mo>-</mo><mi>s</mi><mi>i</mi><mi>n</mi><mi>&amp;theta;</mi></mrow></mtd></mtr><mtr><mtd><mrow><mi>s</mi><mi>i</mi><mi>n</mi><mi>&amp;theta;</mi></mrow></mtd><mtd><mrow><mi>cos</mi><mi>&amp;theta;</mi></mrow></mtd></mtr></mtable></mfenced><mo>&amp;CenterDot;</mo><mfenced open = "[" close = "]"><mtable><mtr><mtd><msub><mi>i</mi><mi>d</mi></msub></mtd></mtr><mtr><mtd><msub><mi>i</mi><mi>q</mi></msub></mtd></mtr></mtable></mfenced></mrow> Among them, θ is the phase angle; The transformation from the αβ rotating coordinate system to the three-phase A, B, and C is Clark transformation, and the transformation relationship is as follows: <mrow><mfenced open = "[" close = "]"><mtable><mtr><mtd><msub><mi>i</mi><mi>A</mi></msub></mrow>mtd></mtr><mtr><mtd><msub><mi>i</mi><mi>B</mi></msub></mtd></mtr><mtr><mtd><msub><mi>i</mi><mi>C</mi></msub></mtd></mtr></mtable></mfenced><mo>=</mo><msqrt><mfrac><mn>2</mn><mn>3</mn></mfrac></msqrt><mfenced open = "[" close = "]"><mtable><mtr><mtd><mn>1</mn></mtd><mtd><mn>0</mn></mtd></mtr><mtr><mtd><mrow><mo>-</mo><mfrac><mn>1</mn><mn>2</mn></mfrac></mrow></mtd><mtd><mfrac><msqrt><mn>3</mn></msqrt><mn>2</mn></mfrac></mtd></mtr><mtr><mtd><mrow><mo>-</mo><mfrac><mn>1</mn><mn>2</mn></mfrac></mrow></mtd><mtd><mrow><mo>-</mo><mfrac><msqrt><mn>3</mn></msqrt><mn>2</mn></mfrac></mrow></mtd></mtr></mtable></mfenced><mo>&amp;CenterDot;</mo><mfenced open = "[" close = "]"><mtable><mtr><mtd><msub><mi>i</mi><mi>d</mi></msub></mtd></mtr><mtr><mtd><msub><mi>i</mi><mi>q</mi></msub></mtd></mtr></mtable></mfenced></mrow> Among them, i _A , i _B , and i _C are the currents of each phase respectively. 3. The method according to claim 2, characterized in that: the loss calculation model of setting up the IGBT power module described in the step 3 specifically includes: The loss P _Module of the IGBT power module includes: the on-state loss P _{IGBT_con generated during the IGBT operation, the turn-on loss P IGBT_on and} the turn-off loss P _{IGBT_off} during the switching transient state; the on-state loss P _{FWD_con} and the reverse recovery loss during the FWD operation P _{FWD_re} : P _Module ＝P _{IGBT_con} +P _{IGBT_on} +P _{IGBT_off} +P _{FWD_con} +P _{FWD_re} The on-state voltage drops V _CE and V _D when the IGBT and FWD are turned on are composed of two parts: the respective threshold voltages V _CEO , V _DO and the voltage drops generated by the on-state resistances R _ch and R _d , and are related to the actual temperature T , the relationship is expressed as follows: T ₀ is the reference temperature, I _C and I _D are the currents passing through the IGBT and FWD respectively, where b _T , m and b _D are temperature-related items that can be obtained by curve fitting; <mrow><msub><mi>V</mi><mrow><mi>C</mi><mi>E</mi></mrow></msub><mrow><mo>(</mo><mi>T</mi><mo>)</mo></mrow><mo>=</mo><msub><mi>V</mi><mrow><mi>C</mi><mi>E</mi><mi>O</mi></mrow></msub><mrow><mo>(</mo><msub><mi>T</mi><mn>0</mn></msub><mo>)</mo></mrow><mo>+</mo><msub><mi>b</mi><mi>T</mi></msub><mo>&amp;CenterDot;</mo><mrow><mo>(</mo><mi>T</mi><mo>-</mo><msub><mi>T</mi><mn>0</mn></msub><mo>)</mo></mrow><mo>+</mo><msub><mi>R</mi><mrow><mi>c</mi><mi>h</mi></mrow></msub><mrow><mo>(</mo><msub><mi>T</mi><mn>0</mn></msub><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><msup><mrow><mo>(</mo><mfrac><mi>T</mi><msub><mi>T</mi><mn>0</mn></msub></mfrac><mo>)</mo></mrow><mi>m</mi></msup><mo>&amp;CenterDot;</mo><msup><msub><mi>I</mi><mi>C</mi></msub><mn>2</mn></msup></mrow> V _D (T)＝V _DO (T ₀ )+b _D ·(TT ₀ )+R _d (T)·I _D ² Calculate the on-state loss, where D _T and D _D are the duty cycle of IGBT and FWD in the unit switching period: P _{IGBT_con} ＝(V _CEO (T)·I _C +R _ch (T)I _C ² )·D _T P _{FWD_con} ＝(V _DO (T)I _D +R _d (T)I _D ² )·D _D When the switching frequency is f _sw , the turn-on power loss P _{IGBT_on} of the IGBT, the turn-off power loss P _{IGBT_off} and the reverse recovery power loss P _{FWD_re} of the FWD can be expressed as follows: P _{IGBT_on} = f _sw E _{IGBT_on} P _{IGBT_off} = f _sw E _{IGBT_off} P _{FWD_re} = f _sw · E _{FWD_re} ; in, <mrow><msub><mi>E</mi><mrow><mi>I</mi><mi>G</mi><mi>B</mi><mi>T</mi><mo>_</mo><mi>o</mi><mi>n</mi></mrow></msub><mo>=</mo><mrow><mo>(</mo><msub><mi>a</mi><mrow><mi>o</mi><mi>n</mi></mrow></msub><msup><msub><mi>I</mi><mi>c</mi></msub><mn>2</mn></msup><mo>+</mo><msub><mi>b</mi><mrow><mi>o</mi><mi>n</mi></mrow></msub><msub><mi>I</mi><mi>c</mi></msub><mo>+</mo><msub><mi>c</mi><mrow><mi>o</mi><mi>n</mi></mrow></msub><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><msub><mi>k</mi><mrow><mi>o</mi><mi>n</mi></mrow></mrow></mrow>msub><mrow><mo>(</mo><mi>T</mi><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><mfrac><mrow><msub><mi>E</mi><mrow><mi>o</mi><mi>n</mi></mrow></msub><mrow><mo>(</mo><msub><mi>R</mi><mi>g</mi></msub><mo>)</mo></mrow></mrow><mrow><msub><mi>E</mi><mrow><mi>o</mi><mi>n</mi></mrow></msub><mrow><mo>(</mo><msub><mi>R</mi><mrow><mi>r</mi><mi>a</mi><mi>t</mi><mi>e</mi><mi>d</mi></mrow></msub><mo>)</mo></mrow></mrow></mfrac><mo>&amp;CenterDot;</mo><mfrac><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi></mrow></msub><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi><mo>_</mo><mi>R</mi><mi>a</mi><mi>t</mi><mi>e</mi><mi>d</mi></mrow></msub></mfrac></mrow> <mrow><msub><mi>E</mi><mrow><mi>I</mi><mi>G</mi><mi>B</mi><mi>T</mi><mo>_</mo><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><mo>=</mo><mrow><mo>(</mo><msub><mi>a</mi><mrow><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><msup><msub><mi>I</mi><mi>c</mi></msub><mn>2</mn></msup><mo>+</mo><msub><mi>b</mi><mrow><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><msub><mi>I</mi><mi>c</mi></msub><mo>+</mo><msub><mi>c</mi><mrow><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><msub><mi>k</mi><mrow><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><mrow><mo>(</mo><mi>T</mi><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><mfrac><mrow><msub><mi>E</mi><mrow><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><mrow><mo>(</mo><msub><mi>R</mi><mi>g</mi></msub><mo>)</mo></mrow></mrow><mrow><msub><mi>E</mi><mrow><mi>o</mi><mi>f</mi><mi>f</mi></mrow></msub><mrow><mo>(</mo><msub><mi>R</mi><mrow><mi>r</mi><mi>a</mi><mi>t</mi><mi>e</mi><mi>d</mi></mrow></msub><mo>)</mi>mo></mrow></mrow></mfrac><mo>&amp;CenterDot;</mo><mfrac><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi></mrow></msub><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi><mo>_</mo><mi>R</mi><mi>a</mi><mi>t</mi><mi>e</mi><mi>d</mi></mrow></msub></mfrac></mrow> <mrow><msub><mi>E</mi><mrow><mi>F</mi><mi>W</mi><mi>D</mi><mo>_</mo><mi>r</mi><mi>e</mi></mrow></msub><mo>=</mo><mrow><mo>(</mo><msub><mi>a</mi><mrow><mi>r</mi><mi>e</mi></mrow></msub><msup><msub><mi>I</mi><mi>D</mi>mi></msub><mn>2</mn></msup><mo>+</mo><msub><mi>b</mi><mrow><mi>r</mi><mi>e</mi></mrow></msub><msub><mi>I</mi><mi>D</mi></msub><mo>+</mo><msub><mi>c</mi><mrow><mi>r</mi><mi>e</mi></mrow></msub><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><msub><mi>k</mi><mrow><mi>r</mi><mi>e</mi></mrow></msub><mrow><mo>(</mo><mi>T</mi><mo>)</mo></mrow><mo>&amp;CenterDot;</mo><mfrac><mrow><msub><mi>E</mi><mrow><mi>r</mi><mi>e</mi></mrow></msub><mrow><mo>(</mo><msub><mi>R</mo>mi><mi>g</mi></msub><mo>)</mo></mrow></mrow><mrow><msub><mi>E</mi><mrow><mi>r</mi><mi>e</mi></mrow></msub><mrow><mo>(</mo><msub><mi>R</mi><mrow><mi>r</mi><mi>a</mi><mi>t</mi><mi>e</mi><mi>d</mi></mrow></msub><mo>)</mi>mo></mrow></mrow></mfrac><mo>&amp;CenterDot;</mo><mfrac><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi></mrow></msub><msub><mi>V</mi><mrow><mi>D</mi><mi>C</mi><mo>_</mo><mi>R</mi><mi>a</mi><mi>t</mi><mi>e</mi><mi>d</mi></mrow></msub></mfrac></mi>mrow> Among them, E _{IGBT_on} is the turn-on energy loss of IGBT, E _{IGBT_off} is the turn-off energy loss of IGBT, E _{FWD_re} is the reverse recovery energy loss of FWD, a _on , b _on , c _on , a _off , b _off , c _off , a _re , b _re , _cre are fitting constants, k _on , k _off , k _re are temperature-related items, E _on (R _g ), E _off (R _g ), E _off (R _g ) and E _on ( R _rated ), E _off (R _rated ), E _off (R _rated ) are the turn-on and turn-off energy consumption of the corresponding IGBT under the actual gate resistance and the reference gate resistance, and the reverse recovery energy loss of FWD, V _{DC_rated} is the reference DC bus value. 4. The method according to claim 3, characterized in that: further, the establishment of the thermal resistance network model of the IGBT power module described in step 4 specifically includes: the model is based on the following assumptions: (1) Neglecting the influence of heat radiation and heat convection, the form of heat transfer in the module is heat conduction: (2) Due to the filling of insulating silica gel, the heat transfer path is from the chip to the substrate: (3) The influence of thermal coupling between chips is negligible: (4) Neglecting the local temperature difference of a single chip, using the lumped parameter method; Based on the above assumptions, a fourth-order Foster thermal resistance network model is established. 5. The method according to claim 4, characterized in that: the junction temperature corresponding to the current motor operating point is calculated by the following method: assuming that the bottom surface temperature of the copper substrate is constant as T _C , the IGBT and FWD are connected from the respective chips to the bottom case The thermal resistances are R _{jc_IGBT} and R _{jc_FWD} , and the junction temperature T _{j_IGBT} and T _{j_FWD} of the IGBT and the anti-parallel diode FWD can be expressed as follows: T _{j_IGBT} ＝T _c +R _{jc_IGBT} P _IGBT T _{j_FWD} ＝T _c +R _{jc_FWD} ·P _FWD Wherein, P _IGBT includes P _{IGBT_con} , P _{IGBT_on} and P _{IGBT_off} , and P _FWD includes P _{FWD_con} and P _{FWD_rr} .