CN107835035B - Low signal-to-noise ratio short frame burst communication open-loop demodulation method and device - Google Patents

Low signal-to-noise ratio short frame burst communication open-loop demodulation method and device Download PDF

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CN107835035B
CN107835035B CN201711041223.2A CN201711041223A CN107835035B CN 107835035 B CN107835035 B CN 107835035B CN 201711041223 A CN201711041223 A CN 201711041223A CN 107835035 B CN107835035 B CN 107835035B
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frequency offset
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CN107835035A (en
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罗士荀
岳平越
王帅
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Beijing Institute of Technology BIT
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
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    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/707—Spread spectrum techniques using direct sequence modulation
    • H04B1/7073—Synchronisation aspects
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/22—Demodulator circuits; Receiver circuits
    • H04L27/227—Demodulator circuits; Receiver circuits using coherent demodulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
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    • H04L2027/0026—Correction of carrier offset

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Abstract

The invention discloses a method and a device for demodulating open loop of low signal-to-noise ratio short frame burst communication, belonging to the technical field of direct sequence spread spectrum communication demodulation. The invention adopts a segmented data processing mode, firstly de-modulates the de-spread data, then carries out zero filling FFT, estimates the frequency deviation change frequency and frequency deviation and carries out compensation. And accurately estimating and compensating each section of frequency deviation by using an exhaustion method, and deblurring the phase by using the polarity of the frame header to finish correct demodulation of data. Compared with the prior art, the invention overcomes the defects of phase ambiguity, low convergence speed and the like of the traditional phase-locked loop coherent demodulation device, can realize the quick search and compensation of frequency deviation and frequency deviation change rate, completes phase identification and phase correction, and ensures that a direct sequence spread spectrum signal can be normally demodulated under the conditions of short time and low signal-to-noise ratio.

Description

Low signal-to-noise ratio short frame burst communication open-loop demodulation method and device
Technical Field
The invention relates to a carrier phase tracking and coherent demodulation method (hereinafter referred to as a demodulation device), in particular to a short frame burst communication open-loop demodulation method and a short frame burst communication open-loop demodulation device with low signal to noise ratio, which are suitable for open-loop search, tracking and symbol coherent demodulation of carrier frequency offset and frequency offset change rate of a short frame burst direct sequence spread spectrum system in high-dynamic and low-signal to noise ratio environments and the like, and belong to the technical field of demodulation in direct sequence spread spectrum communication.
Background
The communication between the high-speed aircrafts and the ground equipment or the high-speed aircrafts has the characteristics of long communication distance, large link attenuation and low signal-to-noise ratio, so a direct sequence spread spectrum communication system is usually adopted; and the communication is often required to ensure the communication concealment and have the anti-detection capability, so that a short frame burst system is adopted. Because the aircraft has high motion speed and complex motion characteristics, first-order or even high-order acceleration exists in relative radial motion between a transmitter and a receiving end on a communication link, on one hand, the high-order acceleration can be ignored due to a short frame burst communication system and short frame communication time, but on the other hand, the receiver receives signals with large Doppler frequency offset and non-negligible Doppler first-order frequency offset change rate (hereinafter referred to as frequency offset change rate); because the signal-to-noise ratio of the received signal is very low, a coherent demodulation mode is adopted, and if a phase-locked loop mode is adopted to realize the tracking of carrier Doppler frequency offset and frequency offset change rate, under a short frame burst communication system, a loop can not be converged timely, so that data can not be demodulated normally, and therefore an open loop search mode with higher search speed and shorter time is adopted to finish coherent demodulation.
A general direct sequence spread spectrum receiver mainly includes an acquisition module, a despreading module, a demodulation module, a decoding module, and the like. The acquisition module is used for acquiring frequency offset and phase information of a received signal and sending the information to the de-spreading module; the despreading module recovers data before spreading by using the captured information; the demodulation module recovers baseband information before modulation by using despread output data; the decoding module completes decoding by utilizing the output of the demodulation module.
Assume that the spreading code of the direct sequence spread spectrum communication receiver is c [ L ], L is 0,1,2, …, L-1, and L is the spreading code length. And c (l) is 0 or 1, using radio technology convention, where c [ l ] is 0 to represent logic positive, c [ l ] is 1 to represent logic negative, and root raised cosine pulse shaping is used, where impulse function of pulse shaping filter is h (t), and baseband waveform expression after shaping is:
Figure GDA0002341816850000021
wherein, b [ m ]]For baseband data symbols, with a period Ts＝LTc，TcThe short frame burst communication system is a chip period and generally has certain requirements on a burst frame structure; for acquisition, the frame communication starts with the pilot signal transmission, i.e. b m]Is always 0. After the pilot signal is over, bm]The M sequence with fixed length or the frame header of Gold code, and the data segment after the frame header is finished. After modulation, the signals emitted by the radio frequency are:
Ssrnd(t)＝cos(2πfRFt)Sb，s(t)
in the above formula, fRFIs a radio frequencyRate, and for convenience of expression, it is assumed that the initial phase of the carrier and the initial phase of the code of the transmission signal are both 0.
Through Doppler effect and signal delay in the wireless transmission process, the signal reaches a receiver, and I/Q two-path orthogonal down-conversion is carried out on a radio frequency signal to obtain a complex baseband signal:
Figure GDA0002341816850000022
in the above formula fβAnd fσRespectively, the Doppler frequency offset change rate and the Doppler frequency offset generated by the relative radial acceleration motion of both sides of the communication link; tau is0Is the code phase delay. And completing two-dimensional search of Doppler frequency offset and pseudo code phase by an acquisition module of the receiver. Wherein, "frequency offset" and "initial frequency offset" are two different concepts once the frequency offset change rate is considered. By default, "frequency offset" refers to instantaneous frequency offset; however, for the situation with frequency offset change rate, the instantaneous frequency offset is a time-varying function, and only the "initial frequency offset" (the instantaneous frequency offset at the beginning of frame burst communication) is constant. The frequency deviation obtained by the capturing device is instantaneous frequency deviation.
The despreading module compensates the input data of the despreading module on a frequency domain by using the rough frequency offset estimation obtained by the capturing device, but the frequency offset change rate is not compensated, and the compensation result is;
Figure GDA0002341816850000023
the upper type
Figure GDA0002341816850000024
For the coarse estimation of the frequency offset obtained by the acquisition module, r n]Is r (t) a digitized expression. To rde_In[n]Carrying out de-spreading;
Figure GDA0002341816850000025
and performing related operation, namely despreading on input data by using the periodic characteristic of the spreading code. Only when rde_In[n]And y [ n ]]When the sequences are identical and their phases are identical to each other, the correlation value is maximized, and d [ n ] is]And updating according to the symbol rate for despreading output results. Considering the effects of the rate of change of frequency offset and residual frequency offset, the despread data output can be written as:
Figure GDA0002341816850000026
in the above equation, β is the digitized rate of change of frequency offset normalized to the symbol rate;
Figure GDA0002341816850000031
ωσnormalization of digital frequency, R, with respect to chip rate for obtaining coarse estimate of frequency offset for acquisition modulecIs the chip rate, Rc＝RsL. Representing the initial (digital) frequency offset of the residual carrier, assuming an initial phase of
Figure GDA0002341816850000032
bnE 0/1 represents the transmitted binary symbols (BPSK modulation); the demodulation module is used for completing frequency offset change rate, frequency offset search, compensation and phase identification, and a demodulated symbol is recorded as z (n), and the expression is as follows:
Figure GDA0002341816850000033
order to
Figure GDA0002341816850000034
Figure GDA0002341816850000035
If β, Δ ω and
Figure GDA0002341816850000036
correctly estimate and compensate, i.e.
Figure GDA0002341816850000037
At this time, the demodulation result
Figure GDA0002341816850000038
The traditional BPSK coherent demodulation module usually uses a phase-locked loop, such as a square loop, a Costas loop, etc., to realize carrier tracking, but cannot effectively solve the problem of "inverted pi", that is, a recovered local carrier may be in phase with or out of phase with a required coherent carrier, and this phase uncertainty will cause that demodulated digital baseband signals are just opposite, and all determined digital symbols are wrong.
Disclosure of Invention
The device aims to overcome the technical defects that the traditional phase-locked loop coherent demodulation device is mainly phase-fuzzy and low in convergence speed, and provides a short frame burst communication open-loop demodulation method and device with a low signal-to-noise ratio, so that frequency offset and frequency offset change rate search and compensation are realized, phase identification and phase correction are completed, and a direct sequence spread spectrum signal can be normally demodulated under the conditions of short time and low signal-to-noise ratio.
The low signal-to-noise ratio short frame burst communication open-loop demodulation method and device comprises a low signal-to-noise ratio short frame burst communication open-loop demodulation device, which is called the device for short; and a short frame burst communication open-loop demodulation method with low signal-to-noise ratio, which is called the method for short.
The device comprises a two-dimensional searching module, a fine searching module 1, a fine searching module 2, a fine searching module 3 and a frame synchronization module;
the fine searching module 1, the fine searching module 2 and the fine searching module 3 are multiplexed, and 3 multiplexed fine searching modules in the device are completely the same;
the input data of the device is the output of the de-spread module, and the output data of the device is the demodulation result and is sent to the decoding module;
the connection relation of each module of the device is as follows:
the two-dimensional searching module is connected with the fine searching module 1; the fine search module 1 is connected with the fine search module 2, the fine search module 2 is connected with the fine search module 3, and the fine search module 3 is connected with the frame synchronization module;
the functions of each module in the device are as follows:
the two-dimensional searching module has the functions of performing two-dimensional searching, finishing joint estimation on the initial frequency offset and the frequency offset change rate of the received signal to obtain the estimation of the initial frequency offset and the frequency offset change rate, and compensating the Doppler frequency offset change rate and the Doppler frequency offset; the fine searching module has the functions of completing the estimation and compensation of frequency offset, phase estimation and phase compensation; the function of the frame synchronization module is to resolve phase ambiguities.
The method is realized by the following technical scheme:
the device adopts a method for processing the input data by the subsection data, and comprises the following steps:
the method comprises the following steps that firstly, a two-dimensional searching module of Doppler frequency offset change rate and Doppler frequency offset change rate is operated based on input data of the device, joint estimation of initial frequency offset and frequency offset change rate of a received signal is completed, and estimation of the initial frequency offset and the frequency offset change rate is obtained;
the input data of the device is expressed as the following formula (1) and is also called as first-stage data;
Figure GDA0002341816850000041
in the above formula (1), Δ ω represents a residual carrier initial frequency offset, which is a digital frequency offset, also called frequency offset;
Figure GDA0002341816850000042
is the initial phase; bnE0/1 represents the transmitted binary symbol, β is the digital frequency deviation change rate, which is abbreviated as frequency deviation change rate, is exponential function;
maximum likelihood estimation of β and Δ ω in equation (1) in white Gaussian noise environment
Figure GDA0002341816850000043
And
Figure GDA0002341816850000044
equivalent to the solution of the two-dimensional optimization problem in the following equation (2):
Figure GDA0002341816850000045
approximate solution can be carried out through two-dimensional grid search, all uncertain ranges of frequency deviation and frequency deviation change rate are divided into two-dimensional plane grids according to certain precision, and the mode of obtaining a two-dimensional plane is as follows: for the frequency deviation change rate, taking Q in the change range
Figure GDA0002341816850000046
Namely, it is
Figure GDA0002341816850000047
According to
Figure GDA0002341816850000048
For x [ n ]]Carrying out Q times of 'de-chirp modulation', short for 'de-chirp', corresponding symbol xq[n]Expressed as the following formula (3):
Figure GDA0002341816850000051
for xq[n]Zero filling is carried out until the length is K points, and then K-point FFT operation is carried out to obtain xq[k](ii) a For xq[k]Modulus is taken, the result is | xq[k]L, |; then, the modulus result | x is obtained from all Q × Kq[k]Finding the maximum and from the position of this maximum on the two-dimensional grid, β and Δ ω estimates are obtained
Figure GDA0002341816850000052
And
Figure GDA0002341816850000053
wherein,
Figure GDA0002341816850000054
representing the result of the step of estimating the rate of change of the frequency offset,
Figure GDA0002341816850000055
representing the estimation result of the pair of initial frequency offsets;
the first segment of data is aimed at obtaining the frequency deviation change frequency and estimation of frequency deviation, and has no need of making subsequent operation on said segment of data, then the baseband waveform of (N +1) symbol is fed into dispreading module, and according to the above-mentioned data it can obtain the estimation result
Figure GDA0002341816850000056
And
Figure GDA0002341816850000057
compensating the frequency deviation change frequency and the frequency deviation of the input data;
and starting from the second section of data, the compensation of the frequency deviation change rate uses the estimation result of the frequency deviation change rate obtained by the first section of data, and the default frequency deviation change rate is compensated. Even if the frequency offset change rate estimation result obtained in the first step has errors, namely residual Doppler frequency offset change rate exists, as long as the processing length and Q value of the second section of data are proper, the influence of the residual Doppler frequency offset change rate on frequency offset estimation can be completely ignored;
each segment of data only needs to estimate and compensate the frequency offset, so that the demodulation of the data is completed;
secondly, operating a fine search module based on the estimation result of the initial frequency offset obtained by the first section of data, and performing compensation based on the accurate estimation result of the frequency offset obtained by the second section of data so as to complete data demodulation;
step two, specifically:
step 2.1, based on the compensation of the estimated value of the initial frequency offset in the step one, obtains the second section of data of formula (4):
Figure GDA0002341816850000058
wherein, Δ ω2Is the frequency offset of the second segment data,
Figure GDA0002341816850000059
Is the initial phase of the second segment of data, and M is the processing length of the segment of data; in order to simplify mathematical derivation, the subscript N +1 of the second segment of data is changed into N which is more than or equal to 0 and less than or equal to M 1 by using a conversion method;
step 2.2, operating a fine search module to obtain accurate estimation of frequency offset;
the fine searching module of the device is operated, and the fine searching module adopts an exhaustion method to obtain the accurate estimation of the frequency deviation, so as to obtain the coherent demodulation result of the second section of data, and the method specifically comprises the following steps:
step 2.21, the second section of data (4) is subjected to square demodulation, and then an inner product is formed by the second section of data and the square of a local inner product template, wherein the expression of the local inner product template is as shown in a formula (5):
Figure GDA0002341816850000061
the specific method for performing inner product with the square of the local inner product template is to carry out inner product on the delta omega2Divided into P small sections at equal intervals and using square of P complex sine waves with different digital frequencies
Figure GDA0002341816850000062
And x2[n]Inner product, and taking the maximum module value of the inner product as the complex sine wave with digital frequency of-2 delta omega2Based on an estimate of
Figure GDA0002341816850000063
For compensating for frequency offset; phase position
Figure GDA0002341816850000064
Estimated according to the following equation (6):
Figure GDA0002341816850000065
the divide by 2 operation in equation (6) above results in phase ambiguity, i.e.
Figure GDA0002341816850000066
And
Figure GDA0002341816850000067
all satisfy the requirement of formula (6) because
Figure GDA0002341816850000068
The initial value is unknown, here
Figure GDA0002341816850000069
Get
Figure GDA00023418168500000610
Or
Figure GDA00023418168500000611
All can be used; when the data is processed to the frame head section, the absolute phase is determined by utilizing the polarity of the correlation result of the frame head, and the absolute phase is not taken in advance
Figure GDA00023418168500000612
The second stage data coherent demodulation result is obtained as the following formula (7):
Figure GDA00023418168500000613
step three, accurately estimating and compensating the frequency offset of the data of the section, and performing ambiguity resolution on the phase of the data of the section, namely the phase of the data of the section is required to be consistent with that of the previous section;
step three, specifically:
the third section of data is:
Figure GDA00023418168500000614
the data in this section are:
Figure GDA00023418168500000615
when the step three is operated for the first time, the data of the section refers to a third section of data, and the rest is done in sequence;
step 3.1: operating a fine search module to obtain accurate estimation of frequency offset; the fine search module adopts an exhaustion method to obtain accurate estimation of frequency deviation, and then obtains a coherent demodulation result of the data, specifically:
carrying out square demodulation on the data of the section, and then carrying out inner product on the data and the square of a local inner product template, wherein the expression of the local inner product template is as shown in a formula (5); the specific method for performing inner product with the square of the local inner product template is to carry out inner product on the delta omega2Divided into P small sections at equal intervals and using square of P complex sine waves with different digital frequencies
Figure GDA00023418168500000616
And x2[n]Inner product, and taking the maximum module value of the inner product as the complex sine wave with digital frequency of-2 delta omega3Based on an estimate of
Figure GDA0002341816850000071
For compensating for frequency offset; phase position
Figure GDA0002341816850000072
Estimated according to the following equation (9):
Figure GDA0002341816850000073
the divide by 2 operation in equation (9) above results in phase ambiguity, i.e.
Figure GDA0002341816850000074
And
Figure GDA0002341816850000075
satisfies the requirement of formula (9), namely: deriving a frequency offset estimate
Figure GDA0002341816850000076
Obtaining phase simultaneously
Figure GDA0002341816850000077
And
Figure GDA0002341816850000078
step 3.2: the phase output in the step 3.1 is deblurred, and a reference phase for deblurring is constructed according to the result of the previous section of frequency offset and the initial phase, because the last phase of the previous section is the initial phase of the current section;
if the second segment of initial phase estimation results in
Figure GDA0002341816850000079
The frequency offset estimation result is
Figure GDA00023418168500000710
The end phase should be (9):
Figure GDA00023418168500000711
this is because the last phase and the initial phase are separated by N-1 symbols; if the despreading process of the spread spectrum system is considered, the initial phase of the third segment and the final phase of the second segment are slightly different, and the following should be done (10):
Figure GDA00023418168500000712
because the second section has phase ambiguity, the third section and the following sections only need to keep a section of continuous phase strictly following the phase of the previous section; the specific treatment process is as follows (11):
Figure GDA00023418168500000713
judgment of
Figure GDA00023418168500000714
And
Figure GDA00023418168500000715
which is closer to
Figure GDA00023418168500000716
If:
Figure GDA00023418168500000717
then get
Figure GDA00023418168500000718
As an estimation result of the initial phase; otherwise, get
Figure GDA00023418168500000719
Then coherently demodulating according to the data of the third segment and according to
Figure GDA00023418168500000720
And compensating the frequency offset, wherein the output of the third stage of coherent demodulation is as follows:
Figure GDA00023418168500000721
step four, based on the subsequent data segment, repeating the step three until the pilot frequency segment data processing is finished; the phase estimation result obtained can keep consistent with the phase of the previous section of data;
step five, step one to step four after the pilot frequency section data processing, enter the data processing of the frame head section, the concrete processing mode is to take the correlation to the frame head data, obtain the frame head correlation result;
wherein, the frame header is usually M sequence or Gold code sequence, and the frame header is the beginning of the data segment;
the frame head outputs data z [ n ] through de-spreading, frequency estimation, compensation and phase estimation, and the correlation operation of the following formula (14) is carried out with a local frame head template:
Figure GDA00023418168500000722
the correlation process of equation (14) is the same as the operation of the despreading process, except that in despreadingIs a correlation operation at the chip level, and the despreading is a time length of one symbol for the correlation length; frame synchronization is a correlation operation performed at the symbol level, and the length of the correlation operation depends on the length of a frame header; r isheader[n]The frame head correlation result is obtained; sheader_tep[n]A frame header local template; m is the length of the frame header;
step six, the frame head correlation result is correlated with the local template, whether the frame synchronization is successful is determined according to whether the absolute value of the correlation result exceeds a preset threshold, and corresponding operation is carried out, specifically: correlation operation result r to frame headerheader[n]Modulus operation, if modulus result exceeds threshold, considering frame synchronization success, jumping to step seven, starting decoding device; otherwise, the data of the frame is considered to be failed to be received, and the step one is returned to wait for the coming of the next frame data;
seventhly, when the frame synchronization is successful, the phase is deblurred according to the polarity of the correlation result; the method specifically comprises the following steps: according to the fact that in order to solve the problem of 'falling pi', when the time exceeds the threshold, the positive and negative of a related result when the threshold is exceeded need to be confirmed; if the polarity is positive, the phase kept at the moment is 0, and z [ n ] is directly output to the decoding device; if the polarity of the correlation result is negative, the phase kept at the moment is pi, the z [ n ] needs to be inverted, and then the inverted z [ n ] is sent to a decoding module; after the decoding is finished, the data of the frame is successfully received, and the device returns to the first step to wait for the arrival of the next frame of data;
and completing the open-loop demodulation method of the short frame burst communication with the low signal-to-noise ratio from the first step to the seventh step.
Advantageous effects
Compared with the prior art, the method and the device for demodulating the burst communication open loop of the short frame with the low signal-to-noise ratio have the following beneficial effects:
1. accurate estimation of Doppler frequency offset and frequency offset change rate can be completed in a short time by using methods such as demodulation line and two-dimensional search;
2. finely estimating and compensating each section of frequency deviation by using an exhaustion method through a sectional processing mode to realize carrier frequency tracking and complete phase identification;
3. the method corrects the phase by using the polarity of the frame header, ensures that data information can be correctly demodulated, and is suitable for open-loop search, tracking and symbol coherent demodulation of carrier frequency offset and frequency offset change rate of a short frame burst direct sequence spread spectrum system in the environments of high dynamics, low signal-to-noise ratio and the like.
Drawings
FIG. 1 is a schematic block diagram of an open-loop demodulation method and apparatus for burst communication of short frames with low SNR according to the present invention and the apparatus in embodiment 1;
FIG. 2 is a diagram of a two-dimensional search plane of frequency offset and frequency offset change rate according to an embodiment of the method and apparatus for open-loop demodulation of short frame burst communication with low SNR;
FIG. 3 is a schematic diagram illustrating the demodulation principle in the first step of the method and apparatus for open-loop demodulation of burst communication of low SNR short frame according to the present invention;
fig. 4 is a schematic diagram of frequency fine search in the method and apparatus for open loop demodulation of short frame burst communication with low snr according to the present invention.
Detailed Description
The device is described in further detail below with reference to examples and figures.
This example takes a direct sequence spread spectrum BPSK communication receiver as an example, and the baseband digital signal processing part of the BPSK communication receiver is shown in fig. 1.
As can be seen from fig. 1, despreading data enters the apparatus, the doppler frequency shift change rate and doppler frequency shift two-dimensional search module, the fine search module, and the frame synchronization module of the apparatus are sequentially operated to complete the demodulation function, and the data is output to the decoding module of the receiver to complete decoding.
The device inputs data parameter indexes: symbol rate Rs2.5 ksps; initial frequency offset Δ f range
Figure GDA0002341816850000091
In Hz. The frequency deviation change rate delta f ranges +/-6 in kHz/s.
Inputting the despread data into the device, wherein the prior information of the data frequency offset and the frequency offset change rate is-130.9140 Hz; Δ Δ f ═ 3253.3 Hz/s. First segment data length128 symbols and all pilots (all "0" of baseband data). After entering a Doppler frequency offset change rate and Doppler frequency offset two-dimensional search module, dividing all uncertain ranges of initial frequency offset and frequency offset change rate into two-dimensional plane grids according to certain precision respectively, and firstly, dividing the two-dimensional plane grids according to certain precision
Figure GDA0002341816850000092
For x [ n ]]The demodulation is performed, as shown in fig. 2, the frequency offset change rate Δ Δ f search range ± 6kHz/s,
Figure GDA0002341816850000093
the search precision is 250Hz/s (the difference between two adjacent Q values is 250Hz/s), and the Q value is
Figure GDA0002341816850000094
Figure GDA0002341816850000095
The demodulation expression is
Figure GDA0002341816850000096
In the above formula
Figure GDA0002341816850000097
For 49 sets of demodulation results xq[n]Make up 128 zeros, do 256 FFT, get 49 groups xq[k]Each group xq[k]Length 256. All the modulus results | x of 49 × 256 are obtainedq[k]And | drawing on the grid, and obtaining the frequency offset and the frequency offset change rate of the signal according to the coordinate corresponding to the maximum value. Referring to FIG. 3, the obtained frequency deviation is-136.7188 Hz, and the frequency deviation change rate is-3000 Hz/s, i.e. the frequency deviation is converted into the frequency deviation estimator obtained in the digital domain
Figure GDA0002341816850000101
Rate of change of frequency offset
Figure GDA0002341816850000102
And respectively compensating the frequency deviation change frequency and the frequency deviation.
Starting from the second stage of data, the data processing length is 64, and the frequency deviation change rate is estimated by using the first stage frequency deviation change rate as a default
Figure GDA0002341816850000103
Compensating, even if estimated in the first stage
Figure GDA0002341816850000104
With an error of at most 500Hz/s, the frequency is accumulated as
Figure GDA0002341816850000105
If the frequency obtained by the frequency offset estimation of the section exceeds 12.8Hz, the frequency estimation of the section is considered to have errors, the previous frequency estimation result is maintained, and only the frequency offset estimated by each section of data is updated and compensated. As known from the knowledge of digital signal processing, the precision of participating in frequency offset compensation after the first stage of data processing is
Figure GDA0002341816850000106
An exhaustive approach is taken to obtain a more accurate estimate of the frequency offset starting from the second segment of data.
The data of the second section of data entering the fine searching module is as follows:
Figure GDA0002341816850000107
referring to FIG. 4, for the second segment of data x [ n ]]Square demodulation is carried out, and then an exhaustion method is used, namely 64 complex sine wave templates with different digital frequencies are used, and the frequency interval between two adjacent templates is
Figure GDA0002341816850000108
Different frequency template squares and x2[n]Inner products are obtained, 64 groups of inner product results are obtained, and the expression of the inner product results is as follows:
Figure GDA0002341816850000109
in the above formula
Figure GDA00023418168500001010
The maximum value of the inner product result is taken to estimate the digital frequency of the complex sine wave to be-2 delta omega2And according to the simulation result,
Figure GDA00023418168500001011
the value of (c) is maximum, when p is 28. So that the frequency of this segment of data is estimated as
Figure GDA00023418168500001012
Figure GDA00023418168500001013
The phase estimation is as follows:
Figure GDA00023418168500001014
division by 2 results in phase ambiguity, since the second segment of data has no data prior information, so the direct selection
Figure GDA00023418168500001015
And (4) finishing. Estimated from this data
Figure GDA00023418168500001016
And compensating the frequency offset. The demodulation result of this segment of data is
Figure GDA00023418168500001017
In a third section of data
Figure GDA00023418168500001018
And
Figure GDA00023418168500001019
the estimation method is the same as the second stage data estimation method, and the frequency offset estimation of the data of the section is obtained through a fine search module
Figure GDA00023418168500001020
Figure GDA00023418168500001021
Compensating for frequency offset
Figure GDA0002341816850000111
Starting from the third segment of data, the data needs to be deblurred to ensure that the phase is consistent with the phase of the previous segment.
The initial phase of the third section is
Figure GDA0002341816850000112
Figure GDA0002341816850000113
Then get
Figure GDA0002341816850000114
As the estimation result of the initial phase, then coherent demodulation is performed according to the data of the third segment, and the demodulation result of the data of the segment is as follows:
Figure GDA0002341816850000115
the processing flow of the fourth and the following sections is completely the same as that of the third section, when the frame synchronization is successful, the polarity of the correlation result is negative, and after the frame head is finished, the demodulation result of the data section needs to invert the z [ n ] symbol and output the inverted z [ n ] symbol to the decoding device to finish the decoding.
Test results
The error rate of the demodulation result of the data segment is counted, 10000 symbols and the signal-to-noise ratio
Figure GDA0002341816850000116
Under the condition, the bit error rate of 0.395 is obtained by simulation, and the theoryThe error rate is 0.375, which basically accords with the theoretical value.
The demodulation device provided by the invention can complete estimation and real-time compensation on the frequency deviation and the frequency deviation change rate in a short time under a lower signal-to-noise ratio, realize accurate tracking of the frequency deviation and the frequency deviation change rate of the signal by a receiver, complete phase identification and phase correction, and ensure that the error rate of the demodulated data conforms to a theoretical value.
In summary, the above description is only a preferred embodiment of the present invention, and is not intended to limit the scope of the present invention. Any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the protection scope of the present invention.

Claims (5)

1. The short frame burst communication open-loop demodulation device with low signal-to-noise ratio is characterized in that: the device comprises a Doppler frequency offset change rate and Doppler frequency offset change rate two-dimensional search module, a fine search module and a frame synchronization module;
the fine searching module is multiplexed, and the multiplexed fine searching modules in the device are completely the same;
the input data of the device is the output of the de-spread module, and the output data of the device is the demodulation result and is sent to the decoding module;
the connection relation of each module of the device is as follows:
the two-dimensional searching module is connected with the fine searching module 1; the fine search module 1 is connected with the fine search module 2, the fine search module 2 is connected with the fine search module 3, and the fine search module 3 is connected with the frame synchronization module;
the functions of each module in the device are as follows:
the two-dimensional search module for the Doppler frequency offset change rate and the Doppler frequency offset change rate has the functions of performing two-dimensional search, completing joint estimation on the initial frequency offset and the frequency offset change rate of the received signal to obtain estimation of the initial frequency offset and the frequency offset change rate, and compensating the Doppler frequency offset change rate and the Doppler frequency offset; the fine searching module has the functions of completing the estimation and compensation of frequency offset, phase estimation and phase compensation; the function of the frame synchronization module is to resolve phase ambiguities.
2. The apparatus for demodulating open loop of short burst communication with low snr according to claim 1, wherein the demodulation method of open loop of short burst communication with low snr is: the method comprises the following steps:
the method comprises the following steps that firstly, a two-dimensional searching module of Doppler frequency offset change rate and Doppler frequency offset change rate is operated based on input data of the device, joint estimation of initial frequency offset and frequency offset change rate of a received signal is completed, and estimation of the initial frequency offset and the frequency offset change rate is obtained;
the input data of the device is also called as first segment data;
secondly, operating the fine search module 1, the fine search module 2 and the fine search module 3 to obtain second-stage data based on an estimation result of the initial frequency offset obtained by the first-stage data, and performing compensation based on an accurate estimation result of the frequency offset obtained by the second-stage data so as to complete data demodulation;
step three, accurately estimating and compensating the frequency offset of the data of the section, and performing ambiguity resolution on the phase of the data of the section, namely the phase of the data of the section is required to be consistent with that of the previous section;
step four, based on the subsequent data segment, repeating the step three until the pilot frequency segment data processing is finished; the phase estimation result obtained can keep consistent with the phase of the previous section of data;
step five, step one to step four after the pilot frequency section data processing, enter the data processing of the frame head section, the concrete processing mode is to take the correlation to the frame head data, obtain the frame head correlation result;
wherein, the frame header is usually M sequence or Gold code sequence, and the frame header is the beginning of the data segment;
the frame head outputs data z [ n ] through de-spreading, frequency estimation, compensation and phase estimation, and the correlation operation of the following formula (14) is carried out with a local frame head template:
Figure FDA0002422322010000021
this disclosureThe correlation process of equation (14) is the same as the operation of the despreading process, except that the despreading is a correlation operation performed at the chip level, and the despreading is a time length of one symbol; frame synchronization is a correlation operation performed at the symbol level, and the length of the correlation operation depends on the length of a frame header; r isheader[n]The frame head correlation result is obtained; sheader_tep[n]A frame header local template; m is the length of the frame header;
step six, the frame head correlation result is correlated with the local template, whether the frame synchronization is successful is determined according to whether the absolute value of the correlation result exceeds a preset threshold, and corresponding operation is carried out, specifically: correlation operation result r to frame headerheader[n]Modulus operation, if modulus result exceeds threshold, considering frame synchronization success, jumping to step seven, starting decoding device; otherwise, the data of the frame is considered to be failed to be received, and the step one is returned to wait for the coming of the next frame data;
seventhly, when the frame synchronization is successful, the phase is deblurred according to the polarity of the correlation result; the method specifically comprises the following steps: according to the fact that in order to solve the problem of 'falling pi', when the time exceeds the threshold, the positive and negative of a related result when the threshold is exceeded need to be confirmed; if the polarity is positive, the phase kept at the moment is 0, and z [ n ] is directly output to the decoding device; if the polarity of the correlation result is negative, the phase kept at the moment is pi, the z [ n ] needs to be inverted, and then the inverted z [ n ] is sent to a decoding module; after the decoding is finished, the data of the frame is successfully received, and the device returns to the first step to wait for the arrival of the next frame of data;
and completing the open-loop demodulation method of the short frame burst communication with the low signal-to-noise ratio from the first step to the seventh step.
3. The open-loop demodulation method for short frame burst communication with low snr according to claim 2, characterized in that: in the first step, the input data of the device is expressed by the following formula (1):
Figure FDA0002422322010000022
in the above formula (1), Δ ω represents the residual carrier initial frequency offset, which is the residueThe carrier initial frequency offset is digital frequency offset, also called frequency offset;
Figure FDA0002422322010000023
is the initial phase; bnE is 0/1 representing the transmitted binary symbol, β is the digital frequency deviation change rate, e is exponential function;
maximum likelihood estimation of β and Δ ω in equation (1) in white Gaussian noise environment
Figure FDA0002422322010000024
And
Figure FDA0002422322010000025
equivalent to the solution of the two-dimensional optimization problem in the following equation (2):
Figure FDA0002422322010000026
specifically, approximate solution is carried out through two-dimensional grid search, all uncertain ranges of frequency deviation and frequency deviation change rate are divided into two-dimensional plane grids according to certain precision, and a two-dimensional plane is obtained in the following mode: for the frequency deviation change rate, taking Q in the change range
Figure FDA0002422322010000031
Namely, it is
Figure FDA0002422322010000032
According to
Figure FDA0002422322010000033
For x [ n ]]Carrying out Q times of 'de-chirp modulation', short for 'de-chirp', corresponding symbol xq[n]Expressed as the following formula (3):
Figure FDA0002422322010000034
for xq[n]To carry outZero filling is carried out until the length is K points, and then K-point FFT operation is carried out to obtain xq[k](ii) a For xq[k]Modulus is taken, the result is | xq[k]L, |; then, the modulus result | x is obtained from all Q × Kq[k]Finding the maximum and from the position of this maximum on the two-dimensional grid, β and Δ ω estimates are obtained
Figure FDA0002422322010000035
And
Figure FDA0002422322010000036
wherein,
Figure FDA0002422322010000037
representing the result of the step of estimating the rate of change of the frequency offset,
Figure FDA0002422322010000038
representing the estimation result of the pair of initial frequency offsets;
the first segment of data is aimed at obtaining the frequency deviation change frequency and estimation of frequency deviation, and has no need of making subsequent operation on said segment of data, then the baseband waveform of (N +1) symbol is fed into dispreading module, and according to the above-mentioned data it can obtain the estimation result
Figure FDA0002422322010000039
And
Figure FDA00024223220100000310
compensating the frequency deviation change frequency and the frequency deviation of the input data;
finishing joint estimation on the initial frequency offset and the frequency offset change rate of the received signal to obtain estimation of the initial frequency offset and the frequency offset change rate, starting from the second section of data, wherein the frequency offset change rate compensation uses the estimation result of the frequency offset change rate obtained by the first section of data, and the default frequency offset change rate is already compensated; even if the frequency offset change rate estimation result obtained in the first step has errors, namely residual Doppler frequency offset change rate exists, as long as the processing length and Q value of the second section of data are proper, the influence of the residual Doppler frequency offset change rate on frequency offset estimation can be completely ignored; and each section of data is only required to estimate and compensate the frequency offset, so that the data demodulation is completed.
4. The open-loop demodulation method for short frame burst communication with low snr according to claim 3, characterized in that: step two, specifically:
step 2.1, based on the compensation of the estimated value of the initial frequency offset in the step one, obtains the second section of data of formula (4):
Figure FDA00024223220100000311
wherein, Δ ω2Is the frequency offset of the second segment data,
Figure FDA00024223220100000312
Is the initial phase of the second segment of data, and M is the processing length of the segment of data; in order to simplify mathematical derivation, the subscript N +1 of the second section of data is changed into N is more than or equal to 0 and less than or equal to M-1 by using a conversion method;
step 2.2, operating a fine search module to obtain accurate estimation of frequency offset;
the fine searching module of the device is operated, and the fine searching module adopts an exhaustion method to obtain the accurate estimation of the frequency deviation, so as to obtain the coherent demodulation result of the second section of data, and the method specifically comprises the following steps:
step 2.21, the second section of data (4) is subjected to square demodulation, and then an inner product is formed by the second section of data and the square of a local inner product template, wherein the expression of the local inner product template is as shown in a formula (5):
Figure FDA0002422322010000041
the specific method for performing inner product with the square of the local inner product template is to carry out inner product on the delta omega2Divided into P small sections at equal intervals and using square of P complex sine waves with different digital frequencies
Figure FDA0002422322010000042
And x2[n]Inner product, and taking the maximum module value of the inner product as the complex sine wave with digital frequency of-2 delta omega2Based on an estimate of
Figure FDA0002422322010000043
For compensating for frequency offset; phase position
Figure FDA0002422322010000044
Estimated according to the following equation (6):
Figure FDA0002422322010000045
the divide by 2 operation in equation (6) above results in phase ambiguity, i.e.
Figure FDA0002422322010000046
And
Figure FDA0002422322010000047
all satisfy the requirement of formula (6) because
Figure FDA0002422322010000048
The initial value is unknown, here
Figure FDA0002422322010000049
Get
Figure FDA00024223220100000410
Or
Figure FDA00024223220100000411
All can be used; when the data is processed to the frame head section, the absolute phase is determined by utilizing the polarity of the correlation result of the frame head, and the absolute phase is not taken in advance
Figure FDA00024223220100000412
The second stage data coherent demodulation result is obtained as the following formula (7):
Figure FDA00024223220100000413
5. the open-loop demodulation method for short frame burst communication with low snr according to claim 4, characterized in that: the third step is specifically as follows:
the data in this section are:
Figure FDA00024223220100000414
when the step three is operated for the first time, the data of the section refers to a third section of data, and the rest is done in sequence;
step 3.1: operating a fine search module to obtain accurate estimation of frequency offset; the fine search module adopts an exhaustion method to obtain accurate estimation of frequency deviation, and then obtains a coherent demodulation result of the data, specifically:
carrying out square demodulation on the data of the section, and then carrying out inner product on the data and the square of a local inner product template, wherein the expression of the local inner product template is as shown in a formula (5); the specific method for performing inner product with the square of the local inner product template is to carry out inner product on the delta omega2Divided into P small sections at equal intervals and using square of P complex sine waves with different digital frequencies
Figure FDA00024223220100000415
And x2[n]Inner product, and taking the maximum module value of the inner product as the complex sine wave with digital frequency of-2 delta omega3Based on an estimate of
Figure FDA0002422322010000051
For compensating for frequency offset; phase position
Figure FDA0002422322010000052
Estimated according to the following equation (9):
Figure FDA0002422322010000053
the divide by 2 operation in equation (9) above results in phase ambiguity, i.e.
Figure FDA0002422322010000054
And
Figure FDA0002422322010000055
satisfies the requirement of formula (9), namely: deriving a frequency offset estimate
Figure FDA0002422322010000056
Obtaining phase simultaneously
Figure FDA0002422322010000057
And
Figure FDA0002422322010000058
step 3.2: the phase output in the step 3.1 is deblurred, and a reference phase for deblurring is constructed according to the result of the previous section of frequency offset and the initial phase, because the last phase of the previous section is the initial phase of the current section;
if the second segment of initial phase estimation results in
Figure FDA0002422322010000059
The frequency offset estimation result is
Figure FDA00024223220100000510
The end phase should be (9):
Figure FDA00024223220100000511
this is because the last phase and the initial phase are separated by N-1 symbols; if the despreading process of the spread spectrum system is considered, the initial phase of the third segment and the final phase of the second segment are slightly different, and the following should be done (10):
Figure FDA00024223220100000512
because the second section has phase ambiguity, the third section and the following sections only need to keep a section of continuous phase strictly following the phase of the previous section; the specific treatment process is as follows (11):
Figure FDA00024223220100000513
judgment of
Figure FDA00024223220100000514
And
Figure FDA00024223220100000515
which is closer to
Figure FDA00024223220100000516
If:
Figure FDA00024223220100000517
then get
Figure FDA00024223220100000518
As an estimation result of the initial phase; otherwise, get
Figure FDA00024223220100000519
Then coherently demodulating according to the data of the third segment and according to
Figure FDA00024223220100000520
And compensating the frequency offset, wherein the output of the third stage of coherent demodulation is as follows:
Figure FDA00024223220100000521
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