CN105162333B - A kind of DAB BDC modulator approaches based on high-frequency ac buck principle - Google Patents
A kind of DAB BDC modulator approaches based on high-frequency ac buck principle Download PDFInfo
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Abstract
The invention discloses a kind of DAB BDC modulator approaches based on high-frequency ac buck principle, energy snubber inductive current is designed in electric current critical continuous conduction mode (BCM) or discontinuous mode (DCM), and controlled using constant frequency, ensure that the switching device in BDC realizes ZVT (ZVS) and Zero Current Switch (ZCS).It is proposed it is a kind of realize the algorithm of device optimal current stress multivariable solution, and current sensor is eliminated in control, directly enhances the dynamic property of system.Compared with prior art, institute's extracting method of the present invention has premium properties.
Description
Technical Field
The invention relates to a DAB-BDC modulation method based on a high-frequency alternating current buck-boost principle, which realizes an algorithm of an optimal current stress multivariable solution of a switching device, omits a current sensor in control, directly improves the dynamic performance of a system, and belongs to the field of control of power electronic converters.
Background
Bidirectional DC/DC converter (BDC) is widely used in aerospace, new energy power generation, ac and DC micro-grid, and has been increasingly researched due to its Bidirectional power flow capability to save system cost, volume and weight.
The BDC is divided into a non-isolated type and an isolated type, and because the fluctuation ranges of the input voltage and the output voltage of the BDC are large and the voltage levels of the BDC and the BDC are generally different greatly, the non-isolated BDC is difficult to realize high efficiency by using single-stage power conversion. And the isolation type BDC can utilize a high-frequency transformer to match the input and output voltage grades, so that the conversion efficiency is higher. Generally, the isolation type BDC can be divided into a voltage type BDC and a current type BDC, a switch tube in the current type BDC is easy to cause a voltage spike, a clamping circuit must be added to realize voltage spike suppression, and the complexity of the circuit is increased; and the voltage type BDC direct current side is provided with a capacitor clamp, so that the voltage stress is small, and the application is wide. The classic isolation type direct current converter can obtain a voltage type BDC after circuit transformation, a general circuit is of a symmetrical structure and comprises a flyback BDC, a forward and flyback mixed BDC, a half-bridge bidirectional converter, a full-bridge bidirectional converter and the like, wherein a Double Active Bridge (DAB) BDC can be widely applied because Zero Voltage Switching (ZVS) can be conveniently realized, and power bidirectional flow can be realized through phase-shift control. The phase and the pulse width of the alternating-current square wave voltage on the primary side and the secondary side of the transformer can be adjusted, and more research results are obtained in the aspect of reducing feedback power, current peak values of devices and total loss in recent years.
In the prior art, the concept of expanding the phase shift is more advanced, which aims to reduce the reflux power and the loss generated by the reflux power and obtain better effect, but the BDC has three independent control variables, namely the duty ratio d1 of a primary side bridge, the duty ratio d2 of a secondary side bridge and the phase shift angle phi between the primary side and the secondary side, and an optimal method for realizing the three variables is not provided, and the duty ratio of one bridge is fixed firstly when the three variables are realized, so the optimal control of the reflux power is not realized; some scholars propose a double phase-shifting control strategy with the aim of realizing minimum peak current of a switching tube, but the duty ratios of the original secondary side full bridge are preset to be equal, so that the optimization can be still further realized; some scholars propose to realize staged frequency conversion control according to the processing power of the converter so as to ensure more optimal efficiency, on one hand, staged control is difficult to realize, and on the other hand, triangular control of current is adopted so that the current stress of a device is larger; in the prior art, various complete loss models of BDC are established, the sizes of d1, d2 and phi are determined according to the design principle of minimum loss, and the optimal control scheme is determined at present, but the operation effect depends on the accuracy of the loss models, and the establishment of the loss models is very complex. In addition, ZVS acquisition in DAB-BDC is load-size dependent, and soft switching is difficult to achieve at light loads.
The traditional DAB-BDC control strategy comprises an output voltage outer ring and a current inner ring, wherein a low-pass filter is usually added to obtain a smoother average value for closed-loop feedback because the feedback quantity of the current inner ring is an inductive current, but the dynamic characteristic of a system can be influenced by the low-pass filter. Therefore, the inductor current prediction control is widely researched, but the method has strong dependence on the accuracy of the current measurement time, and the control effect can be influenced by slight error. Reference to the literature
[1]A.Rodr′1guez，A.V′azquez，D.G.Lamar,et al.Different Purpose Design Strategies and Techniques to Improve the Performance of a Dual Active Bridge With Phase-Shift Control[J].IEEE Trans.on Power Electronics，2015，30(2)：790–804.
[2]B.Zhao，Q.Song，W.Liu.Efficiency Characterization and Optimization of Isolated Bidirectional DC–DC Converter Based on Dual-Phase-Shift Control for DC Distribution Application[J].IEEE Trans.on Pow.Electron，2013，28(4)：1711–1727.
Disclosure of Invention
The purpose of the invention is as follows: aiming at the defects of the existing method, a DAB-BDC modulation strategy based on a high-frequency alternating current buck-boost principle is provided, the energy buffer inductor current is designed in a current critical continuous mode (BCM) or a discontinuous mode (DCM), and constant frequency control is adopted to ensure that a switching device in the BDC realizes ZVS and ZCS. An algorithm for realizing the optimal current stress multivariable solution of the device is provided, a current sensor is omitted in control, and the dynamic performance of the system is directly improved.
The technical scheme is as follows: a DAB-BDC modulation method based on a high-frequency alternating current buck-boost principle is characterized in that the input and output voltages in the DAB-BDC are U in 、U o The voltage on both sides of the buffer inductor is
In the formula, A, B are states of potentials at two sides of an inductor respectively, and n is a transformer transformation ratio;
the control strategy of the DAB-BDC is as follows:
the high-frequency positive and negative cycles are symmetrical, each half cycle comprises 4 modes, and the modulation ratios of the 4 modes are d 1 、d 2 、d 3 、d 4 (ii) a In the positive half cycle, the values of a in 4 modes are 1, 0, and 0,B are 0, 1, and 0, respectively; in the negative half cycle, the values of A under 4 modes are-1, 0 and 0,B are 0, -1 and 0 respectively;
depending on the input voltage, output voltage and output power, all 4 modes mentioned above may not be included in a half switching cycle.
Let T s Is the switching period, y 1 、y 2 Buffering the inductor current i for the first mode end time and the second mode end time, respectively L A value of (d); determining the time and voltage acting on the buffer inductor
Wherein L is inductance of the buffer inductor, and is obtained by equalizing rise and fall of inductor current
Primary side current i of transformer L Only the current in the second mode and the third mode can flow to the load side, and the average value of the two mode currents after being converted by the transformer is equal to the load current to obtain
Substituting (4) into (5) to obtain
Get through solution
d 3 ＝x 5 d 1 +x 6 d 2 (8)
Wherein
x 1 ＝U in (nU in -U o )；
The inductor current i is only in the second mode and the third mode L Can flow to the load side. Therefore, if d 2 +d 3 If the value of (b) is larger, under the condition of the same load current, the current is smoother, and the effective value of the current born by the switching device is smaller; let y = d 2 +d 3 Then, d when y is maximized is obtained 1 、d 2 、d 3 So that the maximum value of y is required below; substituting (7) and (8) into the expression of y to obtain
D is obtained from (9) 1 Derivative of (A) to
Let formula (10) equal to 0 to obtainWhen y obtains an extreme value in a section of interval, d 1 Corresponding value of d 1y
According to nU in And U o There are three cases of the relationship of (A) and (B), and the size of the solution
I.nU in ≥U o
At this time, x is removed 2 <, 0 and x 1 、x 3 ～x 6 Are all larger than zero, so the amount inside the root number in the formula (7) is automatically larger than zero, and d is ensured 2 Greater than zero, d 1 Must satisfy
D is to be 1x1 、d 1y 、d 1z1 Formula (9) is substituted, and the amount corresponding to the maximum value is taken as d 1 Then d is obtained according to (7) and (8) respectively 2 、d 3 The value of (c).
II.nU in ＜U o
At this time, x is removed 1 、x 2 、x 4 、x 6 Are all less than 0,x 3 、x 5 Greater than zero, so except that d is guaranteed 2 When the root number is larger than zero, the amount of the root number in the formula (7) is also ensured to be larger than zero, and the condition that the root number is larger than zero must be met
D is to be 1x2 、d 1y 、d 1z2 Substituting formula (9) with the maximum value of d 1 Then d is obtained according to (7) and (8) respectively 2 、d 3 The value of (c).
It is also possible that the current generation data is calculated to obtain d 1 +d 2 +d 3 > 1, it is said that the switching frequency needs to be changed, and d must be changed to realize constant frequency 1 +d 2 +d 3 Is limited to 1 and this limitation is taken as the solution d 1 The conditions of (a);
III.d 1 +d 2 +d 3 ＝1
at this time
d 3 ＝1-d 1 -d 2 (14)
In this case, equation (4) is still satisfied, and (14) is substituted for (4) to obtain
To ensure d 2 >, 0, then
In this case, the relationship of the formula (6) is satisfied, and (15) is substituted into (6) to obtain
According to the constraint condition (16), obtaining
Wherein
Then, d is obtained from (14) and (15) 2 And d 3 ；
According to the three calculation conditions, d obtained under one condition is finally selected 1 、d 2 And d 3 As the modulation ratio of the final first 3 modes, if d 1 +d 2 +d 3 If 1, then d 4 =0; if d is 1 +d 2 +d 3 <, 1, then d 4 ＝1-(d 1 +d 2 +d 3 ) And d is combined 1 、d 2 、d 3 And d 4 As a source signal, a driving signal of a switching tube is directly realized through a modulation strategy.
Drawings
FIG. 1 is a diagram of a DAB-BDC circuit configuration;
FIG. 2 shows a high frequency AC step-up and step-down modulation strategy, in which (a) is a high frequency AC step-down control mode, (b) is a high frequency AC step-up and step-down control mode, and (c) is a high frequency AC step-up control mode;
FIG. 3 illustrates a unified high frequency AC buck-boost approach;
FIG. 4 is a flow chart of a multivariate solution;
FIG. 5 is an AC buck-boost control strategy for DAB-BDC;
FIG. 6 is a graph showing the variation of the effective value of current with the input voltage;
FIG. 7 is a DPS control strategy;
FIG. 8 is a graph showing the variation of the effective value of the current controlled by the DPS with the input voltage;
figure 9 is a comparison of current stress for DPS control versus the methods presented herein.
Detailed Description
The present invention is further illustrated by the following examples, which are intended to be purely exemplary and are not intended to limit the scope of the invention, as various equivalent modifications of the invention will occur to those skilled in the art upon reading the present disclosure and fall within the scope of the appended claims.
As shown in FIG. 1, the DAB-BDC topology includes a low-side full bridge composed of switching tubes S1-S4, a high-side full bridge composed of switching tubes S5-S8, and a bridge C r1 -C r8 L is an energy buffer inductor corresponding to the equivalent output capacitance of the switch tube, and comprises the sum of the series inductor and the leakage inductor of the transformer, U in And U o Voltage values of low and high voltage, C 1 And C 2 Filter capacitors, i, on the low and high voltage side, respectively L 、i S Are primary and secondary side currents u of the transformer respectively L1 、u S The voltage u is the AC side voltage of the low-voltage side full-bridge circuit and the high-voltage side full-bridge circuit respectively L2 Is a voltage u S The value converted to the primary side, T is a high-frequency transformer, n is the transformer transformation ratio, R L To be a load, I o Is the load current.
In a conventional dual active bridge, the voltage u L1 、u S High-frequency AC voltage with positive and negative width of 50%, and control u L1 、u S The power flow can be realized by the phase shift angle between the U phase and the U phase, but the control method is only carried out on the U phase in ＝U o Higher efficiency can be achieved near/n because the converter under control of this method is in U in And U o When the difference/n is larger, larger feedback power exists. To overcome this problem, the current in the inductor L is 0 if the energy stored in the inductor can be fully discharged during the positive or negative half cycle of the switch, i.e. before the start of the positive and negative half cycles.
Input and output voltage U in DAB-BDC in 、U o The transformation ratio n of the transformer is generally designed to match the input and output voltages in order to improve efficiency, and therefore the voltage u is generally constant L1 And u L2 The amplitude relationship of (2) is not fixed, and in order to ensure that the current of the inductor L is 0 before the end of a half switching period, a control strategy of a classical buck, buck/boost and boost circuit can be used when the current is interrupted. Suppose that the input and output voltages of buck, buck/boost and boost circuits are all U in And U o N, the voltage across the corresponding inductor is
u L ＝AU in -BU o /n (1)
Wherein A, B is the state of the voltage at both sides of the inductor, and the values are shown in table 1, and there are three modes in a switching period, corresponding to the current respectivelyRise, fall and remain at zero. The method is extended to the high-frequency alternating current in DAB-BDC, and the voltage u in the figure 1 is corresponded L1 、u L2 And a current i L The waveforms of (a) are shown in fig. 2, and correspond to high-frequency ac step-down, step-up and step-down, and step-up, respectively. In the figure D bu 、D bb 、D bo The modulation ratios are respectively for voltage reduction, voltage increase and voltage decrease, and voltage increase.
TABLE 1 DC CONVERTER INDUCTOR INPUT/OUTPUT VOLTAGE STATE
Presetting a voltage threshold according to the voltage u L1 And u L2 The amplitude relationship determination circuit of (a) operates in a control mode of the type shown in fig. 2, and how to implement ZVS and ZCS of the device, but this control strategy also has significant disadvantages:
1) Relative to the case of the current continuous state, current i L The peak value of (a) is large, so that the current stress borne by the device is large;
2) Discontinuities in control tend to cause system instability at handover.
In order to keep ZVS and ZCS characteristics of the device and inhibit current stress of the device, on one hand, the inductor is required to be ensured to work in DCM or BCM, and on the other hand, a control strategy corresponding to a unified control high-frequency alternating current buck-boost principle is required to be developed. The voltage across the inductor in DAB-BDC still satisfies equation (1), and the present invention proposes the control strategy shown in table 2.
TABLE 2 unified high frequency AC Buck-boost control strategy
The high frequency positive and negative cycles shown in FIG. 2 are symmetrical, each half cycle comprises 4 modes, and the modulation ratio is d 1 、d 2 、d 3 、d 4 . In the positive half cycle, the first two modes and tables in Table 2Boost in 1 is the same and the modulation ratio is d 2 、d 3 The two modes of (a) are the same as buck in table 1, the two states share a modulation ratio d 2 The mode of (2), that is, the control strategy mentioned in table 2 is essentially a hybrid control mode of buck and boost, and as long as the time ratio occupied by them is changed, the voltage increase or decrease can be easily realized, and the control parameters are continuously changed without jump. At the end of the 3 rd mode, the current i L Becomes zero and preserves the device ZCS characteristics. Depending on the input, output voltage and output power, one or both of the modes shown in table 2 may not be included in a half switching cycle.
According to the control strategy shown in Table 2, the voltage u on both sides of the inductor in one switching period L1 、u L2 And the inductor current i L Is shown in figure 3. In the figure, d 1 、d 2 、d 3 、d 4 Respectively corresponding to the modulation ratios, T, of the 4 modes in the half period in Table 2 s Is the switching period, y 1 、y 2 Current i at the end of the first mode and the end of the second mode, respectively L The value of (c).
Determining from the time and voltage acting on the inductor
In FIG. 3, the increase and decrease of the inductor current are equal to each other to obtain
Current i in fig. 3 L Only the currents of the second mode and the third mode can flow to the load side, i.e. the shaded part in the figure, so that the average of the currents of the two modes after being converted by the transformerA value equal to the load current to
Substituting (4) into (5) to obtain
Get it solved
d 3 ＝x 5 d 1 +x 6 d 2 (8)
Wherein
x 1 ＝U in (nU in -U o )
As can be seen from fig. 3, the inductor current i is only in the second mode and the third mode L Can flow to the load side. Therefore, if d 2 +d 3 If the value of (b) is larger, the current is smoother under the condition of the same load current, and the effective value of the current borne by the switching device is smaller. Let y = d 2 +d 3 Then d when y is maximized is found 1 、d 2 、d 3 So that the maximum value of y is required below. Substituting (7) and (8) to obtain
D is obtained from (9) 1 Derivative of (A) to
Let formula (10) equal to 0 to obtain d corresponding to extreme value in a section of interval 1 Has a value of d 1y
According to nU in And U o There are three cases of the relationship of (A) and (B), and the size of the solution
I.nU in ≥U o
At this time, x is removed 2 < 0 and x 1 、x 3 ～x 6 Are all larger than zero, so the amount inside the root number in the formula (7) is automatically larger than zero, and d is ensured 2 Greater than zero, d 1 Must satisfy
Will d 1x1 、d 1y 、d 1z1 Substituting formula (9) with the maximum value of d 1 Then d is obtained according to (7) and (8) 2 、d 3 The value of (c).
II.nU in ＜U o
At this time, x is removed 1 、x 2 、x 4 、x 6 Are all less than 0,x 3 、x 5 Greater than zero, thus except that d is guaranteed 2 When the root number is larger than zero, the amount of the root number in the formula (7) is also ensured to be larger than zero, and the condition that the root number is larger than zero must be met
Will d 1x2 、d 1y 、d 1z2 Substituting formula (9) with the maximum value of y as d 1 Then d is obtained according to (7) and (8) respectively 2 、d 3 The value of (c).
It is also possible that the current generation data is calculated to obtain d 1 +d 2 +d 3 > 1, it is said that the switching frequency needs to be changed, and d must be changed to realize constant frequency 1 +d 2 +d 3 Is limited to 1 and this limitation is taken as the solution d 1 The conditions of (a);
III.d 1 +d 2 +d 3 ＝1
at this time
d 3 ＝1-d 1 -d 2 (14)
In this case, formula (4) is still satisfied, and (4) is substituted into (14) to obtain
To ensure d 2 >, 0, then
In this case, the relationship of the formula (6) is satisfied, and substitution of (15) into (6) results in
According to the constraint conditions of (16), obtaining
Wherein
Then, d is obtained from (14) and (15) 2 And d 3 ；
According to the three calculation conditions, d obtained under one condition is finally selected 1 、d 2 And d 3 As the modulation ratio of the final first 3 modes, if d 1 +d 2 +d 3 If 1, then d 4 =0; if d is 1 +d 2 +d 3 &1, then d 4 ＝1-(d 1 +d 2 +d 3 ) And d is 1 、d 2 、d 3 And d 4 As a source signal, a driving signal of a switching tube is directly realized through a modulation strategy.
The above solution process is clearly illustrated by the flow chart shown in fig. 4. Note that the calculations in the flow chart replace the load current with the reference current I. The flow shown in fig. 4 is clear and can be easily implemented by a DSP.
In obtaining d 1 、d 2 、d 3 And d 4 And the driving signal of the switching tube is directly realized through a modulation strategy. The control block diagram in the boost mode adopted by the present invention is shown in fig. 5. When operating in buck mode, devices such as batteries are typically charged using the same control scheme except that the reference for the current loop is a constant during constant current charging, and in fig. 5, the reference current I is shown as o * Is the amplitude. Compared with the traditional control method, the method provided by the invention saves current sensors, the current loop does not need current feedback quantity, the modulation ratio is directly obtained through the established mathematical model, and the more accurate feedback quantity does not need to be established through a sensor and a current prediction method, so that the dynamic performance of the system is greatly improved.
Parameter design method
In the DAB-BDC shown in fig. 1, there are two key parameters to be designed, one is the transformer transformation ratio n and the other is the snubber inductance L. Because the current stress of the device is minimum when the input voltage and the converted output voltage on the primary side are equal in the DAB-BDC, the fluctuation ranges of the input voltage and the output voltage can be comprehensively considered, and the input voltage and the converted output voltage on the primary side are equal when the voltage fluctuates by a middle value.
A typical application of DAB-BDC is to charge and discharge an energy storage battery in a direct-current microgrid, and typical input and output voltage parameters are as follows: u shape in In the range of 42-56V, U o The range is 360-400V, and the maximum output power is 500W according to the design of the embodiment. The design of the inductance L is explained with the aid of this parameter.
The intermediate voltages of the input and output voltages are 49V and 380V, respectively, and therefore, the transformer transformation ratio n =380/49=7.75 is determined first. There are no multiple methods for designing the inductor L, and document [1] proposes two design methods, one of which is designed from the viewpoint of reducing reactive power, and the method requires that the inductance value should be as small as possible; the other design is based on increasing the range of ZVS, the inductance value is required to be as large as possible under the condition of ensuring rated output power, and the two methods have advantages and disadvantages respectively. The inductance value is designed in the present embodiment based on the minimum inductor current stress (linear relationship with the switching tube current).
Y is represented by the formulae (2) and (3) 1 、y 2 In FIG. 2, the function of the current in 3 stages in a half cycle is written, and the inductance current i is obtained according to the listed function L Effective value of (I) L Is composed of
Calculation of I L The three-segment modulation ratio d must be known 1 、d 2 、d 3 But calculating d 1 、d 2 、d 3 The process of (a) cannot be solved by using a complete and continuous expression, and on the premise of not affecting the calculation precision, the embodiment adopts matlab/simulink to construct a simulation calculation model shown in a flow chart, so as to obtain d under the conditions of different input voltages and different inductance values 1 、d 2 、d 3 As shown in table 3, it can be seen from table 3 that: not all situations experience the 4 modes shown in fig. 3, in case of low input voltage, light load, only the first 3 modes,corresponding to the ac boost condition shown in fig. 2 (c); in the case of high input voltage and light load, only the last 3 modes exist, which corresponds to the ac step-down case shown in fig. 2 (a). In most cases, however, d 1 +d 2 +d 3 =1, which helps to reduce the current stress of the switching tube. Corresponding to the modulation ratio data and equation (19) in Table 3, I is obtained under different power and different inductance conditions L The curve with the input voltage is shown in fig. 6. It can be seen that a large inductance (e.g., 10 muH inductance in the figure) will cause I to be present at light load over the entire input voltage range L Is small; however, as the power increases, the inductance current effective value I generated by the larger inductance L Faster increase and much larger than for other inductance values; at the same power, an inductance that is too small (e.g., 2.5 muH inductance in the figure) varies strongly across the input voltage range, and I L The value of (c) is also very large. Therefore, the inductance is selected moderately (such as 5 muH and 7.5 muH in the figure), on one hand, the current stress of the device does not change too much in the whole variation range of the input voltage under the same power; on the other hand, in the range from light load to rated load, the current stress of the device is in a relatively optimized state relative to the maximum inductance value and the minimum inductance value. Taking the above situations into consideration, L =6 μ H is selected.
TABLE 3 unified high frequency AC Buck-boost control strategy
It is worth noting that: in the design of the document [1], the range of the device ZVS and the suppression of reactive power are mutually contradictory, and the larger the range of the larger device ZVS, the better the inductance value is required to be; however, in order to suppress the reactive power, it is preferable that the inductance value is smaller, which is a contradiction. In the method, the current stress optimization can be realized by selecting a proper inductance value on one hand, and on the other hand, the wide range of the ZVS of the device can be ensured by selecting a proper inductance value.
Comparison with Current stress of Dual phase Shift control strategy
Among the numerous DAB-BDC improved control strategies, document [2]]The proposed double-Phase-Shift (DPS) control strategy is representative, and the proposed control scheme is shown in fig. 7, in which the modulation ratios of the front and rear full-bridge circuits of the transformer are first set to be equal, i.e. D in the figure 1 The implementation is carried out; on the basis, the high-voltage side and the low-voltage side full bridges are shifted, namely D in the figure 2 And (5) realizing.
The parameters of the converter are the same as described above, using reference [2]]The method for solving the control quantity is provided to obtain a group (D) 1 ，D 2 ) Data, as shown in table 4, according to the data, a curve of the effective value of the inductor current along with the change of the input voltage when the inductor is different is obtained, as shown in fig. 8, the current is different in size when the inductor is different from the light load to the full load, and the inductor is larger when the inductor is light load, so that the smaller current stress can be realized; at full load, the opposite is true.
Considering a moderate inductance value, L =10 μ H is finally taken.
TABLE 4 DPS control strategy
A. Comparison of current stresses
In order to directly compare the method of the present invention with the method of document [2], fig. 9 shows the current stress of two control methods at different powers. It can be seen that the frequency-increasing AC buck-boost circuit has obvious advantages when the circuit is fully loaded; when the load is light, the current stress of the method and the DPS control method is in a staggered state. Overall, the method of the present invention is slightly more advantageous in terms of current stress.
B. Mining of control object potential
In the DPS control shown in fig. 7, the ac-side voltage widths of the high-side and low-side full-bridge circuits are designed to be equal, and therefore, if the pulse width modulation of the high-side full-bridge is increased, the current stress of the switching devices in the converter will be further optimized, but the complexity of the mathematical model for calculating the phase shift time will be multiplied, and it is necessary to consider the increaseMore cases are considered, and it is nearly impossible to calculate three phase shift times. In the high-frequency ac buck-boost control method provided by the present invention, there are 4 control objects, which are d shown in fig. 2 1 、d 2 、d 3 、d 4 Finding out the most (d) among them according to the constraint variables 1 ，d 2 ，d 3 ，d 4 ) And (4) combining. It should be said that in the aspect of mining the potential of the control object, the control strategy based on the high-frequency alternating current buck-boost provided by the invention is due to the DPS control strategy.
C. Soft switching capability comparison
Under the condition of large load to a certain extent, all devices in DAB-BDC controlled by DPS can realize ZVS, but under the condition of light load, the energy stored in the inductor is not enough to draw out the charges stored in the two switch junction capacitors, so that under the condition of light load, the DPS method can not realize the soft switching of the devices [1] . The method provided by the invention designs the inductive current in a DCM or BCM state, so that at the beginning of a half cycle and the moment when the inductive current becomes zero, the device can be ensured to realize ZCS switching no matter in a light load state or a full load state. This feature can effectively improve the efficiency of the converter under light load.
Claims (1)
1. A DAB-BDC modulation method based on a high-frequency alternating-current buck-boost principle is a topology structure of DAB-BDC and comprises a low-voltage side full bridge formed by switching tubes S1-S4, a high-voltage side full bridge formed by switching tubes S5-S8 and a high-voltage side full bridge C r1 -C r8 L is an energy buffer inductor corresponding to the equivalent output capacitance of the switch tube, and comprises the sum of the series inductor and the leakage inductor of the transformer, U in And U o Voltage values of low and high voltage, C 1 And C 2 Filter capacitors, i, on the low and high voltage side, respectively L 、i S Are primary and secondary side currents u of the transformer respectively L1 、u S The AC side voltage u of the full bridge circuit at the low voltage side and the high voltage side respectively L2 Is a voltage u S Converted to the value of the primary side, T is the high frequency transformer, n is the transformer transformation ratio, R L To be a load, I o Is the load current; the input and output voltages in DAB-BDC are U in 、U o The voltage on both sides of the buffer inductor is
In the formula, A, B is the state of the electric potential at two sides of the inductor respectively, and n is the transformer transformation ratio;
the DAB-BDC control strategy is characterized by comprising the following steps:
the high-frequency positive and negative cycles are symmetrical, each half cycle comprises 4 modes, and the modulation ratios of the 4 modes are d 1 、d 2 、d 3 、d 4 (ii) a In the positive half cycle, the values of a in 4 modes are 1, 0, and 0,B are 0, 1, and 0, respectively; in the negative half cycle, the values of A under 4 modes are-1, 0 and 0,B are 0, -1 and 0 respectively;
depending on the input voltage, output voltage and output power, all 4 modes mentioned above may not be included in a half switching cycle.
Let T s Is the switching period, y 1 、y 2 Buffering the inductor current i for the first mode end time and the second mode end time, respectively L A value of (d); determining the time and voltage acting on the buffer inductor
Wherein L is inductance of the buffer inductor, and is obtained by equalizing rise and fall of inductor current
Primary side current i of transformer L Having only a second modeThe current in the state and the third mode time period can flow to the load side, and the average value of the current in the two modes after being converted by the transformer is equal to the load current to obtain
Substituting (4) into (5) to obtain
Get it solved
d 3 ＝x 5 d 1 +x 6 d 2 (8)
Wherein
Wherein, I o Is the load current;
the inductor current i is only in the second mode and the third mode L Can flow to the load side; therefore, if d 2 +d 3 If the value of the voltage is larger, the current is smoother under the condition of the same load current, and the effective value of the current born by the switching device is smaller; let y = d 2 +d 3 Then, d when y is maximized is obtained 1 、d 2 、d 3 So the maximum value of y is required below; substituting (7) and (8) into the expression of y to obtain
D is obtained from (9) 1 Derivative of (A) to
Let equation (10) equal to 0, and obtain the value of y in a segment of interval, d 1 Corresponding value of d 1y
According to nU in And U o There are three cases of the relationship of (c), and the size of the solution
I.nU in ≥U o
At this time, x is removed 2 < 0 and x 1 、x 3 ～x 6 Are all larger than zero, so the amount inside the root number in the formula (7) is automatically larger than zero, and d is ensured 2 Greater than zero, d 1 Must satisfy
Will d 1x1 、d 1y 、d 1z1 Formula (9) is substituted, and the amount corresponding to the maximum value is taken as d 1 Then d is obtained according to (7) and (8) respectively 2 、d 3 The value of (c).
II.nU in ＜U o
At this time, x is removed 1 、x 2 、x 4 、x 6 Are all less than 0,x 3 、x 5 Greater than zero, thus except that d is guaranteed 2 When the root number is larger than zero, the amount of the root number in the formula (7) is also ensured to be larger than zero, and the condition that the root number is larger than zero must be met
D is to be 1x2 、d 1y 、d 1z2 Formula (9) is substituted, and the amount corresponding to the maximum value is taken as d 1 Then d is obtained according to (7) and (8) 2 、d 3 The value of (c).
It is also possible that the current generation data is calculated to obtain d 1 +d 2 +d 3 &1, it is shown that the switching frequency needs to be changed, and d must be changed to realize constant frequency 1 +d 2 +d 3 Is limited to 1 and this limitation is taken as the solution d 1 The conditions of (a);
III.d 1 +d 2 +d 3 ＝1
at this time
d 3 ＝1-d 1 -d 2 (14)
In this case, equation (4) is still satisfied, and (14) is substituted for (4) to obtain
To ensure d 2 >, 0, then
In this case, the relationship of the formula (6) is satisfied, and substitution of (15) into (6) results in
According to the constraint conditions of (16), obtaining
Wherein
Then, d is obtained from (14) and (15) 2 And d 3 ；
According to the three calculation conditions, d obtained under one condition is finally selected 1 、d 2 And d 3 As the modulation ratio of the final first 3 modes, if d 1 +d 2 +d 3 If 1, then d 4 =0; if d is 1 +d 2 +d 3 &1, then d 4 ＝1-(d 1 +d 2 +d 3 ) And d is combined 1 、d 2 、d 3 And d 4 As a source signal, a driving signal of a switching tube is directly realized through a modulation strategy.
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