CN105048501A - LCL-type inverter decoupling control method based on state feedback - Google Patents

Abstract

The invention relates to an LCL-type inverter decoupling control method based on state feedback. The method comprises the following steps of: S1, setting an active component and a reactive component of a net current given value i2, acquiring a monitoring parameter in real time, and obtaining the active component and the reactive component corresponding to the monitoring parameter, wherein the monitoring parameter comprises the net current given value i2, actual voltage values u<c> of the two ends of a capacitor, an actual current value i1 passing through a first inductor and a network voltage e; S2, respectively acquiring an active component and a reactive component of a grid-connected inverter output voltage given value through the state feedback and closed-loop control of the monitoring parameter according to the active component and the reactive component of the net work current given value I2; and S3, carrying out pulse width modulation (PWM) control according to the active component and the reactive component of the grid-connected inverter output voltage given value, and controlling a grid-connected inverter through an output switch-controlled PWM wave. Compared with the prior art, the LCL-type inverter decoupling control method based on state feedback has the advantages that independent control of the active component and the reactive component is achieved, and high output current quality is also guaranteed.

Description

A kind of LCL type inverter decoupling control method based on state feedback

Technical field

The present invention relates to field of inserter control, especially relate to a kind of LCL type inverter decoupling control method based on state feedback.

Background technology

Developing rapidly of generation of electricity by new energy technology, makes parallel network reverse technology become study hotspot.As the key component of grid-connected inverting system, combining inverter adopts pulse modulation technology (PWM) usually, containing high order harmonic component in its output current, cannot meet grid-connected requirement.Therefore, it exports and needs to access filter between electrical network, conventional L-type and LCL type two type filter.LCL type filter has more preferably High frequency filter effect compared with L-type filter, and volume and loss are all less than the L-type filter of equal filter effect.How to attract wide attention by improving control strategy raising LCL type combining inverter performance.The connection diagram of LCL type filter and combining inverter as shown in figure 11, LCL type filter comprises the first inductance L 1, electric capacity CJ and the second inductance L 2, one end of first inductance L 1 connects the output of combining inverter, the other end of the first inductance L 1 connects the positive pole of electric capacity CJ and one end of the second inductance L 2 respectively, the minus earth of electric capacity CJ, the other end of the second inductance L 2 connects electricity consumption end.

According to the difference of reference frame, the controller of combining inverter can be divided into based on static three-phase natural system of coordinates with based on rotation two-phase synchronous coordinate system two class.Under static three phase coordinates, ratio resonant controller (PR) can directly be followed the tracks of AC signal, avoids the coupled problem caused by coordinate transform, but PR existence is difficult to realize the problems such as Digital Discrete.Therefore, the control control strategy based on synchronous coordinate system is selected in most of application scenario.Under synchronous coordinate system, three-phase ac signal becomes two-phase direct current signal through abc/dq coordinate transform.But there is coupling between the dq component that conversion obtains, this problem becomes particularly outstanding in LCL filter situation, has a strong impact on the dynamic property of control system.For coupled problem, coupling terms is used as the design difficulty that scheme that external disturbance item directly ignores can reduce controller to a great extent, the independence realized between dq component controls, but, directly ignore coupling terms and must cause modeling distortion, cause output current quality to reduce.With directly ignore compared with coupling terms, by ssystem transfer function equivalent transformation, the method introducing compensating for coupling item improves model accuracy, can ensure higher output current quality.But the program is in fact carry out alternative original coupling terms by introducing suitable compensation term, to realize Accurate Model, but cannot eliminate the coupling influence between dq component.

Chinese patent CN102545264A discloses a kind of control method of the combining inverter based on feed-forward decoupling of state quantity, comprising: (1) gathers line voltage, current feedback amount and quantity of state; (2) command signal is generated according to current feedback amount; (3) feed-forward signal is tried to achieve according to quantity of state; (4) make command signal superpose with feed-forward signal and obtain modulation signal, generate the switching signal controlling combining inverter according to modulation signal.This patent is by introducing feedforward amount to realize third-order system active damping schemes, and former third-order system is degraded to first-order system by quantity of state feedforward, simplifies Controller gain variations process.But when adopting the control method under synchronous coordinate system, the coupled problem between the dq component brought by coordinate transform is not considered in that patent.

Summary of the invention

Object of the present invention be exactly in order to overcome above-mentioned prior art exist defect and a kind of LCL type inverter decoupling control method based on state feedback is provided, not only ensure that the accuracy of modeling process, guarantee higher networking current quality, and the coupling influence can effectively eliminated between dq component, independence between realization output is meritorious and idle controls, and improves dynamic performance.

Object of the present invention can be achieved through the following technical solutions:

A kind of LCL type inverter decoupling control method based on state feedback, PWM for combining inverter controls, described combining inverter connects electricity consumption end by LCL filter, described LCL filter comprises the first inductance, electric capacity and the second inductance, and this LCL type inverter decoupling control method comprises the following steps:

S1: setting networking given value of current value i ₂real component and idle component, Real-time Collection monitoring parameter, monitoring parameter comprises networking current actual value i ₂, electric capacity both end voltage actual value u _c, flow through the current actual value i of the first inductance ₁with line voltage e, and obtain real component corresponding to monitoring parameter and idle component;

S2: networking given value of current value i ₂real component and idle component obtain combining inverter output voltage set-point respectively by the state feedback of monitoring parameter and closed-loop control real component and idle component;

S3: according to combining inverter output voltage set-point real component and idle component carry out PWM control, the switch control rule PWM ripple of output controls combining inverter.

Combining inverter output voltage set-point is obtained in described step S2 the method of real component be specially:

11) networking given value of current value i ₂real component deduct networking current actual value i ₂real component after carry out outer shroud PI control obtain after state feedback electric capacity both end voltage value u _c' real component;

12) u _c' real component deduct the idle component coupling terms of the second inductance and the real component of e respectively after obtain electric capacity both end voltage set-point real component;

13) electric capacity both end voltage set-point real component deduct electric capacity both end voltage actual value u _creal component after carry out middle ring P and control to obtain the current value i of first inductance after state feedback ₁' real component;

14) i ₁' real component deduct idle component coupling terms and the networking current actual value i of electric capacity respectively ₂real component after obtain and flow through the given value of current value of the first inductance real component;

15) the given value of current value of the first inductance is flow through real component deduct the current actual value i flowing through the first inductance ₁real component after carry out inner ring P control obtain after state feedback combining inverter output voltage values u ₁' real component;

16) u ₁' real component deduct idle component coupling terms and the electric capacity both end voltage actual value u of the first inductance respectively _creal component after obtain combining inverter output voltage set-point real component;

Obtain combining inverter output voltage set-point the method of idle component be specially:

21) networking given value of current value i ₂idle component deduct networking current actual value i ₂idle component after carry out outer shroud PI control obtain after state feedback electric capacity both end voltage value u _c' idle component;

22) u _c' idle component add the real component coupling terms of the second inductance after deduct the idle component of e, obtain electric capacity both end voltage set-point idle component;

23) electric capacity both end voltage set-point idle component deduct electric capacity both end voltage actual value u _cidle component after carry out middle ring P and control to obtain the current value i of first inductance after state feedback ₁' idle component;

24) i ₁' idle component add the real component coupling terms of electric capacity after deduct networking current actual value i ₂idle component, obtain and flow through the given value of current value of the first inductance idle component;

25) the given value of current value of the first inductance is flow through idle component deduct the current actual value i flowing through the first inductance ₁idle component after carry out inner ring P control obtain after state feedback combining inverter output voltage values u ₁' idle component;

26) u ₁' idle component add the real component coupling terms of the first inductance after deduct electric capacity both end voltage actual value u _cidle component, obtain combining inverter output voltage set-point idle component.

Real component coupling terms and the idle component coupling terms of described second inductance are respectively ω L ₂i _2dwith ω L ₂i _2q, wherein, ω is line voltage angular frequency, L ₂be the inductance value of the second inductance, subscript d represents real component, and subscript q represents idle component.

Real component coupling terms and the idle component coupling terms of described electric capacity are respectively ω Cu _cdwith ω Cu _cq, wherein, ω is line voltage angular frequency, and C is the capacitance of electric capacity, and subscript d represents real component, and subscript q represents idle component.

Real component coupling terms and the idle component coupling terms of described first inductance are respectively ω L ₁i _1dwith ω L ₁i _1d, wherein, ω is line voltage angular frequency, L ₁be the inductance value of the first inductance, subscript d represents real component, and subscript q represents idle component.

The Proportional coefficient K that described outer shroud PI controls _p1, integral coefficient K _iand the Proportional coefficient K that middle ring P controls _p2with the Proportional coefficient K that inner ring P controls _p3meet following formula:

$K_{P2}=\frac{0.01K_{P3}^2-6}{2K_{P3}}---\left(1\right)$

$\sqrt{\frac{K_I{K}_{P2}{K}_{P3}}{L_1{\mathrm{CL}}_2}}=2\frac{K_I}{K_{P1}}---\left(2\right)$

$\sqrt{\frac{K_{P2}{K}_{P3}{L}_2+L_1+L_2}{L_1{\mathrm{CL}}_2}}\&CenterDot;\frac{K_{P1}}{K_I}=10---\left(3\right)$

Wherein, L ₁be the inductance value of the first inductance, C is the capacitance of electric capacity, L ₂it is the inductance value of the second inductance.

Compared with prior art, the present invention has the following advantages:

1) by removing the coupling terms that LCL filter is brought, eliminate the coupled problem between dq component under synchronous coordinate system, achieve the dynamic decoupling between dq component, effectively can eliminate the coupling influence between meritorious and idle component, realize independence that is meritorious, idle component and control;

2) by removing the coupling terms that LCL filter is brought, achieving the Accurate Model of system, making the design of control system more accurate, ensure that exporting networking electric current has the higher quality of power supply;

3) by removing the coupling terms that LCL filter is brought, the basis of the system of guarantee Accurate Model achieves the dynamic decoupling between dq component, eliminate the coupling influence between its real component and idle component when current-order changes, improve dynamic performance when current-order changes.

4) in order to ensure the stability of a system, stability margin requirement, to the controling parameters K of outer shroud, middle ring and inner ring _p1, K _i, K _p2, K _p3be optimized design, thus the stability of a system and stability margin are provided.

Accompanying drawing explanation

Fig. 1 is control block diagram of the present invention;

Fig. 2 is the control block diagram directly ignoring coupling terms scheme;

Fig. 3 is the schematic diagram of d axle equivalent transformation;

Wherein, (3a) is d shaft model schematic diagram before equivalent transformation, and (3b) is d shaft model schematic diagram after equivalent transformation;

Fig. 4 is the control block diagram compensating coupling terms scheme;

Fig. 5 is the system block diagram comprising state feedback;

Fig. 6 is L ₁model structural representation after decoupling zero;

Fig. 7 is model structural representation after electric capacity C decoupling zero;

Fig. 8 is inductance L ₂model structural representation after decoupling zero;

Fig. 9 is networking current simulations waveform schematic diagram;

Wherein, (9a) be the networking current simulations waveform schematic diagram of scheme proposed by the invention, (9b) for directly ignoring the networking current simulations waveform schematic diagram of coupling terms scheme, (9c) is the networking current simulations waveform schematic diagram compensating coupling terms scheme;

Figure 10 is networking electric current dq component schematic diagram;

Wherein, (10a) be the networking electric current dq component schematic diagram of scheme proposed by the invention, (10b) for directly ignoring the networking electric current dq component schematic diagram of coupling terms scheme, (10c) is the networking electric current dq component schematic diagram compensating coupling terms scheme;

Figure 11 is the connection diagram of LCL type filter and combining inverter.

In figure, the first inductance in L1:LCL mode filter;

Electric capacity in CJ:LCL mode filter;

The second inductance in L2:LCL mode filter;

Subscript d: real component or d axle component;

Subscript q: idle component or q axle component;

L ₁, L ₂, C:LCL filter parameter, wherein, L ₁be the inductance value of the first inductance, L ₂be the inductance value of the second inductance, C is the capacitance of electric capacity;

ω: line voltage angular frequency;

E: line voltage;

I ₁: the current actual value flowing through the first inductance;

flow through the given value of current value of the first inductance;

I ₁': the current value of first inductance after state feedback;

I ₂: networking current actual value;

networking given value of current value;

I _c: the current actual value flowing through electric capacity;

flow through the current actual value of electric capacity;

U _c: electric capacity both end voltage actual value;

electric capacity both end voltage set-point;

U ₁: combining inverter output voltage actual value;

combining inverter output voltage set-point;

U ₁': combining inverter output voltage values after state feedback;

U _c': electric capacity both end voltage value after state feedback;

K _1P: outer shroud networking current PI controller proportionality coefficient;

K _i: outer shroud networking current PI controller integral coefficient;

K _2P: middle ring capacitance voltage P controller proportionality coefficient;

K _3P: inner ring pusher side inductive current P controller proportionality coefficient;

X ₁: state variable;

the differential of state variable;

U ₁: input variable;

Y ₁: output variable;

A ₁: state matrix;

B ₁: input matrix;

C ₁: output matrix;

H ₁: state feedback matrix;

S: Laplace transform operator.

Embodiment

Below in conjunction with the drawings and specific embodiments, the present invention is described in detail.The present embodiment is implemented premised on technical solution of the present invention, give detailed execution mode and concrete operating process, but protection scope of the present invention is not limited to following embodiment.

For the deficiency of existing conventional method, the present invention proposes a kind of LCL type inverter decoupling control method based on state feedback, PWM for combining inverter controls, as shown in figure 11, combining inverter connects electricity consumption end by LCL filter, LCL filter comprises the first inductance L 1, electric capacity CJ and the second inductance L 2, one end of first inductance L 1 connects the output of combining inverter, the other end of the first inductance L 1 connects the positive pole of electric capacity CJ and one end of the second inductance L 2 respectively, the minus earth of electric capacity CJ, the other end of the second inductance L 2 connects electricity consumption end, dynamic decoupling is realized by adopting state feedback.Below principle is described:

(1) due in the expression formula of coupling terms containing inductance value or capacitance item, and general inductance capacitance value is all less, relative to for, the value of coupling terms is general very little.Thus, directly can ignore when model accuracy is less demanding, and then design the traditional double Closed-loop Control Strategy shown in Fig. 2, in order to the resonance problems suppressing LCL third-order system to cause due to underdamping, inner ring adopts P controller to increase system damping, ensures the stability of a system; Outer shroud adopts PI controller to realize the tracking to given networking current signal.The method can ensure the stability of a system, and realizes the tracking to given output networking electric current.The present invention program adopts state feedback independently to control with the decoupling zero realized between each state variable dq component, because native system state variable is two inductive currents and an electric capacity both end voltage, therefore needs tricyclic structure to realize the strategy proposed in the present invention.Compared with traditional double closed-loop control, the coupled problem that three closed loop strategy in the present invention achieve system Accurate Model and eliminate between dq component.

(2) for d axle, according to the principle of equal effects, the output current i of two block diagrams in Fig. 3 _2dwith input voltage u _1dbetween should have identical transfer function relation, have Fig. 3 (a):

(s ³L ₁CL ₂+sL ₁)i _2d＝u _1d-u _cd-s ²L ₁Ce _d-ωL ₁i _1q+s ²L ₁CωL ₂i _2q-sL ₁ωCu _cq(4)

Have Fig. 3 (b):

(s ³L ₁CL ₂+sL ₁)i _2d＝u _1d-u _cd-s ²L ₁Ce _d-dLCL(5)

Then:

dLCL＝ωL ₁i _1q-s ²L ₁CωL ₂i _2q+sL ₁ωCu _cq(6)

D axle compensating for coupling item dLCL is obtained after eliminating variable s:

dLCL＝dL ₁+dC+dL ₂＝3ωL ₁i _1q-(2ωL ₁+ω ³L ₁CL ₂)·i _2q+3ω ²L ₁C·u _cd-ω ²L ₁Ce _d(7)

Use the same method, q axle compensating for coupling item qLCL can be obtained:

qLCL＝3ωL ₁i _1d-(2ωL ₁+ω ³L ₁CL ₂)·i _2d-3ω ²L ₁C·u _cq+ω ²L ₁Ce _q(8)

According to obtained compensating for coupling item, design the control strategy including compensating for coupling item shown in Fig. 4, and adopt the two close cycles strategy identical with Fig. 2.

(3) under rest frame, have the first inductance L 1:

$\{\begin{array}{c}{u}_{1d}-u_{cd}=L_1\frac{{\mathrm{di}}_{1d}}{dt}-{\mathrm{\&omega;L}}_1{i}_{1q}\\ u_{1q}-u_{cq}=L_1\frac{{\mathrm{di}}_{1q}}{dt}+{\mathrm{\&omega;L}}_1{i}_{1d}\end{array}---\left(9\right)$

In formula, ω L ₁i _1qwith ω L ₁i _1dbe the dq component coupling terms of the first inductance L 1.

Formula (9) is written as state space form:

$\{\begin{array}{c}{\stackrel{\&CenterDot;}{x}}_1=A_1\&CenterDot;x_1+B_1\&CenterDot;u_1\\ y_1=C_1\&CenterDot;x_1\end{array}---\left(10\right)$

Wherein, x ₁=[i _1di _1q] ^t, u ₁=[u _1d-u _cdu _1q-u _cq] ^t, y ₁=[i _1di _1q] ^t, $A_1=\left[\begin{array}{cc}0& \mathrm{\&omega;}\\ -\mathrm{\&omega;}& 0\end{array}\right],$ $B_1=\left[\begin{array}{cc}1/L_1& 0\\ 0& 1/L_1\end{array}\right],C_1=\left[\begin{array}{cc}1& 0\\ 0& 1\end{array}\right].$

Obtain ssystem transfer function matrix:

$G\left(s\right)=\frac{y_1}{u_1}=\left[\begin{array}{cc}\frac{s}{L_1\&CenterDot;(s^2+{\mathrm{\&omega;}}^2)}& \frac{\mathrm{\&omega;}}{L_1\&CenterDot;(s^2+{\mathrm{\&omega;}}^2)}\\ \frac{-\mathrm{\&omega;}}{L_1\&CenterDot;(s^2+{\mathrm{\&omega;}}^2)}& \frac{s}{L_1\&CenterDot;(s^2+{\mathrm{\&omega;}}^2)}\end{array}\right]---\left(11\right)$

Transfer function matrix represented by formula (11) is non-diagonal matrix, shows that this two-output impulse generator linear time invariant system exists coupling, can introduce state feedback and realize diagonalization, and then realize the dynamic decoupling under stricti jurise.Fig. 5 is the system block diagram comprising state feedback, in Fig. 5, and H ₁for state feedback matrix.

After introducing state feedback, ssystem transfer function matrix can be obtained:

G′(s)＝C ₁·(sI-A ₁+B ₁H ₁) ^-1·B ₁(12)

For after guarantee decoupling zero, ssystem transfer function matrix is diagonal matrix, obtain:

$H_1=\left[\begin{array}{cc}0& {\mathrm{\&omega;L}}_1\\ -{\mathrm{\&omega;L}}_1& 0\end{array}\right]---\left(13\right)$

Transfer function matrix after decoupling zero:

$G_1\left(s\right)=\frac{i_1}{u_1^{\&prime;}}=\left[\begin{array}{cc}1/{\mathrm{sL}}_1& 0\\ 0& 1/{\mathrm{sL}}_1\end{array}\right]---\left(14\right)$

Wherein, $u_1^{\&prime;}=\left[\begin{array}{c}{u}_{1d}^{\&prime;}\\ u_{1q}^{\&prime;}\end{array}\right]=\left[\begin{array}{c}{u}_{1d}-u_{cd}-{\mathrm{\&omega;L}}_1{i}_{1q}\\ u_{1q}-u_{cq}+{\mathrm{\&omega;L}}_1{i}_{1d}\end{array}\right].$

After obtaining the first inductance L 1 decoupling zero, structure is illustrated in fig. 6 shown below.

As shown in Figure 7, Figure 8, adopting uses the same method carries out state feedback decoupling to electric capacity CJ, the second inductance L 2, and after the transfer function matrix after state feedback matrix, decoupling zero and decoupling zero, structure is as follows:

$H_2=\left[\begin{array}{cc}0& \mathrm{\&omega;}C\\ -\mathrm{\&omega;}C& 0\end{array}\right]---\left(15\right)$

$G_2\left(s\right)=\frac{u_c}{u_2^{\&prime;}}=\left[\begin{array}{cc}1/sC& 0\\ 0& 1/sC\end{array}\right]---\left(16\right)$

$H_3=\left[\begin{array}{cc}0& {\mathrm{\&omega;L}}_2\\ -{\mathrm{\&omega;L}}_2& 0\end{array}\right]---\left(17\right)$

$G_3\left(s\right)=\frac{i_2}{u_3^{\&prime;}}=\left[\begin{array}{cc}1/{\mathrm{sL}}_2& 0\\ 0& 1/{\mathrm{sL}}_2\end{array}\right]---\left(18\right)$

Wherein, $u_2^{\&prime;}=\left[\begin{array}{c}{i}_{1d}^{\&prime;}\\ i_{1q}^{\&prime;}\end{array}\right]=\left[\begin{array}{c}{i}_{1d}-i_{2d}-{\mathrm{\&omega;Cu}}_{cq}\\ i_{1q}-i_{2q}+{\mathrm{\&omega;Cu}}_{cd}\end{array}\right],u_3^{\&prime;}=\left[\begin{array}{c}{u}_{cd}^{\&prime;}\\ u_{cq}^{\&prime;}\end{array}\right]=\left[\begin{array}{c}{u}_{cd}-e_d-{\mathrm{\&omega;L}}_2{i}_{2q}\\ u_{cq}-e_q+{\mathrm{\&omega;L}}_2{i}_{2d}\end{array}\right].$

Structured flowchart after three equal decoupling zeros of state variable according to Fig. 8 carries out the design of controller, for realizing the method described in the present invention, need all to feed back two inductive currents and capacitance voltage, therefore three closed-loop structure shown in employing Fig. 1, and add in each feedback loop by the decoupling compensation amount analyzing gained above.Meanwhile, for realizing the tracking to networking electric current, outer shroud adopts PI controller, and middle ring and inner ring only need to adopt P controller, adopt three controllers to control a state variable respectively.

To sum up, a kind of LCL type inverter decoupling control method based on state feedback as shown in Figure 1, comprises the following steps:

S1: setting networking given value of current value i ₂real component and idle component, Real-time Collection monitoring parameter, monitoring parameter comprises networking current actual value i ₂, electric capacity CJ both end voltage actual value u _c, flow through the current actual value i of the first inductance L 1 ₁with line voltage e, and obtain real component corresponding to monitoring parameter and idle component.

S2: networking given value of current value i ₂real component and idle component obtain combining inverter output voltage set-point respectively by the closed-loop control of the state feedback of monitoring parameter and setup control parameter real component and idle component.Specific as follows:

Obtain combining inverter output voltage set-point real component comprise the following steps:

11) networking given value of current value i ₂real component deduct networking current actual value i ₂real component after carry out outer shroud PI control obtain after state feedback electric capacity CJ both end voltage value u _c' real component;

12) u _c' real component deduct the idle component coupling terms of the second inductance L 2 and the real component of e respectively after obtain electric capacity CJ both end voltage set-point real component;

13) electric capacity CJ both end voltage set-point real component deduct electric capacity CJ both end voltage actual value u _creal component after carry out middle ring P and control to obtain the current value i of first inductance after state feedback ₁' real component;

14) i ₁' real component deduct idle component coupling terms and the networking current actual value i of electric capacity CJ respectively ₂real component after obtain and flow through the given value of current value of the first inductance L 1 real component;

15) the given value of current value of the first inductance L 1 is flow through real component deduct the current actual value i flowing through the first inductance L 1 ₁real component after carry out inner ring P control obtain after state feedback combining inverter output voltage values u ₁' real component;

16) u ₁' real component deduct idle component coupling terms and the electric capacity CJ both end voltage actual value u of the first inductance L 1 respectively _creal component after obtain combining inverter output voltage set-point real component.

Obtain combining inverter output voltage set-point idle component comprise the following steps:

21) networking given value of current value i ₂idle component deduct networking current actual value i ₂idle component after carry out outer shroud PI control obtain after state feedback electric capacity CJ both end voltage value u _c' idle component;

22) u _c' idle component add the real component coupling terms of the second inductance L 2 after deduct the idle component of e, obtain electric capacity CJ both end voltage set-point idle component;

23) electric capacity CJ both end voltage set-point idle component deduct electric capacity CJ both end voltage actual value u _cidle component after carry out middle ring P and control to obtain the current value i of first inductance after state feedback ₁' idle component;

24) i ₁' idle component add the real component coupling terms of electric capacity CJ after deduct networking current actual value i ₂idle component, obtain and flow through the given value of current value of the first inductance L 1 idle component;

25) the given value of current value of the first inductance L 1 is flow through idle component deduct the current actual value i flowing through the first inductance L 1 ₁idle component after carry out inner ring P control obtain after state feedback combining inverter output voltage values u ₁' idle component;

26) u ₁' idle component add the real component coupling terms of the first inductance L 1 after deduct electric capacity CJ both end voltage actual value u _cidle component, obtain combining inverter output voltage set-point idle component.

Wherein, the real component coupling terms of the second inductance L 2 and idle component coupling terms are respectively ω L ₂i _2dwith ω L ₂i _2q, real component coupling terms and the idle component coupling terms of electric capacity CJ are respectively ω Cu _cdwith ω Cu _cq, real component coupling terms and the idle component coupling terms of the first inductance L 1 are respectively ω L ₁i _1dwith ω L ₁i _1d, ω is line voltage angular frequency, L ₂be the inductance value of the second inductance L 2, C is the capacitance of electric capacity CJ, L ₁be the inductance value of the first inductance L 1, subscript d represents real component, and subscript q represents idle component, eliminates these coupling terms in the inventive method.

According to the stability of a system, stability margin requirement, the Proportional coefficient K that outer shroud PI controls _p1, integral coefficient K _iand the Proportional coefficient K that middle ring P controls _p2with the Proportional coefficient K that inner ring P controls _p3meet following formula:

$K_{P2}=\frac{0.01K_{P3}^2-6}{2K_{P3}}---\left(1\right)$

$\sqrt{\frac{K_I{K}_{P2}{K}_{P3}}{L_1{\mathrm{CL}}_2}}=2\frac{K_I}{K_{P1}}---\left(2\right)$

$\sqrt{\frac{K_{P2}{K}_{P3}{L}_2+L_1+L_2}{L_1{\mathrm{CL}}_2}}\&CenterDot;\frac{K_{P1}}{K_I}=10---\left(3\right)$

In the present embodiment, selected filter parameter L ₁=2mH, C=20uF, L ₂=1mH, selects inner ring P controller parameter K simultaneously _p3=100, calculate the Proportional coefficient K of outer shroud PI controller _p1=15.045, integral coefficient K _i=53190, the Proportional coefficient K of middle ring P controller _p2=0.47.

S3: according to combining inverter output voltage set-point real component and idle component carry out PWM control, the switch control rule PWM ripple of output controls combining inverter.

In l-G simulation test, when setting emulation total time is 0.2s, 0.1s, given active power becomes 3.8kW from rated power 7.6kW step.Scheme proposed by the invention, directly ignore coupling terms scheme and compensate the scheme networking electric current under the same conditions of coupling terms and dq component thereof respectively as shown in Figure 9, Figure 10.Shown in Fig. 9, three kinds of scheme three-phase output networking electric current THD values are as shown in table 1 below, and contrasting known proposed scheme and compensating coupling terms scheme has higher output current quality; Simultaneously, shown in contrast Figure 10, during active power Spline smoothing, output current dq component waveform is known, when active power instruction changes, compensate between coupling terms scheme output current dq component and occur coupling influence, and proposed scheme and ignore coupling terms scheme and can effectively suppress this coupling influence.

Three kinds, table 1 different control strategy networking electric current THD analysis result

Therefore, comparatively conventional method is compared, and method proposed by the invention not only ensure that the accuracy of modeling process, guarantee higher networking current quality, and the coupling influence can effectively eliminated between dq component, the independence between realization output is meritorious and idle controls, and improves dynamic performance.

Claims (6)

1. the LCL type inverter decoupling control method based on state feedback, PWM for combining inverter controls, described combining inverter connects electricity consumption end by LCL filter, described LCL filter comprises the first inductance, electric capacity and the second inductance, it is characterized in that, this LCL type inverter decoupling control method comprises the following steps:

S1: setting networking given value of current value i ₂real component and idle component, Real-time Collection monitoring parameter, monitoring parameter comprises networking current actual value i ₂, electric capacity both end voltage actual value u _c, flow through the current actual value i of the first inductance ₁with line voltage e, and obtain real component corresponding to monitoring parameter and idle component;

S2: networking given value of current value i ₂real component and idle component obtain combining inverter output voltage set-point respectively by the state feedback of monitoring parameter and closed-loop control real component and idle component;

S3: according to combining inverter output voltage set-point real component and idle component carry out PWM control, the switch control rule PWM ripple of output controls combining inverter.

2. a kind of LCL type inverter decoupling control method based on state feedback according to claim 1, is characterized in that, obtain combining inverter output voltage set-point in described step S2 the method of real component be specially:

11) networking given value of current value i ₂real component deduct networking current actual value i ₂real component after carry out outer shroud PI control obtain after state feedback electric capacity both end voltage value u ' _creal component;

12) u ' _creal component deduct the idle component coupling terms of the second inductance and the real component of e respectively after obtain electric capacity both end voltage set-point real component;

13) electric capacity both end voltage set-point real component deduct electric capacity both end voltage actual value u _creal component after carry out middle ring P and control to obtain the current value i ' of first inductance after state feedback ₁real component;

14) i ' ₁real component deduct idle component coupling terms and the networking current actual value i of electric capacity respectively ₂real component after obtain and flow through the given value of current value of the first inductance real component;

15) the given value of current value of the first inductance is flow through real component deduct the current actual value i flowing through the first inductance ₁real component after carry out inner ring P control obtain after state feedback combining inverter output voltage values u ' ₁real component;

16) u ' ₁real component deduct idle component coupling terms and the electric capacity both end voltage actual value u of the first inductance respectively _creal component after obtain combining inverter output voltage set-point real component;

Obtain combining inverter output voltage set-point the method of idle component be specially:

21) networking given value of current value i ₂idle component deduct networking current actual value i ₂idle component after carry out outer shroud PI control obtain after state feedback electric capacity both end voltage value u ' _cidle component;

22) u ' _cidle component add the real component coupling terms of the second inductance after deduct the idle component of e, obtain electric capacity both end voltage set-point idle component;

23) electric capacity both end voltage set-point idle component deduct electric capacity both end voltage actual value u _cidle component after carry out middle ring P and control to obtain the current value i ' of first inductance after state feedback ₁idle component;

24) i ' ₁idle component add the real component coupling terms of electric capacity after deduct networking current actual value i ₂idle component, obtain and flow through the given value of current value of the first inductance idle component;

25) the given value of current value of the first inductance is flow through idle component deduct the current actual value i flowing through the first inductance ₁idle component after carry out inner ring P control obtain after state feedback combining inverter output voltage values u ' ₁idle component;

26) u ' ₁idle component add the real component coupling terms of the first inductance after deduct electric capacity both end voltage actual value u _cidle component, obtain combining inverter output voltage set-point idle component.

3. a kind of LCL type inverter decoupling control method based on state feedback according to claim 2, it is characterized in that, real component coupling terms and the idle component coupling terms of described second inductance are respectively ω L ₂i _2dwith ω L ₂i _2q, wherein, ω is line voltage angular frequency, L ₂be the inductance value of the second inductance, subscript d represents real component, and subscript q represents idle component.

4. a kind of LCL type inverter decoupling control method based on state feedback according to claim 2, it is characterized in that, real component coupling terms and the idle component coupling terms of described electric capacity are respectively ω Cu _cdwith ω Cu _cq, wherein, ω is line voltage angular frequency, and C is the capacitance of electric capacity, and subscript d represents real component, and subscript q represents idle component.

5. a kind of LCL type inverter decoupling control method based on state feedback according to claim 2, it is characterized in that, real component coupling terms and the idle component coupling terms of described first inductance are respectively ω L ₁i _1dwith ω L ₁i _1d, wherein, ω is line voltage angular frequency, L ₁be the inductance value of the first inductance, subscript d represents real component, and subscript q represents idle component.

6. a kind of LCL type inverter decoupling control method based on state feedback according to claim 2, is characterized in that, the Proportional coefficient K that described outer shroud PI controls _p1, integral coefficient K _iand the Proportional coefficient K that middle ring P controls _p2with the Proportional coefficient K that inner ring P controls _p3meet following formula:

$K_{P2}=\frac{0.01K_{P3}^2-6}{2K_{P3}}---\left(1\right)$

$\sqrt{\frac{K_I{K}_{P2}{K}_{P3}}{L_1{\mathrm{CL}}_2}}=2\frac{K_I}{K_{P1}}---\left(2\right)$

$\sqrt{\frac{K_{P2}{K}_{P3}{L}_2+L_1+L_2}{L_1{\mathrm{CL}}_2}}\&CenterDot;\frac{K_{P1}}{K_I}=10---\left(3\right)$

Wherein, L ₁be the inductance value of the first inductance, C is the capacitance of electric capacity, L ₂it is the inductance value of the second inductance.