CN104269869B - The proportional resonant control method of a kind of PWM converter relating to parameter optimization - Google Patents

Abstract

The present invention discloses the proportional resonant control method of a kind of PWM converter relating to parameter optimization. The implementation step of the method comparative example resonant controller in PWM converter has carried out overall design, gives the parameter optimization method of the ratio resonant controller comprising multiple resonant controller. The control needs of PWM converter when described method can not only meet desirable electrical network, it is particularly useful for unbalanced source voltage, passes through operation control containing the PWM converter under the complex electric network operating modes such as low electric harmonic, thus improve the fault operation of power networks ability of equipment. PR controller parameter of the present invention design (optimization) method is simple and feasible, is convenient to through engineering approaches and realizes, can conveniently be transplanted to the application scenarios such as PWM rectifier, combining inverter, active power filter, double-fed wind energy converter.

Description

The proportional resonant control method of a kind of PWM converter relating to parameter optimization

Technical field

The present invention relates to the proportional resonant control method of a kind of PWM converter relating to parameter optimization, effective control of PWM rectifier, PWM inverter or other grid-connected converters when not being only applicable to desired electrical net, the fault that more can improve such grid-connection device under the severe power grid environment such as voltage imbalance, harmonic distortion passes through service ability. Especially, The present invention gives the Parameters design of ratio resonant controller, it is to increase the engineering practicability of described control strategy.

Background technology

PWM converter has widespread use at active power filter, uninterruptible power supply, power-driven system and distributed generation system. No matter being be operated in rectification or inverter mode, during PWM converter controls, a key issue guarantees the rapidity of the DAZ gene of current-order (zero stable state) and regulate process.

Based in the vector control scheme of rotating coordinate transformation, the advantage such as proportional integral (PI) setter regulates because realizing the floating to DC quantity, be easy to tuning and robustness is good, the preferred regulator in Chang Zuowei PWM converter current control system. But this kind of current control scheme has two obviously deficiencies: when 1) there is uneven or harmonic distortion when controlled current flow, pi regulator is lower in high frequency part amplitude nargin, it is difficult to accomplish that quick, floating to current-order regulate; 2) for realizing the conversion of ac sampling signal to DC quantity, and the inverse transformation of direct current best friend's flow required before voltage instruction modulation, need a large amount of rotating coordinate transformations during Digital Implementation, take certain calculating resource.

In order to overcome above 2 deficiencies, the Linear Control device that one is called ratio resonance (proportionalresonant, PR) controller obtains extensive concern. The advantage of PR controller to realize control under static (�� ��) reference frame, and can provide the perfect Gain to positive sequence, the negative phase-sequence exchange signal of resonant frequency simultaneously, and then the floating realizing controlled signal regulates. The resonance portion of PR controller is also referred to as broad sense exchange integrator (generalizedacintegrator, GI), it is the core of controller, when adopting the resonant controller of multiple different resonant frequency in parallel, it is possible to realize concentrated, the quick adjustment to multiple harmonic component.

Although Preliminary Applications has been shown in the test application of PR controller in combining inverter, dynamic electric voltage recovery device, active power filtering device, double-fed wind energy converter etc., but how from system stability angle, this quasi-controller to be carried out parameter optimization, it is focus and the difficult point problem of industry concern always. Especially, when such current transformer grid-connected terminal voltage imbalance (or asymmetric fall), harmonic distortion, PR controller needs to contain multiple higher harmonic components in the signal regulated, and can whether the implementation step of PR controller be correct, the whether convenient key being such method and obtaining effectively execution of Parameters design.

Summary of the invention

The object of the invention is to overcome the deficiencies in the prior art, it is provided that a kind of meter relates to the proportional resonant control method of the PWM converter of parameter optimization, controls flow process by appropriate design, optimizes the parameter of PR controller, it is to increase the fault of PWM converter passes through service ability.

The present invention object be achieved through the following technical solutions: the proportional resonant control method of a kind of PWM converter relating to parameter optimization, comprises the following steps:

1. utilize one group of (three) voltage hall sensor to gather the three-phase voltage U of filter reactance device end of incoming cables_abc, utilize one group of (three) current Hall sensor to gather the tri-phase current I of filter reactance device end of incoming cables_abc, utilize a voltage hall sensor to gather the terminal voltage U of DC bus capacitance_dc;

2. the three-phase voltage U of filter reactance device end of incoming cables step 1 obtained_abcThrough CLARKE conversion, obtain the two-phase voltage U of filter reactance device end of incoming cables under static reference frame_����; The tri-phase current I of the filter reactance device end of incoming cables that step 1 is obtained_sabcThrough CLARKE conversion, obtain the feedback current I of electric current loop under static reference frame_����;

3. the two-phase voltage U of filter reactance device end of incoming cables under static reference frame step 2 obtained_����Send into traditional phaselocked loop (PLL) based on rotating forward synchronous speed rotating frame, obtain the angular velocity omega of electrical network voltage₁With the angular position theta of electrical network voltage;

4. by the voltage instruction of DC bus capacitanceSubtract the terminal voltage U of the DC bus capacitance that step 1 obtains_dc, obtain DC bus-bar voltage error delta U_dc, by �� U_dcSend into Voltage loop proportional integral (PI) setter, obtain the wattful current instruction of electric current loop

5. the wattful current instruction of electric current loop step 4 obtainedWith the referenced reactive current of electric current loopCarry out vector summing, obtain the resultant current instruction of electric current loop

6. utilize the angular position theta of the electrical network voltage that step 3 obtains step 5 to be obtainedCarry out PARK inverse transformation, obtain the reference electric current of electric current loop under static reference frame

7. the reference electric current of electric current loop step 6 obtainedSubtract the feedback current I of the electric current loop that step 2 obtains_����, obtain electric current loop tracking errors �� I_����, by �� I_����Feeding ratio resonance (PR) controller, obtains the output voltage E of ratio resonant controller_����;

Described ratio resonance (PR) controller comprises: a proportioning controller and 4 resonant frequencies are respectively fundamental frequency (50Hz, k=1), 5 frequency multiplication (250Hz, k=5), 7 frequency multiplication (350Hz, k=7), 11 frequency multiplication (550Hz, k=11) resonant controller, its transport function is:

$G_{\mathrm{PR}}\left(s\right)=K_p+\underset{k=\mathrm{1,5,7,11}}{\mathrm{\Σ}}\frac{K_{\mathrm{rk}}s}{s^2+{\left(k{\mathrm{\ω}}_1\right)}^2};$

In formula, k is harmonic order, K_pFor the scale-up factor (or proportioning controller) of PR controller, K_rkFor the resonance coefficient of k frequency multiplication resonant controller in PR controller;

The parameter K of described ratio resonance (PR) controller_p��K_rkDesign carry out in two steps:

(1) root locus method is utilized to optimize the Proportional coefficient K of PR controller_pWith the resonance coefficient K of fundamental frequency resonant controller_r1, namely obtained the closed loop root locus of electric current loop by the open loop transfer function of electric current loop, choose the limit of subsidence ratio �� on root locus=[0.4,0.8], calculate the interval of now electric current loop open-loop gain, and then obtain K_p��K_r1Interval;

(2) multiple frequence resonant controller bandwidth omega is made_bwIt is taken as (0.5��0.7) ��₁, utilize the resonance coefficient K of frequency domain analysis method design multiple frequence resonant controller (k=5 or 7 or 11)_rk, its accounting equation is:

$K_{\mathrm{rk}}=\frac{1}{{\mathrm{\ω}}_o}\sqrt{{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2({\mathrm{\ω}}_o^2{L}_g^2+R_g^2)-K_p^2{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2};$

In formula: ��_o=k ��₁+��_bw/ 2, it is the limiting frequency of multiple frequence resonant controller; L_g��R_gIt is respectively inductance and the resistance of filter reactance device;

8. the two-phase voltage U of filter reactance device end of incoming cables step 2 obtained_����Subtract the output voltage E of the ratio resonant controller that step 7 obtains_����, obtain required space vector modulation voltage V_����;

9. V step 8 obtained_����Carry out space vector modulation (SVPWM), the switch signal of PWM converter can be obtained, it is achieved to effective control of PWM converter.

The invention has the beneficial effects as follows:

1) the operation control of PWM converter when described control method can not only meet desirable electrical network, also it is applicable to unbalanced source voltage, stability contorting containing current transformer under the complex electric network operating modes such as low electric harmonic, thus improves equipment and pass through service ability when fault electrical network;

2) described PR controller parameter design (optimization) scheme not only discusses the parameter designing of fundamental frequency resonant controller, more gives the parameter optimization step of multiple frequence resonant controller; The method is simple and feasible, is convenient to through engineering approaches and realizes, can conveniently be transplanted to the optimization control field of the equipment such as PWM rectifier, invertor, active power filter, double-fed wind energy converter.

Accompanying drawing explanation

Fig. 1 is the PWM rectifier control texture figure that certain rated capacity utilizing the present invention to control method is 5kW;

In figure, the voltage hall sensor 3 of the voltage hall sensor 1 of AC side, the current Hall sensor 2 of AC side, DC side, CLARKE conversion 4, phaselocked loop (PLL) 5, Voltage loop proportional integral (PI) setter 6, PARK inverse transformation 7, ratio resonance (PR) controller 8, DC bus capacitance C, DC side load R_L, space vector modulation SVPWM.

Fig. 2 is different current controller timeconstant��s_iLower electric current loop closed loop root locus bunch.

Fig. 3 is the electric current loop open-loop amplitude phase frequency rational curve frequently of PWM rectifier.

Fig. 4 is the experimental waveform that the voltage symmetry utilizing the present invention to obtain falls PWM rectifier under fault; Wherein, the experimental result of Fig. 4 (A) for obtaining after PR controller parameter optimization, Fig. 4 (B) is the resonance coefficient K of PR controller in Fig. 4 (A)_r1The experimental result obtained after reducing 10 times, Fig. 4 (C) is PR controller resonance coefficient K in Fig. 4 (A)_r1The experimental result obtained after increasing 10 times.

Fig. 5 is that the asymmetric experimental waveform falling PWM rectifier under fault occurs for utilize the present invention to control electrical network voltage that method obtains.

The experimental waveform of PWM rectifier when Fig. 6 is contain 5 subharmonic composition in electrical network voltage; Wherein, Fig. 6 (A) is for PR controller is only containing the experimental result obtained during a fundamental frequency resonator, and Fig. 6 (B) is for PR controller is simultaneously containing the experimental result obtained when a fundamental frequency resonator and a 5 frequency multiplication resonator.

Embodiment

Below in conjunction with accompanying drawing and case study on implementation, the invention will be further described.

Fig. 1 represents the PWM rectifier control texture figure that certain rated capacity utilizing the present invention to control method is 5kW. In figure, U_abc��I_abcIt is respectively the three-phase voltage of filter reactance device end of incoming cables, the tri-phase current of filter reactance device end of incoming cables, U ��_��For two phase voltages of filter reactance device end of incoming cables under static coordinate system, U_dcFor the terminal voltage of DC bus capacitance,For the voltage instruction of DC bus capacitance, �� U_dcFor DC bus-bar voltage error,For the wattful current instruction of electric current loop,For the referenced reactive current of electric current loop,For the resultant current instruction of electric current loop,For the reference electric current of electric current loop, �� I_����For electric current loop tracking errors, E_����For the output voltage of ratio resonant controller, V_����For space vector modulation voltage, ��₁For the circular frequency of electrical network voltage, �� is the angle, position of electrical network voltage.

With reference to Fig. 1, the proportional resonant control method of a kind of PWM converter relating to parameter optimization described in the invention comprises the following steps:

1. utilize one group of (three) voltage hall sensor 1 to gather the three-phase voltage U of filter reactance device end of incoming cables_abc, utilize one group of (three) current Hall sensor 2 to gather the tri-phase current I of filter reactance device end of incoming cables_abc, utilize a voltage hall sensor 3 to gather the terminal voltage U of DC bus capacitance_dc;

2. the three-phase voltage U of filter reactance device end of incoming cables step 1 obtained_abcConvert 4 through CLARKE, obtain the two-phase voltage U of filter reactance device end of incoming cables under static reference frame_����; The tri-phase current I of the filter reactance device end of incoming cables that step 1 is obtained_sabcConvert 4 through CLARKE, obtain the feedback current I of electric current loop under static reference frame_����;

With U_abcCLARKE be transformed to example, its conversion process can represent and is:

$U_{\mathrm{\α\β}}=\frac{2}{3}\left[\begin{array}{ccc}1& -\frac{1}{2}& -\frac{1}{2}\\ 0& \frac{\sqrt{3}}{2}& \frac{\sqrt{3}}{2}\end{array}\right]\·U_{\mathrm{abc}};$

3. the two-phase voltage U of filter reactance device end of incoming cables under static reference frame step 2 obtained_����Send into traditional phaselocked loop (PLL) 5 based on rotating forward synchronous speed rotating frame, obtain the angular velocity omega of electrical network voltage₁With the angular position theta of electrical network voltage;

4. by the voltage instruction of DC bus capacitanceSubtract the terminal voltage U of the DC bus capacitance that step 1 obtains_dc, obtain DC bus-bar voltage error delta U_dc, by �� U_dcSend into Voltage loop proportional integral (PI) setter 6, obtain the wattful current instruction of electric current loop

5. the wattful current instruction of electric current loop step 4 obtainedWith the referenced reactive current of electric current loopCarry out vector summing, obtain the resultant current instruction of electric current loop

Vector summing algorithm can represent:

$I_{\mathrm{dq}}^{*}=I_d^{*}+j\·I_q^{*};$

6. utilize the angular position theta of the electrical network voltage that step 3 obtains step 5 to be obtainedCarry out PARK inverse transformation 7, obtain the reference electric current of electric current loop under static reference frame

Wherein, PARK inverse transformation process can represent and is:

$I_{\mathrm{\α\β}}^{*}=\left[\begin{array}{cc}\cos \mathrm{\θ}& -\sin \mathrm{\θ}\\ \sin \mathrm{\θ}& \cos \mathrm{\θ}\end{array}\right]\·I_{\mathrm{dq}}^{*};$

7. the reference electric current of electric current loop step 6 obtainedSubtract the feedback current I of the electric current loop that step 2 obtains_����, obtain electric current loop tracking errors �� I_����, by �� I_����Feeding ratio resonance (PR) controller 8, obtains the output voltage E of ratio resonant controller_����;

Described ratio resonance (PR) controller 8 comprises: a proportioning controller and 4 resonant frequencies are respectively fundamental frequency (50Hz, k=1), 5 frequency multiplication (250Hz, k=5), 7 frequency multiplication (350Hz, k=7), 11 frequency multiplication (550Hz, k=11) resonant controller, its transport function is:

$G_{\mathrm{PR}}\left(s\right)=K_p+\underset{k=\mathrm{1,5,7,11}}{\mathrm{\Σ}}\frac{K_{\mathrm{rk}}s}{s^2+{\left(k{\mathrm{\ω}}_1\right)}^2};$

In formula, k is harmonic order, K_pFor the scale-up factor (or proportioning controller) of PR controller, K_rkFor the resonance coefficient of k frequency multiplication resonant controller in PR controller;

The parameter designing of described ratio resonance (PR) controller comprises following two steps:

7.1 utilize the Proportional coefficient K of root locus method design PR controller_pWith the resonance coefficient K of fundamental frequency resonator (k=1)_r1, namely obtained the closed loop root locus of electric current loop by the open loop transfer function of electric current loop, choose the limit of subsidence ratio �� on root locus=[0.4,0.8], obtain the interval of now electric current loop open-loop gain, thus calculate K_p��K_r1; It is specially:

When PR controller is only containing fundamental frequency resonator (k=1), its transport function G_PRS () can simplify and be expressed as:

$G_{\mathrm{PR}}\left(s\right)=K_p+\frac{K_{r1}s}{s^2+{\mathrm{\ω}}_1^2};$

The then electric current loop open loop transfer function H of PWM rectifier in Fig. 1_OLS () has following expression:

$\begin{array}{c}{H}_{\mathrm{OL}}\left(s\right)=G_{\mathrm{PR}}\left(s\right)\·G_p\left(s\right)\\ =\frac{K_p{s}^2+K_{i}s+K_p{\mathrm{\ω}}_1^2}{(s^2+{\mathrm{\ω}}_1^2)(s{L}_g+R_g)}\end{array};$

In formula: G_pS transport function that () is PWM rectifier, can represent in the implementation case and be:

$G_p\left(s\right)=\frac{1}{s{L}_g+R_g};$

In formula: L_g��R_gIt is respectively inductance and the resistance of filter reactance device;

Make ��_i=K_p/K_i, it is the equivalent time constant of PR controller, then electric current loop open loop transfer function H_OLS () can represent again:

$H_{\mathrm{OL}}\left(s\right)=\frac{K(s^2+1/{\mathrm{\τ}}_i\·s+{\mathrm{\ω}}_1^2)}{(s^2+{\mathrm{\ω}}_1^2)({\mathrm{sL}}_g+R_g)};$

Like this, such as known ��_iSpan, then can obtain the root locus of electric current loop when electric current loop open loop equivalent gain K increases to infinity by zero. For the implementation case, the inductance value L of filter reactance device_g=3.5mH, resistance value R_g=0.5 ��, then the timeconstant�� of filter reactance device_l=L_g/R_g=5ms. Current controller design based on pole zero cancellation thought makes �� often_i=��_l. Here �� can be made_iWith ��_lValue, at the same order of magnitude, gets ��_iIt is respectively 2ms, 4ms, 6ms and 8ms. Like this according to the open loop transfer function H of electric current loop_OLS (), can obtain the closed loop root locus bunch of electric current loop shown in Fig. 2, in figure, and p₁��p₂And p₃Represent the limit of electric current loop; z₁�� z₂And z₃Represent the zero point of electric current loop.

With reference to Fig. 2, when electric current loop open loop equivalent gain K increases to infinite by zero, the equivalent time constant, �� of 4 PR controllers_iUnder, the characteristic root of system is distributed in the Left half-plane in s territory all the time, and illustrative system is all stable. But different ��_iUnder value, the damping characteristic of electric current loop but has notable difference: ��_iGet 4ms, when 6ms, 8ms, no matter K how value, the subsidence ratio of electric current loop is less than 0.4 all the time; And ��_iWhen getting 2ms, the subsidence ratio of electric current loop can more than 0.8. For general Controlling System, often wishing that subsidence ratio is selected within 0.4��0.8, at this moment system overshoot is little, and response speed is also very fast. A little obtain by root locus is got, at ��_iIn=2ms situation, can obtaining the damping characteristic expected as K �� [2.38,5.77], corresponding PR controller scale-up factor span is K_p�� [0.6,1.4], the span of the resonance coefficient of fundamental frequency resonator is K_r1�� [300,700]. This is K in this case study on implementation_p��K_r1Desirable span.

7.2 make multiple frequence resonant controller bandwidth omega_bwIt is taken as (0.5��0.7) ��₁, utilize the resonance coefficient K of frequency domain analysis method design multiple frequence resonant controller (k=5 or 7 or 11)_rk, its accounting equation is:

$K_{\mathrm{rk}}=\frac{1}{{\mathrm{\ω}}_o}\sqrt{{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2({\mathrm{\ω}}_o^2{L}_g^2+R_g^2)-K_p^2{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2};$

In formula: ��_o=k ��₁+��_bw/ 2, it is the limiting frequency of multiple frequence resonant controller;

K in this step_rkThe origin of accounting equation is:

At the limiting frequency �� of multiple frequence resonant controller_o=k ��₁+��_bw/ 2 places, the open loop transfer function H of electric current loop_OLS the absolute value of the amplitude domain degree of () is 1, namely have:

$\left|H_{\mathrm{OL}}\left({\mathrm{j\ω}}_o\right)\right|=\left|\frac{-K_p{\mathrm{\ω}}_o^2+{\mathrm{j\ω}}_o{K}_{\mathrm{rk}}+K_p{k}^2{\mathrm{\ω}}_1^2}{(-{\mathrm{\ω}}_o^2+k^2{\mathrm{\ω}}_1^2)({\mathrm{j\ω}}_o{L}_g+R_g)}\right|=1;$

Arrive resonance coefficient K can be obtained fom the above equation_rkAccounting equation, namely

$K_{\mathrm{rk}}=\frac{1}{{\mathrm{\ω}}_o}\sqrt{{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2({\mathrm{\ω}}_o^2{L}_g^2+R_g^2)-K_p^2{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2};$

According to the PR controller Proportional coefficient K that step 7.1 obtains_p, fundamental frequency resonant controller resonance coefficient K_r1Span, the scale-up factor of selected one group of PR controller, the resonance coefficient of fundamental frequency resonator be: K_p=0.8, K_r1=400; With multiple frequence resonant controller bandwidth omega in season_bw=0.6 ��₁. Then according to the resonance coefficient K of multiple frequence resonator_rkAccounting equation, can obtain in the implementation case 5 corresponding frequencys multiplication, 7 frequencys multiplication, 11 frequency multiplication resonators resonance coefficient be respectively: K_r5=473, K_r7=701, K_r11=1135.

It should be noted that, step 7.1 utilizes root locus method to obtain the resonance coefficient of fundamental frequency resonator (k=1), but this method and be not suitable for the asking for of resonance coefficient of multiple frequence resonant controller. This is because now electric current loop open loop transfer function contains multiple variable to be optimized, its equivalence open-loop gain K not easily obtains. Considering that the resonant frequency 250Hz of 5 frequencys multiplication (or other high frequencys multiplication) resonator is away from the resonant frequency 50Hz of fundamental frequency resonator, its width frequency, phase frequency rational curve can be similar to one " flex point " regarding as on fundamental frequency resonant frequency response characteristic extended line. Now, as the coefficient of multiple frequence resonator being optimized and will become very simple and feasible in conjunction with frequency domain analysis method.

8. the two-phase voltage U of filter reactance device end of incoming cables step 2 obtained_����Subtract the output voltage E of the ratio resonant controller that step 7 obtains_����, obtain required space vector modulation voltage V_����;

9. V step 8 obtained_����Carry out space vector modulation (SVPWM), the switch signal of PWM rectifier (current transformer) can be obtained, it is achieved to effective control of PWM rectifier.

Fig. 3 gives the width phase frequency rational curve frequently of electric current loop open loop transfer function in the implementation case. From amplitude-frequency characteristic curve, after 3 multiple frequence (5 frequencys multiplication, 7 frequencys multiplication, 11 frequencys multiplication) resonators are introduced, fundamental frequency (50Hz) electric current signal can not only be provided the perfect Gain by PR controller, also can 5 times, 7 times and 11 order harmonic components fully be regulated simultaneously, and from amplitude-frequency characteristic curve tendency, 3 harmonic controllers are except causing one " point peak " at resonant frequency annex, do not change the main PR controller width characteristic general trend of phase frequency frequently, this also illustrates the reasonableness of above-mentioned multiple frequence frequency resonator parameter optimization method.

Fig. 4 is the experimental waveform that the voltage symmetry utilizing the present invention to obtain falls PWM rectifier under fault. Wherein, Fig. 4 (A) is (K after PR controller parameter optimization_p=0.8, K_r1=400) experimental result obtained, the resonance coefficient that Fig. 4 (B) is PR controller reduces 10 times of (K_p=0.8, K_r1=40) experimental result obtained, the resonance coefficient that Fig. 4 (C) is PR controller increases 10 times of (K_p=0.8, K_r1=4000) experimental result obtained. In figure, U_abFor two phase voltages of filter reactance device end of incoming cables, �� I_������I_��It is respectively �� axle, the beta-axis component of electric current loop tracking errors; The same Fig. 1 of other symbol implications. I from figure_abcWaveform is visible, compares Fig. 4 (C), and in Fig. 4 (A), Fig. 4 (B), the tri-phase current of filter reactance device end of incoming cables is more symmetrical, sinusoidal; Meanwhile, occur from fault and recover rear �� I_������I_��Waveform visible, compare Fig. 4 (B), in Fig. 4 (A), Fig. 4 (C), the dynamic adjustments time of PWM rectifier is relatively short; In general, Fig. 4 (A), namely adopts the PR controller after parameter optimization can obtain desirable steady-state current waveform and dynamic responding speed faster, thus describes the reasonableness of Parameters design of the present invention.

Fig. 5 is that the asymmetric experimental waveform falling PWM rectifier under fault occurs for utilize the present invention to control electrical network voltage that method obtains. �� I from figure_������I_��Waveform it may be seen that PR controller can realize quick, the floating to electric current positive sequence component, negative phase-sequence component regulates simultaneously.

Fig. 6 is the experimental waveform of electrical network voltage PWM rectifier when containing 5 subharmonic composition. Wherein, Fig. 6 (A) is for PR controller is only containing the experimental result obtained during a fundamental frequency resonator, and Fig. 6 (B) is for PR controller is simultaneously containing the experimental result obtained when a fundamental frequency resonator and a 5 frequency multiplication resonator. Obtain through Fourier analysis, I in Fig. 6 (A)_abcTotal harmonic distortion rate (THD) be 13.19%, and I in Fig. 6 (B)_abcTotal harmonic distortion rate (THD) be only 2.87%. Obviously, after adding 5 frequency multiplication resonators, the humorous wave component in electric current is significantly suppressed. This has also confirmed the validity of 5 frequency multiplication resonator parameter designing of the present invention again.

To sum up, a kind of meter of the present invention relates to the proportional resonant control method of the PWM converter of parameter optimization, can not only realize ideal electrical network when PWM converter operation control, also it is applicable to unbalanced source voltage (or asymmetric fall), containing the stability contorting under the complex electric network operating modes such as low electric harmonic, contributes to improving the fault operation of power networks ability of PWM converter; Especially, PR controller parameter of the present invention design (optimization) method is simple and feasible, is extremely conducive to the through engineering approaches of this quasi-controller to realize.

Being described for PWM rectifier though the present invention controls method, its design implementation step, particularly parameter optimization method are equally applicable to the control occasion of PWM inverter or other grid-connected converters.

Claims (3)

1. one kind relates to the proportional resonant control method of the PWM converter of parameter optimization, it is characterised in that, comprise the following steps:

A1. the three-phase voltage U that three is one group of voltage hall sensor collection filter reactance device end of incoming cables of a group is utilized_abc, utilize the tri-phase current I that three is one group of current Hall sensor collection filter reactance device end of incoming cables of a group_abc, utilize a voltage hall sensor to gather the terminal voltage U of DC bus capacitance_dc;

The three-phase voltage U of the filter reactance device end of incoming cables A2. steps A 1 obtained_abcThrough CLARKE conversion, obtain the two-phase voltage U of filter reactance device end of incoming cables under static reference frame_����; The tri-phase current I of the filter reactance device end of incoming cables that steps A 1 is obtained_abcThrough CLARKE conversion, obtain the feedback current I of electric current loop under static reference frame_����;

A3. the two-phase voltage U of filter reactance device end of incoming cables under static reference frame steps A 2 obtained_����Send into traditional phaselocked loop based on rotating forward synchronous speed rotating frame, obtain the angular velocity omega of electrical network voltage₁With the angular position theta of electrical network voltage;

A4. by the voltage instruction of DC bus capacitanceSubtract the terminal voltage U of the DC bus capacitance that steps A 1 obtains_dc, obtain DC bus-bar voltage error delta U_dc, by �� U_dcSend into Voltage loop proportional and integral controller, obtain the wattful current instruction of electric current loop

The wattful current instruction of the electric current loop A5. steps A 4 obtainedWith the referenced reactive current of electric current loopCarry out vector summing, obtain the resultant current instruction of electric current loop

A6. the angular position theta of the electrical network voltage that steps A 3 obtains is utilized step 5 to be obtainedCarry out PARK inverse transformation, obtain the reference electric current of electric current loop under static reference frame

The reference electric current of the electric current loop A7. steps A 6 obtainedSubtract the feedback current I of the electric current loop that steps A 2 obtains_����, obtain electric current loop tracking errors �� I_����, by �� I_����Feeding ratio resonant controller, obtains the output voltage E of ratio resonant controller_����;

The two-phase voltage U of the filter reactance device end of incoming cables A8. steps A 2 obtained_����Subtract the output voltage E of the ratio resonant controller that steps A 7 obtains_����, obtain required space vector modulation voltage V_����;

A9. V steps A 8 obtained_����Carry out space vector modulation, obtain the switch signal of PWM converter, the effective control to PWM converter can be realized.

2. the proportional resonant control method of a kind of PWM converter relating to parameter optimization according to claim 1, it is characterized in that, in described steps A 7, described ratio resonant controller comprises a proportioning controller and 4 resonant frequencies are respectively the resonant controller of fundamental frequency, 5 frequencys multiplication, 7 frequencys multiplication, 11 frequencys multiplication, and its transport function is:

$G_{PR}\left(s\right)=K_p+\underset{k=1,5,7,11}{\Σ}\frac{K_{rk}s}{s^2+{\left({\mathrm{k\ω}}_1\right)}^2};$

In formula, k is harmonic order, K_pFor the scale-up factor of ratio resonant controller, K_rkFor the resonance coefficient of k frequency multiplication resonant controller.

3. the proportional resonant control method of a kind of PWM converter relating to parameter optimization according to claim 2, it is characterised in that, described ratio resonant controller comprises following parameter designing step:

B1. root locus method is utilized to obtain the Proportional coefficient K of ratio resonant controller_pWith the resonance coefficient K of fundamental frequency resonant controller_r1; It is specially: the closed loop root locus being obtained electric current loop by the open loop transfer function of electric current loop, chooses the limit of subsidence ratio �� on root locus=[0.4,0.8], obtain the interval of now electric current loop open-loop gain, and then K can be obtained_p��K_r1Span;

B2. multiple frequence resonant controller bandwidth omega is made_bwIt is taken as (0.5��0.7) ��₁, utilize the resonance coefficient K of frequency domain analysis method design multiple frequence resonant controller_rk, its accounting equation is:

$K_{rk}=\frac{1}{{\mathrm{\ω}}_o}\sqrt{{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2({\mathrm{\ω}}_o^2{L}_g^2+R_g^2)-K_p^2{(k^2{\mathrm{\ω}}_1^2-{\mathrm{\ω}}_o^2)}^2};$

In formula: ��_o=k ��₁+��_bw/ 2, it is the limiting frequency of multiple frequence resonant controller; L_g��R_gIt is respectively inductance and the resistance of filter reactance device.