CN103856432B

CN103856432B - Micro-strip resonance coherent demodulator for AMPSK modulating signals

Info

Publication number: CN103856432B
Application number: CN201410002001.XA
Authority: CN
Inventors: 郑祖翔; 吴乐南
Original assignee: Southeast University
Current assignee: Southeast University
Priority date: 2014-01-03
Filing date: 2014-01-03
Publication date: 2017-05-10
Anticipated expiration: 2034-01-03
Also published as: CN103856432A

Abstract

The invention discloses a micro-strip resonance coherent demodulator for AMPSK modulating signals. After the demodulator directly extracts a coherent carrier from analog signals received through an antenna, the coherent carrier and the received AMPSK modulating signals are superposed, wherein the coherent carrier and a received signal carrier are the same in frequency and strictly opposite in phase. Therefore, an AMPSK modulating signal carrier is restrained, an AMPSK phase modulation time interval is expanded, the signals are demodulated, no analog-digital conversion or down-conversion needs to be carried out on the received signals, and the demodulator can complete coherent carrier extraction and AMPSK modulating signal coherent demodulation only through a micro-strip open-circuit line. According to the micro-strip resonance coherent demodulator for the AMPSK modulating signals, on the premise that the advantages of being high in the spectrum utilization rate of the AMPSK modulating signals and high in information transmission rate are kept, the problems that a traditional demodulation method based on a digital shock filter is limited by analog-digital conversion and the processing rate, and the microwave frequency band modulation signals are hard to demodulate are solved.

Description

Microstrip resonance coherent demodulator for AMPSK modulation signal

Technical Field

The invention relates to a micro-strip resonance coherent demodulator for AMPSK modulation signals, which is a demodulator for directly and coherently demodulating the AMPSK modulation signals in a microwave frequency band by using a micro-strip resonator and belongs to the signal receiving and demodulating technology in digital communication.

Background

The rapidly growing demand for broadband wireless services puts increasing demands on wireless communication, which directly results in the increasing congestion of the radio frequency in the air, especially as third generation (3G) and fourth generation (4G) broadband mobile communication networks develop, the continuous spectrum of the lower frequency band is almost exhausted. The auction price of 20-year usage right of European 10MHz spectrum is up to 40 hundred million Euros, and in China, the frequency point and bandwidth of 'gold frequency band' below 1GHz are hard to buy at a cost. Therefore, like energy and water resources, spectrum is also an important strategic resource of the country, compressing the wireless transmission spectrum to the maximum extent has important practical significance and direct economic benefit, and the spectrum utilization rate has become a core competitive index and a key common technology of a new generation of information transmission systems.

1. Asymmetric binary phase shift keying modulation

In digital communication systems, the process of moving up a baseband signal representing binary data to a given transmission frequency band is called modulation, while the reverse process is called demodulation. In order to improve the spectrum utilization, a series of binary phase shift keying modulation methods with asymmetric modulation periods of data "0" and "1" have been developed, such as:

the Chinese patent number is ZL 200710025203.6, the invention name is 'unified orthogonal Binary shift Keying modulation and demodulation method', the disclosed unified Asymmetric Binary Phase Shift Keying (ABPSK) modulation;

② China patent number ZL200910033322.5, invention name 'spectrum tightening extension Binary Phase Shift Keying modulation and demodulation method', discloses continuous Phase extension Binary Phase Shift Keying (CP-EBPSK) modulation and a plurality of variants thereof.

In the chinese patent application No. 201210243474.X entitled "digital filter bank for demodulating multiple ABPSK signals", the above two modulations are collectively expressed as:

s₀(t)＝Asinω_ct，0≤t＜T

wherein s is₀(t) and s₁(t) modulation waveforms representing symbols "0" and "1", respectively; omega_cFor carrier angular frequency, T_c＝2π/ω_cFor the carrier period, T ═ NT_cFor a symbol period, τ ═ KT_cFor modulation intervalB-A is the amplitude of carrier keying, and sigma is the phase of carrier keying, when the modulated waveform is a hard jump, sigma ∈ [0, pi]When the modulation waveform is continuous, σ ═ ξ · Δ sin (η× 2 π f)_ct), 0 ≦ Δ ≦ 1, 0 ≦ η ≦ 1, and the value of ξ∈ { -1,1}, i.e., the phase modulation polarity, may be controlled with a pseudo-random sequence.

2. Asymmetric multi-element phase shift keying modulation

If the position of the modulation interval tau in the code element period T in the multivariate information symbol Keying (1) formula is used, a series of Asymmetric Multivariate Phase Shift Keying (AMPSK) modulations can be obtained, and the expression is as follows:

wherein s is_k(t) represents a modulation waveform of symbol "k", k being 0,1, …, M-1; r is_gIs a symbol guard interval control factor, r is more than or equal to 0_gLess than 1; the remaining parameters are defined as in formula (1). From r_gAnd the integer M, N, K constitute a "modulation parameter" that varies the signal bandwidth, transmission efficiency, and demodulation performance.

According to the contents of chinese patent No. ZL200710025202.1 and the invention named "multiple Phase Shift Keying modulation and demodulation method", taking the Phase modulation angle σ ═ pi and a ═ B ═ 1, one of the most commonly used multiple Phase Shift Keying (MPPSK) can be obtained, and the expression is as follows:

in particular, when M ═ 2 and r_gAt 0, MPPSK modulation degenerates to the common Extended binary phase shift keying (EBPSK: Extended BPSK) modulation, which is expressed as follows:

3. AMPSK/ABPSK modulated signal demodulation

The power spectrum of the AMPSK/ABPSK modulated signal shows the distinct characteristics of high carrier and low sidebands, and a very high spectrum utilization rate can be obtained, as shown in fig. 1. On the other hand, however, the waveform difference between data "0" and "non-0" of these modulated signals is small, which brings about a great challenge to demodulation.

Chinese patent No. ZL200910029875.3, entitled "impact filtering method for enhancing asymmetric binary modulation signal", proposes an Infinite Impulse Response (IIR) narrow-band digital band-pass filter based on a class of special designs, which is composed of a single zero point and a multi-pole point, and can present an extremely narrow notch-frequency selection characteristic at the center frequency thereof, so that the filtering output waveform of the input modulation signal generates an obvious and strong parasitic amplitude modulation impact at the information modulation position, as shown in fig. 2, the difference between "0" and "non-0" symbols is highlighted, which facilitates the demodulation decision of the modulation signal, and is called as a digital impact filter.

As can be seen from expressions (1) to (4), T is NT for the same symbol period_c＝2πN/ω_cThe transmission code rate of ABPSK modulation is:

R_bB＝f_c/N (5a)

and the transmission code rate of AMPSK modulation is increased to:

R_bM＝(f_c/N)log₂M＝(log₂M)R_bB(5b)

thus, the carrier frequency f of the signal_cThe higher the code rate of the AMPSK/ABPSK modulated signal, the more suitable this regime is for operation at higher frequencies, such as Radio Frequency (RF) or high Intermediate Frequency (IF).

However, in order to perform digital processing on a communication signal, an analog-to-digital converter (ADC) is first used to convert a received analog signal into a digital signal, but the ADC is generally large in size, high in power consumption, and expensive, and when a signal carrier frequency is increased to a microwave or millimeter wave frequency band, a typical ADC device cannot directly sample a radio frequency signal, and the advantages of the digital impact filter cannot be achieved. Therefore, in order to take advantage of the AMPSK/ABPSK modulation in higher frequency bands, a completely new demodulation scheme suitable for microwave and millimeter wave band receivers must be explored.

Disclosure of Invention

The purpose of the invention is as follows: in order to overcome the defects in the prior art, the invention provides the micro-strip resonance coherent demodulator for the AMPSK modulation signal, which is a brand new demodulation scheme suitable for microwave and millimeter wave frequency band receivers and can play the great advantages of AMPSK/ABPSK modulation in higher frequency bands.

The technical scheme is as follows: in order to achieve the purpose, the invention adopts the technical scheme that:

the demodulator is used for directly demodulating the analog signals received by the antenna and does not perform analog-to-digital conversion or down-conversion on the received analog signals; the demodulation method comprises the following steps: the demodulator directly extracts a coherent carrier which is the same frequency as and strictly opposite to a received signal carrier from an analog signal received by the antenna, and then superposes the coherent carrier with the received AMPSK modulation signal, so that the AMPSK modulation signal carrier is inhibited, the phase modulation time interval is amplified, and the signal demodulation is realized.

The method can simultaneously complete the extraction of the coherent carrier wave and the coherent demodulation of the AMPSK modulation signal only by depending on a section of microstrip open line.

The AMPSK modulated signal is transmitted in a symbol period NT_cThe simplified expression within is:

wherein s is_k(t) represents a modulation waveform of symbol "k", k being 0,1, …, M-1; r is_gIs a symbol guard interval control factor, r is more than or equal to 0_g＜1；ω_cFor carrier angular frequency, T_c＝2π/ω_cFor the carrier period, T ═ NT_cFor a symbol period, τ ═ KT_cIs the modulation interval, B-A is the amplitude of carrier keying, and sigma is the phase of carrier keying, when the modulation waveform is a hard jump, sigma ∈ [0, pi]When the modulation waveform is continuous, ± ξ · Δ sin (η× 2 π f)_ct), 0 ≦ Δ ≦ 1, 0 ≦ η ≦ 1, and the value of ξ∈ { -1,1} i.e., the phase modulation polarity can be controlled by a pseudo-random sequence, r ≦ Δ ≦ 1, and_gand the integer M, N, K constitute modulation parameters that change the signal bandwidth, transmission efficiency, and demodulation performance.

Has the advantages that: compared with the prior art, the microstrip resonance coherent demodulator for the AMPSK modulation signal provided by the invention has the following advantages: 1. the applicable frequency range of the AMPSK/ABPSK modulation mode is expanded, so that the application of the AMPSK/ABPSK modulation mode in microwave, millimeter wave frequency ranges and high and medium frequency conditions becomes possible; according to the scheme, the received AMPSK/ABPSK analog signal is directly demodulated on the RF/IF, the limitation of the traditional digital impact filtering method by the ADC sampling rate is eliminated, the system structure is simplified, and the hardware cost is reduced; 2. the design method is suitable for a wide frequency range from several GHz to hundreds of GHz, the design of the demodulator can be completed by simply modifying the size parameters of the microstrip demodulation circuit aiming at different working frequencies in practice, other modules are not required to be added, and the flexibility is strong; 3. the remarkable characteristic that AMPSK/ABPSK modulation signal energy is concentrated near a carrier frequency is combined, carrier energy of a received signal is fully utilized for demodulation, a receiver does not need to additionally generate a coherent carrier, and the structure is obviously simplified; 4. the demodulation circuit is simple and cheap, has small volume, light weight and low power consumption, and is convenient for analog-digital mixed integration; 5. the negative coherent output waveform is consistent with the impulse filtering output waveform and can be directly compatible with a demodulator based on digital impulse filtering.

Drawings

Fig. 1 is a power spectrum of a signal measured at a carrier frequency of about 62.5 MHz: fig. 1(a) is EBPSK modulation, K: N is 3:1600, code rate is 53.5kbps, power bandwidth is-60 dB is 326Hz, and spectrum utilization is 164 bps/Hz; fig. 1(b) shows MPPSK modulation, where K: N is 3:1800, M is 512, code rate 428kbps, power bandwidth of-60 dB is 478Hz, and spectrum utilization rate is 895 bps/Hz;

fig. 2 shows modulation data corresponding to (a) and an impulse waveform of an EBPSK modulated signal actually measured at a carrier frequency of about 62.5MHz, a code rate of 856kbps, and a K: N: 10:100 after passing through (b) a digital impulse filter;

fig. 3 is a waveform diagram of the EBPSK modulated signal (for example, N is 20, and K is 2) demodulation process according to the present invention, where the following 3 diagrams respectively show: the waveform of the EBPSK modulation signal, the sine wave waveform which has the same frequency and is opposite to the carrier wave of the input signal and the demodulation output waveform.

Fig. 4 is a waveform diagram of a CP-EBPSK modulated signal (for example, N is 20, K is 2, and Δ is 0.5) demodulation process according to the present invention, where the upper 3 diagrams are respectively: CP-EBPSK modulation signal waveform, sine wave waveform with same frequency and opposite phase with input signal carrier, demodulation output waveform;

fig. 5 is a waveform diagram of an MPPSK modulation signal (for example, M is 11, K is 2, and N is 20) demodulation process according to the present invention, where the following 3 diagrams respectively show: MPPSK modulation signal waveform, sine wave waveform which has the same frequency and phase reversal with input signal carrier, and demodulation output waveform;

fig. 6 is a schematic diagram of instantaneous waveforms of voltage (shown by a solid line in the figure) and current (shown by a broken line in the figure) along the line when the long line terminal is open, wherein the time sequence of the waveforms of the voltage and the current is as follows: t is t₁→t₂→t₃→t₄→t₅；

FIG. 7 is a graph showing the complex amplitude distribution of voltage (shown as a solid line) and current (shown as a dashed line) along the line when the long line terminal is open;

FIG. 8 is an AMPSK modulated signal decomposition diagramThe upper 3 drawings and the lower 3 drawings are respectively as follows: AMPSK-modulated signal waveform, carrier frequency componentWaveform and transition componentA waveform;

fig. 9 is a schematic diagram of a process of superimposing an incident wave voltage and a reflected wave voltage at a position λ/4 away from an open terminal on a long line when an AMPSK modulation signal is input, where fig. 9(a) shows a waveform and a phase relationship between the incident wave and the reflected wave at the position, and fig. 9(b) is a synthesized waveform obtained by superimposing the incident wave and the reflected wave;

FIG. 10 is a schematic diagram of a demodulation circuit, in which the length of the open line of the terminal is required to be λ/4 (corresponding to the carrier frequency of the modulated signal), and the connection manner and length of the input transmission line and the output transmission line can be flexibly designed according to the actual application requirements;

fig. 11 is a schematic diagram of an embodiment of the present invention based on a microstrip circuit, in which 1 is a microwave dielectric plate, 2 is a microstrip line circuit, 3 represents a free space, and the positions indicated by arrows are the input and output terminals of the demodulation circuit, respectively;

FIG. 12(a) is a schematic diagram of the physical dimensions of a microstrip resonant demodulation circuit with a theoretical calculated operating frequency of 2.45 GHz; FIG. 12(b) is a schematic diagram of the physical dimensions of the microstrip resonant demodulation circuit obtained by rounding off the theoretical dimensions shown in FIG. 12(a) and retaining the 2 bits after the decimal point for the sake of processing accuracy;

fig. 13 is a schematic diagram of input and output waveforms of the microstrip resonant demodulation circuit with rounded parameters shown in fig. 12 (b): FIG. 13(a) is the waveform of an EBPSK modulated signal at a carrier frequency of 2.45GHz, and FIG. 13(b) is the output response of the microstrip resonant demodulation circuit to the input signal; it can be seen that a significant "bump" waveform is produced in the output response corresponding to the phase jump of the input signal;

FIG. 14 is a schematic diagram of the demodulation output waveform of the microstrip resonant demodulation circuit for EBPSK signal with carrier frequency of 45GHz according to the demodulation concept of the present invention;

FIG. 15 is a waveform of an output of a microstrip resonant demodulation circuit for demodulating an EBPSK signal having a carrier frequency of 60GHz according to the demodulation concept of the present invention;

fig. 16 is a waveform of an output of a microstrip resonant demodulation circuit for demodulating an EBPSK signal having a carrier frequency of 100GHz according to the demodulation concept of the present invention.

Detailed Description

The present invention will be further described with reference to the accompanying drawings.

The demodulator directly demodulates the analog signals received by the antenna without performing analog-to-digital conversion or down-conversion on the received analog signals; the demodulation method comprises the following steps: the demodulator directly extracts a coherent carrier which is the same frequency as and strictly opposite to a received signal carrier from an analog signal received by the antenna, and then superposes the coherent carrier with the received AMPSK modulation signal, so that the AMPSK modulation signal carrier is inhibited, the phase modulation time interval is amplified, and the signal demodulation is realized.

wherein s is_k(t) represents a modulation waveform of symbol "k", k being 0,1, …, M-1; r is_gIs a symbol guard interval control factor, r is more than or equal to 0_g＜1；ω_cFor carrier angular frequency, T_c＝2π/ω_cIs a carrier wave periodPeriod, T ═ NT_cFor a symbol period, τ ═ KT_cIs the modulation interval, B-A is the amplitude of carrier keying, and sigma is the phase of carrier keying, when the modulation waveform is a hard jump, sigma ∈ [0, pi]When the modulation waveform is continuous, ± ξ · Δ sin (η× 2 π f)_ct), 0 ≦ Δ ≦ 1, 0 ≦ η ≦ 1, and the value of ξ∈ { -1,1} i.e., the phase modulation polarity can be controlled by a pseudo-random sequence, r ≦ Δ ≦ 1, and_gand the integer M, N, K constitute modulation parameters that change the signal bandwidth, transmission efficiency, and demodulation performance.

Now, the idea of a novel demodulation scheme for AMPSK/ABPSK modulated signals is explained by taking EBPSK modulated signals as an example. The obvious characteristic that the code element of the EBPSK modulation signal ' 0 ' defined by the combination formula (4) is a single sine wave, the code element of ' 1 ' is a sine wave containing a plurality of reverse phases, the EBPSK modulation signal is added with a sine wave (hereinafter referred to as ' negative coherent signal ') which has the same frequency with the carrier wave of the EBPSK signal but is strictly reverse in phase at a receiving end, and as a result, the carrier wave of the EBPSK modulation signal is cancelled out in a reverse phase manner, and the signal amplitude at the position of the phase modulation of the code element ' 1 ' is multiplied in the same phase, so that the difference between the code elements ' 0 ' and ' 1. This is in contrast to conventional coherent demodulation schemes that use multiplication of a co-frequency in-phase carrier with the input signal followed by low-pass filtering. Since the various modulation schemes of equations (1) to (4) differ only in phase jump position and phase modulation scheme, the scheme is applicable to various AMPSK/ABPSK modulation schemes. Fig. 3 to fig. 5 respectively show the processing effect of the scheme on the EBPSK, CP-EBPSK and MPPSK modulated signals.

In order to ensure stable and reliable demodulation performance and simplify the structure of the receiver as much as possible, the scheme fully utilizes the carrier energy of the received AMPSK/ABPSK modulation signal to directly obtain the negative coherent signal without regeneration at a receiving end. This is both the focus of consideration in the design of a particular demodulation circuit and the difference between the present invention and conventional coherent demodulation methods.

Based on the negative coherent demodulation idea, in the embodiment, the AMPSK/ABPSK demodulator with the carrier frequency of 2.45GHz is designed by using a microstrip circuit, and a specific microstrip circuit structure and the demodulation effect of the microstrip circuit structure on the AMPSK/ABPSK modulated signals are given.

1. Principle of design

In the long line theory, the reflection coefficient (z) at a certain point on the long line is a parameter describing the relative amplitude and phase relationship between the reflected wave and the incident wave at that point, and is a function of the position z. The voltage reflection coefficient (z) at the position along the line z is defined as the complex vector of the voltage of the reflected wave at the positionComplex vector with incident wave voltageIf the origin of the abscissa z is selected as the long line terminal, the complex vectors of the incident wave and the reflected wave voltage at the terminal are respectively set asThe voltage reflection coefficient (z) at a distance z from the terminal may be represented

WhereinRepresenting the amplitudes of the incident wave and the reflected wave voltage at the terminal respectively, β is a phase shift constant representing the phase lag per unit distance, the current reflection coefficient is equal to the voltage reflection coefficient in a mode, and the phase difference is pi. the terminal voltage reflection coefficient can be obtained by taking z as 0 according to the formula (6)_tComprises the following steps:

for the purpose of illustrating the principles of the present invention, several relationships known in the art are set forth below without undue experimentation.From the long line theory, the input impedance Z at a distance Z from the terminal_in(z) can be expressed as:

wherein Z is₀Is the characteristic impedance of the long line, Z_LIs the load impedance of the long wire termination. Reflection coefficient (Z) at distance Z from terminal and input impedance Z_in(z) is in the relationship:

from this, the terminal reflection coefficient at the terminal, i.e., z is 0, can be obtained_tAnd a load impedance Z_LHas the relation of

When the transmission line ends short (Z)_L0) or open circuit (Z)_LInfinity), the transmission line terminal reflection coefficient can be obtained from equation (10)_t1. Considering that the present invention uses microstrip circuit as a design example, and since short circuit in microstrip circuit is more difficult to process than open circuit, the following theoretical analysis is only performed for the case of open termination (in principle, the λ/4 open termination line has the same property as the λ/2 short termination line).

When the transmission line is open-circuited at its termination, the load impedance Z_LInfinity, the terminal reflection coefficient from the formula (10)_t1 is known from formula (7)Similarly, with the terminal as the origin of the abscissa z, the complex vector expression of the synthesized wave of the voltage and the current along the line is:

taking the modulus of the formulas (11) and (12) to obtain:

then the instantaneous expressions of the voltage and current along the line are:

as can be seen from the equations (15) and (16), the phases of the voltage and the current along the line in the space domain and the time domain when the terminal is open are respectively in two independent factors, and the analysis by taking u (z, t) as an example shows that the change rule in the space domain along with the abscissa z is cos β z, and the change rule in the time t is cos β zSpecifically, when a certain point coordinate z on the long line satisfies β z ═ n (2n +1) pi/2, (2n +1) λ/4, and n ═ 0,1,2 …, the voltage at the point is always 0, and when a certain point coordinate z satisfies β z ═ n pi, z ═ n (λ/2), and n ═ 0,1,2 …, the voltage amplitude at the point is always maximumThe standing wave is always constant, i.e. formed along the line. The distribution law of the instantaneous value of the current along the line is similar to that of the voltage and changes according to a sine law along the abscissa z. The instantaneous values of the line voltage and the current when the long line terminal is open are shown in fig. 6.

As can be seen from fig. 6, the phase difference between the voltage and the current along the line when the terminal is open is pi/2 in the space domain and the time domain, and the phase difference between the pi/2 in the space domain makes the current corresponding to the maximum amplitude of the voltage along the line constantly be 0, which are called as voltage antinode and current node. The current amplitude corresponding to the position along the line where the voltage amplitude is constant at 0 is the maximum, and the positions are called as a voltage wave node and a current wave antinode. From the formulas (13) and (14), the antinodes of the line voltageAt nodeAntinode of current along the lineAt nodeThe complex amplitude distribution of the voltage and current along the line is shown in fig. 7. The phase difference of the time domain pi/2 ensures that the complex vector product of the voltage and the current along the line is always a pure imaginary number, so that only energy is stored on the line and no energy is transmitted, namely standing waves are formed on the line. The input impedance is purely reactive everywhere along the line. Will Z_LCan be obtained by substituting formula (8) with ∞

Z_in(z)＝-jZ₀cotβz (17)

It can be seen that the input impedance along the line varies with the abscissa z according to the negative cotangent law. Therefore, when the open path length z is (2n +1) λ/4, n is 0,1,2 …, β z is (2n +1) pi/2, cot β z is 0, the input impedance is 0, and the long line is equivalent to a series resonant circuit; when the open path length z is n (λ/2), n is 0,1,2 …, β z is n pi, cot β z is ± infinity, the input impedance is infinite, and the long path is equivalent to a parallel resonant circuit.

Now calculate when the terminal is open (Z)_LInfinity), voltage wave node nearest to the open end, i.e., voltage reflection coefficient at z λ/4Obtained by the formula (6):

namely:

equations (18) and (19) show that the reflected wave voltage is out of phase with the incident wave voltage by pi at a distance of lambda/4 from the terminal when the terminal is open. Therefore, the synthesized wave voltage here is a superposition of the incident wave voltage and the reflected wave voltage lagging behind the incident wave voltage by 1/2 carrier cycles. Namely:

therefore, the composite wave voltage is always 0 here for a sinusoidal signal, and becomes one node. Since the waveform of the AMPSK modulated signal is very similar to that of a sinusoidal signal, and only a short-time phase jump exists at a position corresponding to modulation information, and the energy of the AMPSK modulated signal is highly concentrated on a carrier frequency, for the AMPSK signal, at a position distant from the open-circuit terminal λ/4, the reflected wave can be regarded as an inverted coherent carrier wave by utilizing the superposition property of the reflected wave and the incident wave, and the coherent carrier wave and the input signal are superposed to play a role in suppressing the carrier wave and amplifying the phase jump, thereby completing the demodulation of the signal.

For ease of analysis, the AMPSK signal is decomposed into carrier frequency componentsAnd a hopping componentTwo parts are schematically shown in figure 8. Then, for both the incident wave and the reflected wave, the following can be decomposed:

when the input is AMPSK modulation signal, the voltage of the synthesized wave along the line isAccording to equation (20), at a distance of lambda/4 from the open circuit terminalCan be expressed as

As can be seen from equation (17), when the open line length is fixed (λ/4), the open line of fixed length forms a resonant tank only for an input signal of a specific wavelength (that is, a specific frequency). I.e. only signals of a certain frequency will form a standing wave in the open circuit line, and resonance will occur. The length of the open-circuit line is lambda if the combined type (20), (21) is set_c/4 and carrier frequency f of input AMPSK signal_cCorrespondingly, only the carrier frequency component in the input signalResonance will occur and therefore the reflected wave contains only the carrier frequency component. So is from the open terminal lambda_cThe resultant wave at/4 will be the incident wave and the carrier frequency component lagging behind the incident wave1/2 cycles of the reflected wave. Namely:

fig. 9 shows a schematic diagram of the superposition process of the incident wave and the reflected wave of the AMPSK signal represented by equation (23). It can be seen that when an AMPSK modulated signal is input, λ is at a distance open-circuit terminal_cThe voltage of the synthesized wave at the/4 position only contains jump components of the AMPSK signalThe 'impact' waveform and the position thereof contain all modulation information, and the amplitude of the jump component is 2 times of the amplitude of the input signal, which is beneficial to directly carrying out threshold judgment to realize the demodulation of the modulation information in the signal, thereby greatly simplifying the structure of the receiver.

In an actual circuit, in order to extract the 'impulse' waveform at a position which is lambda/4 away from an open circuit terminal for further processing, a transmission line with the same characteristic impedance is connected in parallel at the other end of a lambda/4 open circuit line to be used as an input line and an output line of a demodulation circuit. In fact, since the lambda/4 open circuit is equivalent to a series resonant circuit, the carrier frequency component of the input AMPSK modulated signal will resonate within the lambda/4 open circuit, forming a standing wave. While the hopping component of the AMPSK modulated signal can be transmitted over the input-output transmission line. Therefore, the λ/4 open circuit is the key of the demodulation circuit, and the specific circuit structure can be flexibly designed according to the actual requirement, and is not limited to the form given in this embodiment. Fig. 10 shows a schematic diagram of the structure of the resonant demodulation circuit after the input-output transmission line is introduced.

2. Design scheme of microstrip resonance demodulation circuit

The resonant demodulation circuit structure given according to fig. 10The design of the embodiment of the invention is based on a microstrip circuit. In the embodiment, the dielectric substrate parameter is selected as the relative dielectric constant_r22, loss tangent tan D0.0009, substrate thickness h 0.508mm, and copper layer thickness 35 μm.

In order to extract the enhanced signal with "impact" in the λ/4 open-ended microstrip line (which has the same property as the λ/2 short-ended microstrip line in principle) for further processing, and in consideration of reducing the physical size of the resonance demodulation circuit, the present embodiment adopts a λ/4 microstrip transmission line having the same characteristic impedance as the λ/4 open-ended microstrip line as the input end of the resonance demodulation circuit (using a microstrip transmission line of any length as the input end does not affect the final output effect), and simultaneously uses a λ/4 microstrip transmission line as the output end of the resonance demodulation circuit (using a microstrip transmission line of any length as the input end does not affect the final output effect), and the structure of the whole microstrip resonance demodulation circuit is shown in fig. 11.

3. Design parameter of microstrip resonance demodulation circuit

After parameters such as a dielectric substrate, a working frequency, characteristic impedance, electrical length and the like of the microstrip circuit are determined, the size parameters of the corresponding microstrip line can be rapidly calculated by using the current mainstream radio frequency simulation software (such as ADS radio frequency simulation software of Agilent). For specific operation, reference is made to the usage description of each software, and details are not repeated here. In an actual microstrip circuit, a microstrip line with an open terminal is not an ideal open circuit, an edge field effect exists at the open end of the microstrip, and the microstrip line can be equivalent to a ground capacitor or a microstrip line with a length of delta l, so that the length obtained by theoretical calculation is longer than the actual length, and therefore, the shortened length delta l representing the edge capacitance effect can be determined by an experimental method and can also be calculated by the following empirical formula:

wherein h is the dielectric substrate thickness, W is the microstrip line width, and_reis a medium baseThe effective relative dielectric constant of the sheet can be calculated using the following formula:

wherein,_ris the relative dielectric constant of the dielectric substrate. Thus, the guided wavelength λ in the medium can be calculated by the following equation, where λ₀Wavelength of electromagnetic wave in free space:

the physical size of the microstrip resonant demodulation circuit with the operating frequency of 2.45GHz obtained by calculation and correction is shown in fig. 12 (a). The physical size of the lambda/4 terminal open-circuit microstrip line is as follows: the width W is 1.520940mm, and the length L is 22.3950 mm; the input and output ends are all lambda/4 microstrip transmission lines with the same characteristic impedance as the open-circuit microstrip lines.

The physical size parameters of the microstrip demodulation circuit shown in fig. 12(a) are theoretically calculated values, which are accurate to 6 bits after decimal point, and the accuracy cannot be achieved by the actual processing technology. In this embodiment, the theoretically calculated physical size of the microstrip line is retained to 2 bits after the decimal point based on the rounding principle, so as to obtain the size parameter shown in fig. 12 (b). The physical size of the rounded lambda/4 terminal open-circuit microstrip line is as follows: the width W is 1.52mm, and the length L is 22.40 mm; the input and output ends are all lambda/4 microstrip transmission lines with the same characteristic impedance as the open-circuit microstrip lines.

4. Simulation result of microstrip resonance demodulation circuit

A microstrip resonant demodulation circuit with rounded parameters as shown in fig. 12(b) was simulated, and the output response of this circuit to an EBPSK-modulated signal with a carrier frequency of 2.45GHz as shown in fig. 13(a) is shown in fig. 13 (b). It can be seen that, since the λ/4 open-ended microstrip line forms a series resonant loop with the carrier frequency component of the 2.45GHz modulation signal, the carrier frequency component will form a standing wave on the λ/4 open-ended line, and the hopping component will continue to propagate along the transmission line, thereby generating an "impulse" waveform as shown in fig. 13(b) at the output end of the microstrip line, which is consistent with the theoretical analysis result shown in fig. 9 (b). The 'impact' waveform is subjected to threshold judgment, and the demodulation of the EBPSK signal can be realized. The simulation result also shows that the microstrip resonance demodulation circuit designed based on the negative coherent demodulation principle provided by the invention has no strict requirements on the processing technology and allows certain errors.

The demodulation performance of the microstrip circuit structure (the specific microstrip line size is recalculated) at different carrier frequencies is given below to illustrate the applicability of the negative coherent demodulation principle proposed by the present invention to a wide frequency range.

1)f_cThe physical size of the rounded lambda/4 terminal open-circuit microstrip line is as follows: the width W is 1.70mm and the length L is 1.18 mm. The output waveform is shown in fig. 14.

2)f_c60GHz, after rounding, the physical size of the lambda/4 terminal open-circuit microstrip line is as follows: the width W is 1.80mm and the length L is 0.88 mm. The output waveform is shown in fig. 15.

3)f_cThe physical size of the rounded lambda/4 terminal open-circuit microstrip line is as follows: the width W is 2.00mm and the length L is 0.52 mm. The output waveform is shown in fig. 16.

Fig. 13 to 16 show that the demodulation principle is applicable to a wide frequency range from several GHz to hundreds of GHz, and the design of the demodulation circuit can be completed only by calculating the size parameters of the microstrip line in combination with different operating frequencies in practice, which is very flexible.

The above description is only of the preferred embodiments of the present invention, and it should be noted that: it will be apparent to those skilled in the art that various modifications and adaptations can be made without departing from the principles of the invention and these are intended to be within the scope of the invention.

Claims

The demodulation method of the micro-strip resonance coherent demodulator for the AMPSK modulation signal, wherein the AMPSK modulation represents asymmetric multivariate phase shift keying modulation; the method is characterized in that: the demodulator demodulates the analog signal received by the antenna directly, and does not perform analog-to-digital conversion or down-conversion on the received analog signal; the demodulation method comprises the following steps: the demodulator directly extracts a coherent carrier which is the same frequency as and strictly opposite to a received signal carrier from an analog signal received by the antenna, and then superposes the coherent carrier with the received AMPSK modulation signal, so that the AMPSK modulation signal carrier is inhibited, the phase modulation time interval is amplified, and the signal demodulation is realized.
2. The demodulation method of the microstrip resonant coherent demodulator of AMPSK modulated signals according to claim 1, characterized in that: the AMPSK modulated signal is transmitted in a symbol period NT_cThe simplified expression within is:

$s_{k} (k) = \{\begin{matrix} \begin{matrix} A {sinω}_{c} t, & 0 \leq t \leq {NT}_{c} & k = 0 \end{matrix} \\ \{\begin{matrix} A {sinω}_{c} t, & 0 \leq t \leq (k - 1) {KT}_{c} \\ B \sin (ω_{c} t &PlusMinus; σ), & (k - 1) {KT}_{c} < t < (k - r_{g}) {KT}_{c} & 1 \leq k \leq M - 1 \\ A {sinω}_{c} t, & (k - r_{g}) {KT}_{c} \leq t < {NT}_{c} \end{matrix} \end{matrix}$

wherein s is_k(t) represents a modulation waveform of symbol "k", k being 0,1, …, M-1; r is_gIs a symbol guard interval control factor, r is more than or equal to 0_g＜1；ω_cFor carrier angular frequency, T_c＝2π/ω_cFor the carrier period, T ═ NT_cFor a symbol period, τ ═ KT_cIs the modulation interval, B-A is the amplitude of carrier keying, and sigma is the phase of carrier keying, when the modulation waveform is a hard jump, sigma ∈ [0, pi]When the modulation waveform is continuous, ± ξ · Δ sin (η× 2 π f)_ct), 0 ≦ Δ ≦ 1, 0 ≦ η ≦ 1, and the value of ξ∈ { -1,1} i.e., the phase modulation polarity can be controlled by a pseudo-random sequence, r ≦ Δ ≦ 1, and_gand the integer M, N, K constitute modulation parameters that change the signal bandwidth, transmission efficiency, and demodulation performance.