CN103856432B - Micro-strip resonance coherent demodulator for AMPSK modulating signals - Google Patents

Abstract

The invention discloses a microstrip resonant coherent demodulator for an AMPSK modulated signal. The demodulator directly extracts a coherent carrier wave with the same frequency as the carrier wave of the received signal and strictly in reverse phase from the analog signal received by the antenna, and then combines it with the receiver The received AMPSK modulation signal is superimposed, thereby suppressing the AMPSK modulation signal carrier and amplifying the AMPSK phase modulation period to realize signal demodulation without analog-to-digital conversion or down-conversion of the received signal, and the demodulator only uses a section of microstrip The coherent carrier extraction and the coherent demodulation of the AMPSK modulated signal are completed at the same time by opening the line. On the premise of maintaining the advantages of asymmetric multiple phase shift keying (AMPSK) modulated signal spectrum utilization rate and fast information transmission rate, the invention solves the problem that the traditional demodulation method based on digital shock filter is limited by analog-to-digital conversion and The processing speed is difficult to be competent for the demodulation of the microwave frequency band modulation signal.

Description

Microstrip Resonant Coherent Demodulator for AMPSK Modulated Signals

technical field

The invention relates to a microstrip resonant coherent demodulator for AMPSK modulated signals, which is a demodulator for directly coherently demodulating microwave frequency band AMPSK modulated signals by using a microstrip resonator, and belongs to the signal reception demodulation technology in digital communication.

Background technique

The rapidly growing demand for broadband wireless services puts forward higher and higher requirements for wireless communication, which directly leads to more and more crowded radio frequencies in the air, especially with the third generation (3G) and fourth generation (4G) broadband mobile With the development of communication networks, the contiguous spectrum of lower frequency bands is almost exhausted. The auction price for the 20-year use right of 10MHz spectrum in Europe has reached 4 billion euros. In my country, it is difficult to buy the frequency and bandwidth of the "golden frequency band" below 1GHz. Therefore, like energy and water resources, spectrum is also an important strategic resource of the country. Compressing wireless transmission spectrum to the maximum has important practical significance and direct economic benefits. Spectrum utilization has become the core competitive index and Key common technologies.

1. Asymmetric binary phase shift keying modulation

In a digital communication system, the process of moving the baseband signal representing binary data up to a given transmission frequency band is called modulation, and the opposite process is called demodulation. In order to improve spectrum utilization, a series of binary phase shift keying modulation methods with asymmetric modulation periods of data "0" and "1" have appeared, such as:

① The unified asymmetric binary phase shift keying (ABPSK: Asymmetric Binary PhaseShift Keying) modulation;

② In the Chinese Patent No. "ZL200910033322.5" and the title of the invention "Spectrum Compressed Extended Binary Phase Shift Keying Modulation and Demodulation Method", the disclosed continuous phase extended binary phase shift keying (CP-EBPSK: ContinuousPhase-Extended Binary Phase Shift Keying) modulation, and its various variants.

In the Chinese patent application number "201210243474.X" and the invention title "Digital Filter Bank for Demodulating Multiple ABPSK Signals", the above two modulations are uniformly expressed as:

s ₀ (t)=Asinω _c t, 0≤t<T

Among them, s ₀ (t) and s ₁ (t) respectively represent the modulation waveform of symbol "0" and "1"; ω _c is the carrier angular frequency, T _c =2π/ω _c is the carrier period, T=NT _c is the symbol period, τ=KT _c is the modulation interval; BA is the amplitude of carrier keying, σ is the phase of carrier keying: when the modulation waveform is a hard transition, σ∈[0,π]; and when the modulation waveform When continuous, σ=±ξ·Δsin(η×2πf _c t), 0≤Δ≤1, 0≤η≤1, and the value of ξ∈{-1,1}, that is, the phase modulation polarity can be a pseudo-random sequence to control.

2. Asymmetric multiple phase shift keying modulation

If the position of the modulation interval τ in the symbol period T in the formula (1) is controlled by the multi-element information symbol keying, a series of asymmetric multi-element phase shift keying (AMPSK: Asymmetric M-ary Phase Shift Keying) modulation can be obtained, Its expression is as follows:

Among them, s _k (t) represents the modulation waveform of symbol "k", k=0,1,...,M-1; r _g is the symbol guard interval control factor, 0≤r _g <1; the definition of other parameters Same as formula (1). The "modulation parameters" that change signal bandwidth, transmission efficiency, and demodulation performance are formed by r _g and integers M, N, and K.

According to the patent content of the Chinese patent number "ZL200710025202.1" and the invention name "Multiple Position Phase Shift Keying Modulation and Demodulation Method", taking the phase modulation angle σ=π and A=B=1, an optimal Commonly used multiple position phase shift keying (MPPSK: M-ary Phase Position Shift Keying), its expression is as follows:

In particular, when M=2 and r _g =0, the MPPSK modulation degenerates into a common extended binary phase shift keying (EBPSK: Extended BPSK) modulation, and its expression is as follows:

3. AMPSK/ABPSK modulated signal demodulation

The power spectrum of the AMPSK/ABPSK modulated signal shows the distinctive characteristics of high carrier and low sideband, which can get a high spectrum utilization rate, as shown in Figure 1. But on the other hand, the waveform difference between the data "0" and "non-zero" of these modulated signals is very small, which brings great challenges to demodulation.

In the Chinese patent number "ZL200910029875.3" and the title of the invention "Impact Filtering Method for Enhancing Asymmetric Binary Modulation Signals", a narrow-band digital band-pass filter based on a special design of infinite impulse response (IIR) is proposed The filter is composed of a single zero point and a multi-pole point, and can present a very narrow notch-frequency-selective characteristic at its center frequency, so that the filtered output waveform of the input modulation signal produces obvious and strong spurious amplitude modulation shocks at the information modulation, such as As shown in Figure 2, it highlights the difference between "0" and "non-zero" symbols, which facilitates the demodulation judgment of the modulated signal, and is called a digital impact filter.

It can be seen from formulas (1)~(4), for the same symbol period T=NT _c =2πN/ω _c , the transmission code rate of ABPSK modulation is:

R _bB =f _c /N (5a)

The transmission code rate of AMPSK modulation is increased to:

R _bM = (f _c /N) log ₂ M = (log ₂ M) R _bB (5b)

Therefore, the higher the carrier frequency f _c of the signal, the higher the code rate of the AMPSK/ABPSK modulated signal, so this system is more suitable for working at a higher frequency, such as radio frequency (RF) or intermediate frequency (IF).

However, in order to digitally process the communication signal, an analog-to-digital converter (ADC) must first be used to convert the received analog signal into a digital signal, but the ADC is usually large in size, high in power consumption, and expensive, and when the signal carrier frequency increases After reaching the microwave and millimeter wave frequency bands, ordinary ADC devices cannot directly sample radio frequency signals, and the advantages of the digital impact filter cannot be brought into play. Therefore, in order to take advantage of the huge advantages of AMPSK/ABPSK modulation in higher frequency bands, it is necessary to explore a new demodulation scheme suitable for microwave and millimeter wave frequency band receivers.

Contents of the invention

Purpose of the invention: In order to overcome the deficiencies in the prior art, the present invention provides a microstrip resonant coherent demodulator for AMPSK modulated signals, which is a new demodulation scheme suitable for microwave and millimeter wave frequency band receivers, and can be used in Higher frequency bands take advantage of AMPSK/ABPSK modulation.

Technical scheme: in order to achieve the above object, the technical scheme adopted in the present invention is:

The microstrip resonant coherent demodulator of AMPSK modulation signal, demodulator directly demodulates the analog signal that antenna receives, does not carry out analog-to-digital conversion or down-conversion to the received analog signal; Described demodulation method is: demodulation The device directly extracts the coherent carrier with the same frequency as the received signal carrier and is strictly opposite to the received signal carrier from the analog signal received by the antenna, and then superimposes it with the received AMPSK modulated signal, thereby suppressing the AMPSK modulated signal carrier and amplifying the phase modulation period to achieve Signal demodulation.

The above method can simultaneously complete the extraction of coherent carrier and the coherent demodulation of AMPSK modulated signal only relying on a section of microstrip open line.

The simplified expression of the AMPSK modulated signal in a symbol period NT _c is:

Among them, s _k (t) represents the modulation waveform of symbol "k", k=0,1,...,M-1; r _g is the symbol guard interval control factor, 0≤r _g <1; ω _c is the carrier Angular frequency, T _c = 2π/ω _c is the carrier period, T = NT _c is the symbol period, τ = KT _c is the modulation interval; BA is the amplitude of carrier keying, σ is the phase of carrier keying: when the modulation waveform When it is a hard transition, σ∈[0,π]; when the modulation waveform is continuous, σ=±ξ·Δsin(η×2πf _c t), 0≤Δ≤1, 0≤η≤1, and ξ∈{ The value of -1,1}, that is, the polarity of phase modulation, can be controlled by a pseudo-random sequence; r _g and integers M, N, and K constitute modulation parameters that change signal bandwidth, transmission efficiency, and demodulation performance.

Beneficial effects: the microstrip resonant coherent demodulator of the AMPSK modulation signal provided by the present invention has the following advantages compared with the prior art: 1. Expand the applicable frequency band of the AMPSK/ABPSK modulation mode, so that it can be used in the microwave and millimeter wave frequency bands And the application in the case of high and medium frequency becomes possible; this scheme directly demodulates the received AMPSK/ABPSK analog signal on the RF/IF, and gets rid of the limitation of the ADC sampling rate by the traditional digital shock filter method, simplifies the system structure and reduces the Hardware cost; 2. It is suitable for a wide frequency range from a few GHz to hundreds of GHz. In practice, the design of the demodulator can be completed by simply modifying the size parameters of the microstrip demodulation circuit for different operating frequencies, without adding other Module, strong flexibility; 3. Combined with the remarkable feature that the energy of AMPSK/ABPSK modulated signal is concentrated near the carrier frequency, the carrier energy of the received signal is fully utilized for demodulation, and the receiver does not need to generate additional coherent carrier waves, and the structure is significantly simplified; 4. Solution The modulation circuit is simple and cheap, small in size, light in weight, and low in power consumption, which is convenient for analog-digital hybrid integration; 5. The negative coherent output waveform is consistent with the output waveform of the shock filter, and can be directly compatible with the demodulator based on the digital shock filter.

Description of drawings

Figure 1 is the measured signal power spectrum on a carrier frequency of about 62.5MHz: Figure 1(a) is EBPSK modulation, K:N=3:1600, code rate 53.5kbps, -60dB power bandwidth 326Hz, spectrum utilization rate 164bps/Hz ; Fig. 1 (b) is MPPSK modulation, K:N=3:1800, M=512, code rate 428kbps,-60dB power bandwidth 478Hz, spectral efficiency 895bps/Hz;

Fig. 2 is the shock waveform after the measured about 62.5MHz carrier frequency, 856kbps code rate, K:N=10:100 EBPSK modulation signal passes through (b) digital shock filter and the modulation data corresponding to (a);

Fig. 3 is the waveform schematic diagram of the demodulation process of the EBPSK modulation signal (taking N=20, K=2 situation as an example) provided by the present invention, and 3 figures from top to bottom are respectively: EBPSK modulation signal waveform, and input signal The sine wave waveform with the same carrier frequency and opposite phase, and the demodulated output waveform.

Fig. 4 is the waveform diagram of the demodulation process of the CP-EBPSK modulation signal (with N=20, K=2, Δ=0.5 situation being example) provided by the present invention, 3 pieces of figures from top to bottom are respectively: CP-EBPSK Modulation signal waveform, sine wave waveform with the same frequency and opposite phase as the input signal carrier, and demodulated output waveform;

Fig. 5 is the waveform schematic diagram of the demodulation process of the MPPSK modulation signal (with M=11, K=2, N=20 situation being example) provided by the present invention, 3 pieces of figures from top to bottom are respectively: MPPSK modulation signal waveform, The sine wave waveform with the same frequency and opposite phase as the input signal carrier, and the demodulated output waveform;

Figure 6 is a schematic diagram of the instantaneous waveforms of the voltage (indicated by the solid line in the figure) and current (indicated by the dotted line in the figure) along the line when the terminal of the long line is open. The time sequence of the voltage and current waveforms shown in the figure is: t ₁ → t ₂ →t ₃ →t ₄ →t ₅ ;

Fig. 7 is a schematic diagram of the complex amplitude distribution of the voltage (shown by the solid line in the figure) and current (shown by the dotted line in the figure) along the line when the long-term terminal is open;

Figure 8 is a schematic diagram of AMPSK modulation signal decomposition. The three figures from top to bottom are: AMPSK modulation signal waveform, carrier frequency component Waveform and Transition Components waveform;

Figure 9 is a schematic diagram of the superimposition process of the incident wave voltage and the reflected wave voltage at the open circuit terminal λ/4 on the long line when the AMPSK modulation signal is input. Figure 9(a) shows the waveform and phase relationship between the incident wave and the reflected wave at this place. Figure 9(b) is the composite waveform after the incident wave and the reflected wave are superimposed;

Figure 10 is a schematic diagram of the structure of the demodulation circuit, in which the length of the terminal open line is required to be λ/4 (corresponding to the carrier frequency of the modulation signal), and the connection mode and length of the input transmission line and output transmission line can be flexibly designed according to the needs of actual applications ;

Figure 11 is a schematic diagram of an embodiment based on a microstrip circuit provided by the present invention, wherein 1 is a microwave dielectric board, 2 is a microstrip line circuit, 3 represents free space, and the arrows indicate the input and output terminals of the demodulation circuit respectively;

Figure 12(a) is a schematic diagram of the physical dimensions of a microstrip resonant demodulation circuit with a working frequency of 2.45 GHz obtained through theoretical calculations; Figure 12(b) is based on the theoretical dimensions shown in Figure 12(a) for the sake of processing accuracy Schematic diagram of the physical size of the microstrip resonant demodulation circuit obtained by rounding up and retaining 2 decimal places;

Figure 13 is a schematic diagram of the input and output waveforms of the microstrip resonant demodulation circuit after the rounding of the parameters given in Figure 12(b): Figure 13(a) is the EBPSK modulation signal waveform with a carrier frequency of 2.45GHz, and Figure 13(b ) is the output response of the microstrip resonant demodulation circuit to the input signal; it can be seen that an obvious "shock" waveform is produced in the output response corresponding to the phase jump of the input signal;

Fig. 14 is the demodulation output waveform of the EBPSK signal whose carrier frequency is 45GHz according to the microstrip resonant demodulation circuit designed according to the demodulation train of thought of the present invention;

Fig. 15 is the demodulation output waveform of the EBPSK signal whose carrier frequency is 60 GHz to the microstrip resonant demodulation circuit designed according to the demodulation train of thought of the present invention;

Fig. 16 is a demodulation output waveform of an EBPSK signal with a carrier frequency of 100 GHz by a microstrip resonant demodulation circuit designed according to the demodulation idea of the present invention.

detailed description

The present invention will be further described below in conjunction with the accompanying drawings.

The microstrip resonant coherent demodulator of AMPSK modulation signal, demodulator directly demodulates the analog signal that antenna receives, need not carry out analog-to-digital conversion or down conversion to the analog signal that receives; The method for described demodulation is: The demodulator directly extracts the coherent carrier with the same frequency as the received signal carrier and is strictly opposite to the received signal carrier from the analog signal received by the antenna, and then superimposes it with the received AMPSK modulated signal, thereby suppressing the AMPSK modulated signal carrier and amplifying the phase modulation period, Realize the demodulation of the signal.

The above method can simultaneously complete the extraction of coherent carrier and the coherent demodulation of AMPSK modulated signal only relying on a section of microstrip open line.

The simplified expression of the AMPSK modulated signal in a symbol period NT _c is:

Among them, s _k (t) represents the modulation waveform of symbol "k", k=0,1,...,M-1; r _g is the symbol guard interval control factor, 0≤r _g <1; ω _c is the carrier Angular frequency, T _c = 2π/ω _c is the carrier period, T = NT _c is the symbol period, τ = KT _c is the modulation interval; BA is the amplitude of carrier keying, σ is the phase of carrier keying: when the modulation waveform When it is a hard transition, σ∈[0,π]; when the modulation waveform is continuous, σ=±ξ·Δsin(η×2πf _c t), 0≤Δ≤1, 0≤η≤1, and ξ∈{ The value of -1,1}, that is, the polarity of phase modulation, can be controlled by a pseudo-random sequence; r _g and integers M, N, and K constitute modulation parameters that change signal bandwidth, transmission efficiency, and demodulation performance.

Now take the EBPSK modulation signal as an example to explain the idea of this new demodulation scheme for AMPSK/ABPSK modulation signals. Combined with the remarkable feature that the "0" symbol of the EBPSK modulation signal defined by formula (4) is a single sine wave, and the "1" symbol is a sine wave containing several anti-phase cycles, the EBPSK modulation signal is combined with a and EBPSK signal carrier with the same frequency but strictly anti-phase sine wave (hereinafter referred to as "negative coherent signal") is added, the result is that the carrier wave of the EBPSK modulation signal is canceled out in reverse phase, and the signal amplitude at the "1" symbol phase modulation is multiplied in phase , making the difference between "0" and "1" symbols more significant, which is beneficial to direct threshold judgment. This is different from the conventional coherent demodulation method that uses the same frequency and in-phase carrier to multiply the input signal and then low-pass filter. Since the various modulation modes of formulas (1)-(4) differ only in the phase jump position and the phase modulation mode, this scheme is applicable to various AMPSK/ABPSK modulation modes. Figures 3 to 5 respectively show the processing effects of the schemes on EBPSK, CP-EBPSK and MPPSK modulated signals.

In order to ensure stable and reliable demodulation performance and simplify the receiver structure as much as possible, this project makes full use of the carrier energy of the received AMPSK/ABPSK modulated signal to directly obtain the negative coherent signal without having to regenerate it at the receiving end. This is not only the focus of consideration in the specific demodulation circuit design, but also the difference between the present invention and the traditional coherent demodulation method.

Based on the negative coherent demodulation idea below, in the embodiment, the carrier frequency is 2.45 GHz by using the microstrip circuit to design the AMPSK/ABPSK demodulator, and provide the specific microstrip circuit structure and its effect on the AMPSK/ABPSK modulated signal demodulation effect.

1. Design principle

In the long-line theory, the reflection coefficient Γ(z) somewhere on the long-line is a parameter describing the relative amplitude and phase relationship between the reflected wave and the incident wave, and is a function of the position z. Now the analysis is carried out on a uniform and lossless long line, and the voltage reflection coefficient Γ(z) along the line z is defined as the reflected wave voltage complex vector and the incident wave voltage complex vector If the origin of the abscissa z is selected at the terminal of the long line, and the complex vectors of the incident wave and reflected wave voltages at the terminal are respectively Then the voltage reflection coefficient Γ(z) at the distance from the terminal z can be expressed as

in Represent the amplitude of the incident wave and reflected wave voltage at the terminal respectively, β is the phase shift constant, indicating the phase lag per unit distance; the current reflection coefficient and the voltage reflection coefficient have the same modulus, and the phase difference is π. From formula (6), taking z=0, the terminal voltage reflection coefficient _Γt can be obtained as:

In order to illustrate the principle of the present invention, some relational expressions known in the art are given below without proof. According to the long-line theory, the input impedance Z _in (z) at the distance from the terminal z can be expressed as:

Among them, Z ₀ is the characteristic impedance of the long line, and Z _L is the load impedance of the long line terminal. The relationship between the reflection coefficient Γ(z) at the distance terminal z and the input impedance Z _in (z) is:

Thus, the relationship between the terminal reflection coefficient Γ _t at the terminal, that is, z=0, and the load impedance Z _L can be obtained as

When the transmission line terminal is short circuited (Z _L =0) or open circuited (Z _L =∞), the modulus of the reflection coefficient of the transmission line terminal |Γ _t |=1 can be obtained from formula (10). In view of the fact that the present invention uses a microstrip circuit as a design example, and because a short circuit in a microstrip circuit is more difficult to process than an open circuit, the following text only focuses on the terminal open circuit (in principle, the λ/4 terminal open circuit has the same properties as the λ/2 terminal short circuit ) is analyzed theoretically.

When the terminal of the transmission line is open, the load impedance Z _L = ∞, the terminal reflection coefficient Γ _t = 1 from formula (10), and from formula (7) Similarly, with the terminal as the origin of the abscissa z, the complex vector expression of the voltage and current composite wave along the line is:

Take the modulus of formula (11), (12), get:

Then the instantaneous expressions of voltage and current along the line are:

It can be seen from equations (15) and (16) that when the terminal is open circuit, the phases of the voltage and current along the line in the space domain and the time domain are respectively in two independent factors. Taking u(z,t) as an example for analysis: in space On the domain, that is, the change law with the abscissa z is cosβz, and the change law with time t is Therefore, the passage of time does not affect the distribution of u(z, t) along the line. Specifically, when the coordinate z of a certain point on the long line satisfies βz=(2n+1)π/2, z=(2n+1)λ/4, n=0,1,2..., then the total voltage of this point is 0; and when the coordinate z of a certain point satisfies βz=nπ, z=n(λ/2), n=0,1,2..., the voltage amplitude of this point is always the largest. That is to say, the positions of the maximum and minimum voltage amplitudes on the transmission line are always the same, that is, standing waves are formed along the line. The distribution law of the instantaneous current value along the line is similar to that of the voltage, and changes according to the sinusoidal law with the abscissa z. Figure 6 shows the instantaneous values of voltage and current along the line when the terminal of the long line is open.

It can be seen from Figure 6 that when the terminal is open, the phase difference between the voltage and current along the line is π/2 in the space domain and the time domain, and the phase difference of π/2 in the space domain makes the current corresponding to the maximum voltage amplitude along the line constant to 0. These positions are called voltage antinodes and current wave nodes. The corresponding current amplitude is the largest along the line where the voltage amplitude is constant at 0, and these positions are called voltage wave nodes and current wave antinodes. From equations (13) and (14), it can be obtained that at the antinode of the voltage along the line Node antinode of current along the line Node The complex amplitude distribution of voltage and current along the line is shown in Figure 7. The phase difference of π/2 in the time domain makes the complex vector product of the voltage and current along the line always a pure imaginary number, so there is only energy storage but no energy transmission along the line, that is, a standing wave is formed along the line. Then the input impedance everywhere along the line is pure reactance. Substituting Z _L = ∞ into formula (8) can get

Z _in (z)＝-jZ ₀ cotβz (17)

It can be seen that the input impedance along the line changes with the abscissa z according to the law of negative cotangent. Therefore, when the open line length z=(2n+1)λ/4, n=0,1,2..., βz=(2n+1)π/2, cotβz=0, the input impedance is 0, and the long line is equivalent It is a series resonant circuit; when taking the open line length z=n(λ/2), n=0,1,2..., βz=nπ, cotβz=±∞, the input impedance is infinite, and the long line is equivalent to a parallel connection resonant circuit.

Now calculate the voltage wave node closest to the open circuit terminal when the terminal is open (Z _L =∞), that is, the voltage reflection coefficient at z=λ/4 From formula (6):

which is:

Equations (18) and (19) show that when the terminal is open, at a distance of λ/4 from the terminal, the phase difference between the reflected wave voltage and the incident wave voltage is π. Therefore, the composite wave voltage here is the superposition of the incident wave voltage and the reflected wave voltage lagging behind the incident wave voltage by 1/2 carrier period. which is:

Therefore, for a sinusoidal signal, the synthesized wave voltage here is always 0, which becomes a wave node. Since the waveform of the AMPSK modulation signal is very "similar" to the sinusoidal signal, there is only a short-term phase jump at the corresponding position of the modulation information, and its energy is highly concentrated in the carrier frequency. Therefore, for the AMPSK signal, at a distance of λ/4 from the open terminal At the position, the superposition property of the reflected wave and the incident wave can be used to treat the reflected wave as an anti-phase coherent carrier. By superimposing the coherent carrier and the input signal, the carrier can be suppressed and the phase jump can be amplified. Signal demodulation.

For ease of analysis, the AMPSK signal is decomposed into carrier frequency components and jump components The two parts are schematically shown in Figure 8. Then the incident wave and reflected wave can be decomposed as follows:

When the input is AMPSK modulation signal, the synthetic wave voltage along the line is According to formula (20), at a distance of λ/4 from the open circuit terminal can be expressed as

It can be seen from formula (17) that when the length of the open line is fixed (λ/4), the open line with a fixed length only constitutes a resonant circuit for an input signal of a specific wavelength (that is, a specific frequency). That is, only a signal of a specific frequency will form a standing wave in the open line and resonate. Combining formulas (20) and (21), if the length of the open line is set to λ _c /4 corresponding to the carrier frequency f _c of the input AMPSK signal, then there is only the carrier frequency component in the input signal Resonance will occur so that only the carrier frequency component is contained in the reflected wave. Therefore, the synthesized wave at λ _c /4 distance from the open terminal will be the incident wave and the carrier frequency component lagging behind the incident wave Superposition of reflected waves of 1/2 period. which is:

FIG. 9 shows a schematic diagram of the superposition process of the incident wave and the reflected wave of the AMPSK signal represented by formula (23). It can be seen that when an AMPSK modulation signal is input, only the jump component of the AMPSK signal is contained in the synthetic wave voltage at the distance from the open circuit terminal λ _c /4 The "shock" waveform and its position in the circuit contain all the modulation information, and the amplitude of the jump component is twice the amplitude of the input signal, which is conducive to directly performing threshold judgment to realize the demodulation of the modulation information in the signal, thus greatly improving the Simplifies the receiver structure.

In the actual circuit, in order to extract the "impact" waveform at the distance of λ/4 from the open circuit terminal for further processing, it is necessary to connect a transmission line with the same characteristic impedance at the other end of the λ/4 open circuit as the input and output of the demodulation circuit Wire. In fact, since the λ/4 open line is equivalent to a series resonant circuit, the carrier frequency component of the input AMPSK modulation signal will resonate in the λ/4 open line to form a standing wave. The jump component in the AMPSK modulation signal can be transmitted through the input and output transmission lines. Therefore, the λ/4 open line is the key to the demodulation circuit, and the specific circuit structure can be flexibly designed according to actual needs, and is not limited to the form given in this embodiment. FIG. 10 shows a schematic structural diagram of the resonant demodulation circuit after the input and output transmission lines are introduced.

2. Design scheme of microstrip resonant demodulation circuit

According to the structural diagram of the resonant demodulation circuit shown in FIG. 10 , the design of the embodiment of the present invention is based on the microstrip circuit. The parameters of the dielectric substrate selected in the embodiment are relative permittivity ε _r =22, loss tangent tanD=0.0009, substrate thickness h=0.508 mm, and copper clad layer thickness is 35 μm.

In order to further process the enhanced signal with "impact" in the λ/4 open-circuit microstrip line (in principle, it has the same properties as the λ/2-terminal short-circuit microstrip line), and to reduce the resonance demodulation circuit Considering the physical size, this embodiment adopts the λ/4 microstrip transmission line having the same characteristic impedance as the λ/4 terminal open circuit microstrip line as the input end of the resonant demodulation circuit (using the microstrip transmission line of any length as the input end does not affect the final output effect), and use the λ/4 microstrip transmission line as the output end of the resonant demodulation circuit (using the microstrip transmission line of any length as the input end does not affect the final output effect), the structure of the entire microstrip resonant demodulation circuit is shown in Figure 11 shown.

3. Design parameters of microstrip resonant demodulation circuit

After determining the parameters of the dielectric substrate, operating frequency, characteristic impedance, and electrical length of the microstrip circuit, the size of the corresponding microstrip line can be quickly calculated using the current mainstream RF simulation software (such as ADS RF simulation software from Agilent). parameter. For specific operations, refer to the instructions of each software, and will not repeat them here. In the actual microstrip circuit, the open-circuited microstrip line is not an ideal open circuit, and there is a fringe field effect at the open-circuit end of the microstrip, which can be equivalent to a grounded capacitor or a microstrip line with a length of Δl, so the theoretical calculation is The length of is longer than the actual length, therefore, the shortened length Δl representing the fringe capacitance effect can be determined experimentally, or can be calculated using the following empirical formula:

Among them, h is the thickness of the dielectric substrate, W is the width of the microstrip line, and _εre is the effective relative permittivity of the dielectric substrate, which can be calculated by the following formula:

Among them, ε _r is the relative permittivity of the dielectric substrate. Therefore, the guided wavelength λ in the medium can be calculated by the following formula, where _λ0 is the wavelength of electromagnetic waves in free space:

After calculation and correction, the physical size of the microstrip resonant demodulation circuit with an operating frequency of 2.45GHz is shown in Figure 12(a). The physical dimensions of the λ/4-terminated open-circuit microstrip line are: width W=1.520940mm, length L=22.3950mm; both input and output ends use a λ/4 microstrip transmission line with the same characteristic impedance as the open-circuit microstrip line.

The physical size parameters of the microstrip demodulation circuit shown in Figure 12(a) are theoretically calculated values, accurate to 6 decimal places, which cannot be achieved by actual processing technology. During implementation, the theoretically calculated physical dimensions of the microstrip line should be appropriately rounded according to the processing technology level. This embodiment is based on the rounding principle, and the theoretically calculated physical dimensions are kept to 2 decimal places, and the result shown in Figure 12(b) Dimensional parameters shown. After rounding, the physical dimensions of the λ/4-terminated open-circuit microstrip line are: width W = 1.52mm, length L = 22.40mm; both input and output ends use a λ/4 microstrip transmission line with the same characteristic impedance as the open-circuit microstrip line.

4. Simulation results of microstrip resonant demodulation circuit

Simulation is carried out on the microstrip resonant demodulation circuit whose parameters are rounded as shown in Figure 12(b). The output response of this circuit to the EBPSK modulation signal with a carrier frequency of 2.45GHz as shown in Figure 13(a) is shown in the figure 13(b). It can be seen that since the λ/4-terminated open circuit microstrip line forms a series resonant circuit for the carrier frequency component of the 2.45GHz modulation signal, the carrier frequency component will form a standing wave on the λ/4 open circuit, while the jump component will continue along the transmission line Propagate, so that the "impact" waveform shown in Figure 13(b) is generated at the output of the microstrip line, which is consistent with the theoretical analysis results shown in Figure 9(b). The demodulation of the EBPSK signal can be realized by making a threshold judgment on the "impact" waveform. The simulation results also show that the microstrip resonant demodulation circuit designed based on the principle of negative coherence demodulation proposed by the present invention does not have strict requirements on the processing technology, and certain errors are allowed.

The demodulation performance of the above-mentioned microstrip circuit structure (the size of the microstrip line has been recalculated) under different carrier frequencies is given below to illustrate the applicability of the inverse coherent demodulation principle proposed by the present invention to a wide frequency range.

1) f _c = 45 GHz, the physical dimensions of the λ/4-terminated open circuit microstrip line after rounding are: width W = 1.70 mm, length L = 1.18 mm. The output waveform is shown in Figure 14.

2) f _c =60 GHz, the physical dimensions of the λ/4 open-circuit microstrip line after rounding are: width W=1.80 mm, length L=0.88 mm. The output waveform is shown in Figure 15.

3) f _c =100 GHz, the physical dimensions of the λ/4-terminated open-circuit microstrip line after rounding are: width W=2.00 mm, length L=0.52 mm. The output waveform is shown in Figure 16.

Figures 13 to 16 show that the demodulation principle is applicable to a wide frequency range from a few GHz to hundreds of GHz. In practice, it is only necessary to calculate the size parameters of the microstrip line in combination with different operating frequencies to complete the design of the demodulation circuit. flexible.

The above is only a preferred embodiment of the present invention, it should be pointed out that for those of ordinary skill in the art, without departing from the principle of the present invention, some improvements and modifications can also be made, and these improvements and modifications are also possible. It should be regarded as the protection scope of the present invention.

Claims (2)

1. The demodulation method of the microstrip resonant coherent demodulator of AMPSK modulation signal, AMPSK modulation represents asymmetric multivariate phase-shift keying modulation; It is characterized in that: demodulator directly demodulates the analog signal that antenna receives, The analog-to-digital conversion or down-conversion is not performed on the received analog signal; the demodulation method is: after the demodulator directly extracts the coherent carrier wave with the same frequency as the received signal carrier and strictly opposite to the received signal carrier from the analog signal received by the antenna, and then Superimposed with the received AMPSK modulation signal, thereby suppressing the carrier of the AMPSK modulation signal and amplifying the phase modulation period to realize demodulation of the signal. 2. the demodulation method of the microstrip resonant coherent demodulator of AMPSK modulation signal according to claim 1, it is characterized in that: described AMPSK modulation signal, the simplified expression in a symbol period NT _c is: ${\mathrm{<span\; class="notranslate">\; the\; s</span>}}_{\mathrm{<span\; class="notranslate">\; k</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\left\{\begin{array}{c}\begin{array}{ccc}\mathrm{<span\; class="notranslate">\; A</span>}{\mathrm{<span\; class="notranslate">\; sin\&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; ,</span>& \mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; \&le;</span>{\mathrm{<span\; class="notranslate">\; NT</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}& \mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}\end{array}\\ \left\{\begin{array}{ccc}\mathrm{<span\; class="notranslate">\; A</span>}{\mathrm{<span\; class="notranslate">\; sin\&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; ,</span>& \mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; \&le;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; KT</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}& \\ \mathrm{<span\; class="notranslate">\; B</span>}\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; \&PlusMinus;</span>\mathrm{<span\; class="notranslate">\; \&sigma;</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>& <span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; KT</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; <</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; <</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; g</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; KT</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}& \mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; m</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}\\ \mathrm{<span\; class="notranslate">\; A</span>}{\mathrm{<span\; class="notranslate">\; sin\&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; ,</span>& <span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; r</span>}}_{\mathrm{<span\; class="notranslate">\; g</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; KT</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; <</span>{\mathrm{<span\; class="notranslate">\; NT</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}& \end{array}\right.\end{array}\right.$ Among them, s _k (t) represents the modulation waveform of symbol "k", k=0,1,...,M-1; r _g is the symbol guard interval control factor, 0≤r _g <1; ω _c is the carrier Angular frequency, T _c = 2π/ω _c is the carrier period, T = NT _c is the symbol period, τ = KT _c is the modulation interval; BA is the amplitude of carrier keying, σ is the phase of carrier keying: when the modulation waveform When it is a hard transition, σ∈[0,π]; when the modulation waveform is continuous, σ=±ξ·Δsin(η×2πf _c t), 0≤Δ≤1, 0≤η≤1, and ξ∈{ The value of -1,1}, that is, the polarity of phase modulation, can be controlled by a pseudo-random sequence; r _g and integers M, N, and K constitute modulation parameters that change signal bandwidth, transmission efficiency, and demodulation performance.