CN103746629A - Model prediction-based current control method for flux-switching permanent magnet synchronous machine - Google Patents

Abstract

The invention discloses a model prediction-based current control method for a flux-switching permanent magnet synchronous machine. The current control method comprises the following steps: performing combination on states of a switching tube of a machine inverter to obtain eight groups of switching vector signals; predicting the p phase winding current of the next moment k+1 under each group of switching vector signals at the current moment k; obtaining current predicted values of a d (direct) axis and a (quadrature) axis at the next moment k+1 under each group of switching vector signals; calculating the predictive current error of the d axis and the predictive current error of the q axis at the next moment k+1 under each group of switching vector signals; calculating a cost function under each group of switching vector signals; selecting the switching vector signals when the cost function is minimum, and controlling the on-off of the switching tube of the inverter in real time by taking the switching vector signals as drive signals of an inverter switch to complete the current control of the flux-switching permanent magnet synchronous machine. According to the method, the switching vector changes of the inverter can be accurately controlled, so that the torque fluctuation and the flux fluctuation of the machine are minimal.

Description

The current control method of the switch flux-linkage permagnetic synchronous motor based on model prediction

Technical field

The invention belongs to switch flux-linkage permagnetic synchronous motor technical field, more specifically, relate to a kind of current control method of the switch flux-linkage permagnetic synchronous motor based on model prediction.

Background technology

The energy is as one of the most popular topic in the world today, the research of environmental protection and energy saving technology and the application problem that countries in the world are all attached great importance at present especially.Industrial and social sustainable development, constantly the consumption main body motor of electric power energy has been proposed to new requirement, under prerequisite as much as possible, motor not only will have simply firmly mechanical structure, higher torque density, and will possess the advantages such as robustness is good, fault-tolerant ability is strong.

The stator yoke portion that the scholars such as the Lipo professor of U.S. Wisconsin university and Liao Yuefeng propose embeds novel doubly-salient permagnetic synchronous motor (Doubly Salient Permanent Magnet Synchronous Machine the is called for short DSPMSM) structure of permanent magnet and has opened up the frontier of doubly salient permanent magnet motor research.Not only efficiency is high for the type motor, dynamic property is good, and has stronger anti-external disturbance ability.But due to the particularity of structure, also there is the problems such as torque pulsation is large in the type motor.

Switch flux-linkage permagnetic synchronous motor (Flux-Switching Permanent Magnet Synchronous Machine is called for short FSPMSM) design philosophy research to single-phase doubly salient permanent magnet generator from people such as Rauch the earliest.Be subject to the restriction of scientific and technological level, working condition at that time, this motor volume is large, and running performance index is poor, is not therefore receiving publicity at that time.1997, the people such as Hoang turned one's attention to the FSPMSM based on the work of magnetic linkage switching mode again, and proposed 12/10(rotor/stator) utmost point three-phase FSPMSM, simultaneously preliminary emulation has been proved FSPMSM and has been had the advantages such as good mechanical property, high torque density.The various versions of FSPMSM obtain broad research all over the world subsequently.Various countries scholar, for the design principle of FSPMSM, has proposed different version, for example 6/4 utmost point, 6/5 utmost point, 8/6 utmost point, composite excitation structure etc.

Novel switched magnetic linkage permanent magnetic synchronous motor structure is also constantly being suggested, and continued to optimize and improved, thereby improve the driving force (W.Xu of motor, J.Zhu, Y.Zhang, and J.Hu, " Cogging torque reduction for radially laminated flux-switching permanent magnet machine with 12/14 poles; " in Proc.IEEE Industrial Electronics Society (IECON), Nov.2011, pp.3465-3470.).But, only from structural topology and Electromagnetic Optimum Design equal angles, be difficult to obviously improve the performance index of FSPMSM, thereby must, in conjunction with senior control strategy, further strengthen the driving force of FSPMSM.

Summary of the invention

For above defect or the Improvement requirement of prior art, the invention provides a kind of current control method of the switch flux-linkage permagnetic synchronous motor based on model prediction, can realize the accurate control to FSPMSM output current.The present invention is by the size of reference current under dq axis coordinate system relatively and predicted current, and take optimal cost function as object, accurately control inverter switching vector selector changes, and makes motor torque fluctuation and magnetic linkage fluctuation minimum.

For achieving the above object, according to one aspect of the present invention, provide a kind of current control method of switch flux-linkage permagnetic synchronous motor, it is characterized in that, comprised the steps:

(1) the switching tube state of motor inverter is combined, obtain eight groups of inverter switching device vector signals;

(2), at current time k, under each group inverter switching device vector signal, gather the p phase winding voltage of switch flux-linkage permagnetic synchronous motor respectively

p phase winding electric current

with rotor electrical angle speed omega _r, predict the p phase winding electric current of next moment k+1

wherein, p represents motor A, B and C phase;

(3) respectively under each group inverter switching device vector signal, according to the p phase winding electric current of next moment k+1

in conjunction with rotor winding principle of coordinate transformation, obtain the d axle of next moment k+1 and the current forecasting value i of q axle _d ^k+1and i _q ^k+1;

(4) at each, organize under inverter switching device vector signal respectively, calculate the d axle predicted current error of next moment k+1

with q axle predicted current error

wherein, for d axle reference current,

for q axle reference current;

(5) at each, organize under inverter switching device vector signal respectively, function assesses the cost

wherein, k ₁for the ratio of nominal torque and specified magnetic linkage, N is positive integer, and A is the regulating error coefficient of N state dq shaft current, with

be respectively the current forecasting value of d axle and the q axle in k+N moment;

(6) choose cost function G inverter switching device vector signal hour, set it as the driving signal of inverter switching device, the break-make of control inverter switching tube, completes the Current Control to switch flux-linkage permagnetic synchronous motor in real time.

Preferably, in described step (5),

with

expression formula be:

$\left\{\begin{array}{c}{i}_d^{k+N}=i_d^k+(N-1)\×{\mathrm{\Δi}}_d^{k+1}\\ i_q^{k+N}=i_q^k+(N-1)\×{\mathrm{\Δi}}_q^{k+1}\end{array}\right.,$

Wherein,

with

be respectively d axle and the q shaft current value of current time k,

for from current time k to next moment k+1, the change amount of d shaft current,

for from current time k to next moment k+1, the change amount of q shaft current.

Preferably,

with

measured current value by current time k obtains through coordinate transform.

Preferably, in described step (2), the p phase winding electric current of next moment k+1

for:

$i_p^{k+1}=i_p^k+\frac{1}{L_p}(u_p^k-R_p{i}_p^k-i_p^k\frac{{\mathrm{dL}}_p}{{\mathrm{d\θ}}_r}{\mathrm{\ω}}_r-\frac{{\mathrm{d\ψ}}_{\mathrm{pm},p}}{{\mathrm{d\θ}}_r}{\mathrm{\ω}}_r)*T_s,$

Wherein, L _pfor p phase winding self-induction, R _pfor p phase winding resistance, θ _rfor motor rotor position angle, ψ _{pm, p}for motor p phase permanent magnetism magnetic linkage, T _sfor switch periods.

Preferably, in described step (3), the current forecasting value i of d axle and q axle _d ^k+1and i _q ^k+1can be expressed as:

$\left[\begin{array}{c}{i}_d^{k+1}\\ i_q^{k+1}\end{array}\right]=\frac{2}{3}\left[\begin{array}{ccc}\cos {\mathrm{\θ}}_e& \cos ({\mathrm{\θ}}_e-2\mathrm{\π}/3)& \cos ({\mathrm{\θ}}_e+2\mathrm{\π}/3)\\ \sin {\mathrm{\θ}}_e& \sin ({\mathrm{\θ}}_e-2\mathrm{\π}/3)& \sin ({\mathrm{\θ}}_e+2\mathrm{\π}/3)\end{array}\right]\×\left[\begin{array}{c}{i}_A^{k+1}\\ i_B^{k+1}\\ i_C^{k+1}\end{array}\right],$

Wherein, θ _efor motor synchronous electric angle, i _a ^k+1, i _b ^k+1and i _c ^k+1be respectively A phase, B phase and the C phase winding electric current of next moment k+1.

Preferably, in described step (4), by motor reference rotation velocity and output speed, by pi regulator, obtained.

In general, the above technical scheme of conceiving by the present invention compared with prior art, has following beneficial effect:

1, can realize the accurate control to FSPMSM output current.Take FSPMSM equivalent mathematical model as basis, can effectively dope under eight kinds of inverter switching states, the value of the motor output current in next moment under three phase coordinate systems, and in conjunction with rotor winding principle of coordinate transformation, current forecasting value under three phase coordinate systems is carried out to coordinate transform, obtain the current forecasting value under dq axis coordinate system.

2, accurately control inverter switching vector selector changes, and makes motor torque fluctuation and magnetic linkage fluctuation minimum.Passing ratio integration (PI) adjuster comparison reference rotation velocity and real-time rotate speed obtain reference current, introduce cost function (Cost Function) as the modulation strategy that reduces motor torque fluctuation and magnetic linkage fluctuation, by the size of reference current and predicted current under comparison dq axis coordinate system, select hour corresponding switching vector selector signal of cost function, control inverter work in real time.

Accompanying drawing explanation

Fig. 1 is induction machine rotor winding coordinate transform schematic diagram;

Fig. 2 is the current control method flow chart of the switch flux-linkage permagnetic synchronous motor based on model prediction of the embodiment of the present invention;

Fig. 3 is the current control system structural representation of switch flux-linkage permagnetic synchronous motor.

Embodiment

In order to make object of the present invention, technical scheme and advantage clearer, below in conjunction with drawings and Examples, the present invention is further elaborated.Should be appreciated that specific embodiment described herein, only in order to explain the present invention, is not intended to limit the present invention.In addition,, in each execution mode of described the present invention, involved technical characterictic just can combine mutually as long as do not form each other conflict.

The basic mathematical equation of switch flux-linkage permagnetic synchronous motor is as follows:

Permanent magnetism magnetic linkage equation:

$\left\{\begin{array}{c}{\mathrm{\ψ}}_{\mathrm{pm},A}={\mathrm{\ψ}}_m\cos {\mathrm{\θ}}_e\\ {\mathrm{\ψ}}_{\mathrm{pm},B}={\mathrm{\ψ}}_m\cos ({\mathrm{\θ}}_e-2\mathrm{\π}/3)\\ {\mathrm{\ψ}}_{\mathrm{pm},C}={\mathrm{\ψ}}_m\cos ({\mathrm{\θ}}_e+2\mathrm{\π}/3)\end{array}\right.---\left(1\right)$

Inductance equation:

$\left\{\begin{array}{c}{L}_A=L_0-L_m\cos \left(2{\mathrm{\θ}}_e\right)\\ L_B=L_0-L_m\cos (2{\mathrm{\θ}}_e+2\mathrm{\π}/3)\\ L_C=L_0-L_m\cos (2{\mathrm{\θ}}_e-2\mathrm{\π}/3)\end{array}\right.---\left(2\right)$

Wherein, θ _e=P _rθ _r, P _rand θ _rrespectively the rotor number of teeth and rotor electromagnetic point, ψ _mpermanent magnetism magnetic linkage amplitude, L ₀and L _mrespectively inductance initial value and amplitude.

Mutual inductance equation:

$\left\{\begin{array}{c}{M}_{\mathrm{ab}}=M_{\mathrm{ba}}=M_0-M_m\cos (2{\mathrm{\θ}}_e-2\mathrm{\π}/3)\\ M_{\mathrm{bc}}=M_{\mathrm{cb}}=M_0-M_m\cos \left(2{\mathrm{\θ}}_e\right)\\ M_{\mathrm{ca}}=M_{\mathrm{ac}}=M_0-M_m\cos (2{\mathrm{\θ}}_e+2\mathrm{\π}/3)\end{array}\right.---\left(3\right)$

Wherein, M ₀and M _mrespectively mutual inductance initial value and amplitude.

According to Electrical Motor theory, the voltage equation that the present invention further obtains FSPMSM is:

$\left[\begin{array}{c}{u}_A\\ u_B\\ u_C\end{array}\right]=\left[\begin{array}{ccc}{R}_s& & \\ & R_s& \\ & & R_s\end{array}\right]\left[\begin{array}{c}{i}_A\\ i_B\\ i_C\end{array}\right]+\frac{d}{\mathrm{dt}}\left[\begin{array}{c}{\mathrm{\ψ}}_A\\ {\mathrm{\ψ}}_B\\ {\mathrm{\ψ}}_C\end{array}\right]---\left(4\right)$

Wherein, u _a, u _band u _cbe respectively A, B and C phase voltage, R _sfor armature winding phase resistance.

The torque equation of tradition permagnetic synchronous motor is:

$T_e=\frac{1}{2}{\left[I\right]}^T\left(\frac{\∂}{{\∂\mathrm{\θ}}_r}\left[L\right]\right)\left[I\right]+\left(\frac{\∂}{{\∂\mathrm{\θ}}_r}{\left[{\mathrm{\ψ}}_{\mathrm{pm}}\right]}^T\right)\left[I\right]---\left(5\right)$

Traditional permagnetic synchronous motor inductance parameters is fixed numbers, not with rotor position angle θ _rvariation and change.Because FSPMSM rotor is all double-salient-pole structures, permanent magnet embeds in each stator tooth, and permanent magnetism magnetic linkage, inductance and mutual inductance are all with rotor position angle θ _rchanging and change, is the function of rotor-position.Therefore formula (5) can not be directly in order to solve the electromagnetic torque of FSPMSM.

To FSPMSM, magnetic linkage, voltage and the torque equation under dq axle and ABC axle re-starts derivation and explanation in the present invention, specific as follows.

Dq axle (two-phase) coordinate system FSPMSM equivalent equation is as follows:

Magnetic linkage equation:

$\left\{\begin{array}{c}{\mathrm{\ψ}}_d={\mathrm{\ψ}}_m+L_d{i}_d\\ {\mathrm{\ψ}}_q=L_q{i}_q\end{array}\right.---\left(6\right)$

Write as matrix form:

$\left[\begin{array}{c}{\mathrm{\&psi;}}_d\\ {\mathrm{\&psi;}}_q\end{array}\right]=\left[\begin{array}{cc}{\mathrm{<figure-callout\; id="0"\; label="L\; d"\; filenames="HDA0000458156320000031.png"\; state="\{\{state\}\}">L</figure-callout>}}_d& 0\\ 0& L_q\end{array}\right]\left[\begin{array}{c}{i}_d\\ i_q\end{array}\right]+\left[\begin{array}{c}{\mathrm{\&psi;}}_m\\ 0\end{array}\right]---\left(7\right)$

Wherein, L _dand L _qbe respectively motor d axle self-induction and q axle self-induction after dq conversion, i _dand i _qbe respectively the electric current that transforms to d axle and q axle.Induction machine rotor winding principle of coordinate transformation as shown in Figure 1.

Voltage equation:

$\left\{\begin{array}{c}{u}_d={\mathrm{Ri}}_d+\frac{{\mathrm{d\&psi;}}_d}{\mathrm{dt}}-{\mathrm{\&omega;}}_r{\mathrm{\&psi;}}_q\\ u_q={\mathrm{Ri}}_q+\frac{{\mathrm{d\&psi;}}_q}{\mathrm{dt}}+{\mathrm{\&omega;}}_r{\mathrm{\&psi;}}_d\end{array}\right.---\left(8\right)$

In above formula, u _dand u _qfor transforming to the electric moter voltage of d axle and q axle, ω _rfor rotor electrical angle speed.

Torque equation:

$\begin{array}{c}{T}_e=\frac{1}{2}{\left[I\right]}^T\left(\frac{\&PartialD;}{{\&PartialD;\mathrm{\&theta;}}_r}\left[L\right]\right)\left[I\right]+\left(\frac{\&PartialD;}{{\&PartialD;\mathrm{\&theta;}}_r}{\left[{\mathrm{\&psi;}}_{\mathrm{pm}}\right]}^T\right)\left[I\right]\\ =\frac{3}{2}{P}_r{\mathrm{\&psi;}}_m{I}_q-\frac{3}{2}{P}_r({2M}_m+L_m)I_d{I}_q\\ =\frac{3}{2}{P}_r[{\mathrm{\&psi;}}_m{I}_q+(L_d-L_q)I_d{I}_q]\end{array}---\left(9\right)$

Because the relative self-induction of motor mutual inductance is smaller, the present invention ignores mutual inductance, and further magnetic linkage, voltage and torque under motor ABC axle (single-phase) coordinate system is derived, specific as follows.

ABC axle (single-phase) coordinate system FSPMSM equivalent equation:

Magnetic linkage equation:

The single-phase magnetic linkage of FSPMSM comprises armature reaction magnetic linkage and two parts of permanent magnetism magnetic linkage, is:

ψ _p=ψ _pm,p+L _pi _p (10)

Wherein, p represents motor A, B and C phase, i _pfor p phase winding electric current, L _pfor p phase winding self-induction, ψ _pfor motor p phase magnetic linkage, ψ _{pm, p}for motor p phase permanent magnetism magnetic linkage.

Voltage equation:

$\begin{array}{c}{u}_p=R_p{i}_p+\frac{d}{\mathrm{dt}}(L_p{i}_p+{\mathrm{\&psi;}}_{\mathrm{pm},p})\\ =R_p{i}_p+L_p\frac{{\mathrm{di}}_p}{\mathrm{dt}}+i_p\frac{{\mathrm{dL}}_p}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r+\frac{{\mathrm{d\&psi;}}_{\mathrm{pm},p}}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r\end{array}---\left(11\right)$

Wherein, R _pfor p phase winding resistance.

Torque equation:

FSPMSM output electromagnetic torque is comprised of reluctance torque and permanent-magnet torque two parts, and wherein reluctance torque is by changing inductance and electric current acting in conjunction produces, and permanent-magnet torque is produced by the interaction of permanent magnetism magnetic linkage and armature supply, and correlated expression formula is:

$T_{\mathrm{ep}}=\frac{1}{2}{i}_p^2\frac{{\&PartialD;L}_p}{{\&PartialD;\mathrm{\&theta;}}_r}+i_p\frac{{\&PartialD;\mathrm{\&psi;}}_{\mathrm{pm},p}}{{\&PartialD;\mathrm{\&theta;}}_r}=T_{\mathrm{rp}}+T_{\mathrm{mp}}---\left(12\right)$

In above formula, T _rpfor reluctance torque component, T _mpfor permanent-magnet torque component.

The equation of motion:

$T_e=T_L+\mathrm{F\&omega;}+J\frac{\mathrm{d\&omega;}}{\mathrm{dt}}---\left(13\right)$

In above formula, T _efor motor output electromagnetic torque, T _lfor load torque, J is system moment of inertia, and F is system friction coefficient, and ω is rotor mechanical angular speed.

According to formula (13), the total output electromagnetic torque of motor is:

T _e=T _ea+T _eb+T _ec (14)

Model Predictive Control is that a kind of being based upon set up the control algolithm on the basis of model to studied object, and the existing historical data of Main Basis and want the object reaching dopes the development trend of key message.In concrete forecasting process, different models, can adopt different prediction mode and different feedback strategies to form different predictive control algorithms.Similarly, accurate Mathematical Modeling will largely determine the accuracy of prediction.Therefore, Model Predictive Control has good adaptability to complication system.

As shown in Figure 2, the current control method of the switch flux-linkage permagnetic synchronous motor based on model prediction of the embodiment of the present invention comprises the steps:

(1) the switching tube state of motor inverter is combined, obtain eight groups of inverter switching device vector signals.

(2), at current time k, under each group inverter switching device vector signal, gather the p phase winding voltage of switch flux-linkage permagnetic synchronous motor respectively

p phase winding electric current

with rotor electrical angle speed omega _r, predict the p phase winding electric current of next moment k+1

for:

$i_p^{k+1}=i_p^k+\frac{1}{L_p}(u_p^k-R_p{i}_p^k-i_p^k\frac{{\mathrm{dL}}_p}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r-\frac{{\mathrm{d\&psi;}}_{\mathrm{pm},p}}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r)*T_s$

Wherein, L _pfor p phase winding self-induction, R _pfor p phase winding resistance, θ _rfor motor rotor position angle, ψ _{pm, p}for motor p phase permanent magnetism magnetic linkage, T _sfor switch periods.

Particularly, be Accurate Prediction FSPMSM winding current value at a time, by formula (11), obtain current change quantity as follows:

$\frac{{\mathrm{di}}_p}{\mathrm{dt}}=(u_p-R_p{i}_p-i_p\frac{{\mathrm{dL}}_p}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r-\frac{{\mathrm{d\&psi;}}_{\mathrm{pm},p}}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r)/L_p---\left(15\right)$

be the p phase winding electric current of current sampling instant k, the p phase winding electric current that is obtained next moment k+1 by formula (15) is:

$i_p^{k+1}=i_p^k+\frac{1}{L_p}(u_p^k-R_p{i}_p^k-i_p^k\frac{{\mathrm{dL}}_p}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r-\frac{{\mathrm{d\&psi;}}_{\mathrm{pm},p}}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r)*T_s---\left(16\right)$

(3) at each, organize under inverter switching device vector signal respectively, according to the p phase winding electric current of next moment k+1, in conjunction with rotor winding principle of coordinate transformation, obtain the dq shaft current predicted value of next moment k+1:

$\left[\begin{array}{c}{i}_d^{k+1}\\ i_q^{k+1}\end{array}\right]=\frac{2}{3}\left[\begin{array}{ccc}\cos {\mathrm{\&theta;}}_e& \cos ({\mathrm{\&theta;}}_e-2\mathrm{\&pi;}/3)& \cos ({\mathrm{\&theta;}}_e+2\mathrm{\&pi;}/3)\\ \sin {\mathrm{\&theta;}}_e& \sin ({\mathrm{\&theta;}}_e-2\mathrm{\&pi;}/3)& \sin ({\mathrm{\&theta;}}_e+2\mathrm{\&pi;}/3)\end{array}\right]\&times;\left[\begin{array}{c}{i}_A^{k+1}\\ i_B^{k+1}\\ i_C^{k+1}\end{array}\right]---\left(17\right)$

Wherein,

with

be respectively the current forecasting value of d axle and q axle, θ _efor motor electrical degree,

with

be respectively A phase, B phase and the C phase winding electric current of next moment k+1.

(4) at each, organize under inverter switching device vector signal respectively, calculate the d axle predicted current error of next moment k+1 with q axle predicted current error

wherein, for d axle reference current,

for q axle reference current, by motor reference rotation velocity and output speed, by pi regulator, obtained.

Current error is defined as the poor of the current phasor of current real-time measurement and reference current vector, and object is that current error is controlled to minimum, to reach the object of accurate control.Current error is mainly calculated by dq shaft current amount.

The d shaft current error of current time k

with q shaft current error

be respectively:

$\left\{\begin{array}{c}{e}_d^k=i_d^{*}-i_d^k\\ e_q^k=i_q^{*}-i_q^k\end{array}\right.---\left(18\right)$

From current time k to next moment k+1, the change amount of d shaft current

change amount with q shaft current

be respectively:

$\left\{\begin{array}{c}{\mathrm{\&Delta;i}}_d^{k+1}=i_d^{k+1}-i_d^k\\ {\mathrm{\&Delta;i}}_q^{k+1}=i_q^{k+1}-i_q^k\end{array}\right.---\left(19\right)$

Obtain the d axle predicted current error of next moment k+1 with q axle predicted current error

be respectively:

$\left\{\begin{array}{c}{e}_d^{k+1}=e_d^k-{\mathrm{\&Delta;i}}_d^{k+1}\\ e_q^{k+1}=e_q^k-{\mathrm{\&Delta;i}}_q^{k+1}\end{array}\right.---\left(20\right)$

(5) at each, organize under inverter switching device vector signal respectively, function assesses the cost wherein, k ₁for the ratio of nominal torque and specified magnetic linkage, N is positive integer.Numerical value for N is selected, and Main Basis motor is for the demand of control precision, and A is the regulating error coefficient of N state dq shaft current, and this coefficient can carry out correction adjustment according to control precision, with be respectively the current forecasting value of d axle and the q axle in k+N moment, take current predicated error as benchmark, infer the current forecasting value of the n-hour, object is the control precision in order to regulate most current cost function.

By linear resolution, obtain

with

expression formula be:

$\left\{\begin{array}{c}{i}_d^{k+N}=i_d^k+(N-1)\&times;{\mathrm{\&Delta;i}}_d^{k+1}\\ i_q^{k+N}=i_q^k+(N-1)\&times;{\mathrm{\&Delta;i}}_q^{k+1}\end{array}\right.---\left(21\right)$

Wherein,

with

measured current value by current time k obtains through coordinate transform.

Model prediction Current Control (Model Predictive Current Control, be called for short MPCC) except reducing current error, improve the accuracy of prediction, can also reduce by cost function (Cost Function) frequency of inverter switching device.The effect of cost function is exactly in assessment current error, selects optimum switching vector selector (best performance), guarantees pulsating torque and the pulsation magnetic linkage total amount minimum of FSPMSM, and can effectively reduce inverter switching device loss.

The cost function of the embodiment of the present invention has been considered reference value, predicted value and weighted factor, and object is to make the relevant output characteristic of motor strictly follow the tracks of designated value.

(6) choose cost function G inverter switching device vector signal hour, set it as the driving signal of inverter switching device, the break-make of control inverter switching tube, completes the Current Control to switch flux-linkage permagnetic synchronous motor in real time.

Selective rule for inverter switching device vector signal is as follows:

Cost function minimum value G _min=min{G ₁, G ₂g ₈, wherein, G ₁, G ₂g ₈be respectively 8 groups of cost functions that inverter switching device vector signal is corresponding.With G _min=G ₁for example, switching signal [0 0 1] is selected for control inverter switch motion.

As shown in Figure 3, the current control system of the switch flux-linkage permagnetic synchronous motor based on model prediction mainly comprises that motor body model (FSPMSM), current forecasting module, coordinate transformation module, pi regulator and cost function minimize module (Cost Function Minimization).Motor body model is mainly the simulation model of building according to motor mathematical model, its objective is the research in order to simplify motor control strategy.Current forecasting module is the main body of implementation model prediction, and its function is exactly to survey motor speed and electric current as basis, to go out the motor electric current (A, B, C axle) in next moment by motor body model prediction.By coordinate transformation module, just can obtain dq shaft current predicted value and the measured current component at dq axle.Pi regulator has considered the state of whole control system, by relatively reference rotation velocity and actual measurement rotating speed obtain the current reference amount of motor at dq axle.Cost function, by the correlation of systematic comparison computing reference electric current, measured current and predicted current, obtains the required switching signal of inverter switch device.By the effect of above each functional module, finally realized the Current Control of the switch flux-linkage permagnetic synchronous motor based on model prediction.

Those skilled in the art will readily understand; the foregoing is only preferred embodiment of the present invention; not in order to limit the present invention, all any modifications of doing within the spirit and principles in the present invention, be equal to and replace and improvement etc., within all should being included in protection scope of the present invention.

Claims (6)

1. a current control method for the switch flux-linkage permagnetic synchronous motor based on model prediction, is characterized in that, comprises the steps:

(1) the switching tube state of motor inverter is combined, obtain eight groups of inverter switching device vector signals;

(2), at current time k, under each group inverter switching device vector signal, gather the p phase winding voltage of switch flux-linkage permagnetic synchronous motor respectively

p phase winding electric current

with rotor electrical angle speed omega _r, predict the p phase winding electric current of next moment k+1 wherein, p represents motor A, B and C phase;

(3) respectively under each group inverter switching device vector signal, according to the p phase winding electric current of next moment k+1 in conjunction with rotor winding principle of coordinate transformation, obtain the d axle of next moment k+1 and the current forecasting value i of q axle _d ^k+1and i _q ^k+1;

(4) at each, organize under inverter switching device vector signal respectively, calculate the d axle predicted current error of next moment k+1

with q axle predicted current error

wherein,

for d axle reference current,

for q axle reference current;

(5) at each, organize under inverter switching device vector signal respectively, function assesses the cost

wherein, k ₁for the ratio of nominal torque and specified magnetic linkage, N is positive integer, and A is the regulating error coefficient of N state dq shaft current,

with

be respectively the current forecasting value of d axle and the q axle in k+N moment;

(6) choose cost function G inverter switching device vector signal hour, set it as the driving signal of inverter switching device, the break-make of control inverter switching tube, completes the Current Control to switch flux-linkage permagnetic synchronous motor in real time.

2. the current control method of switch flux-linkage permagnetic synchronous motor as claimed in claim 1, is characterized in that, in described step (5), with expression formula be:

$\left\{\begin{array}{c}{i}_d^{k+N}=i_d^k+(N-1)\&times;{\mathrm{\&Delta;i}}_d^{k+1}\\ i_q^{k+N}=i_q^k+(N-1)\&times;{\mathrm{\&Delta;i}}_q^{k+1}\end{array}\right.,$

Wherein,

with

be respectively d axle and the q shaft current value of current time k,

for from current time k to next moment k+1, the change amount of d shaft current,

for from current time k to next moment k+1, the change amount of q shaft current.

3. the current control method of switch flux-linkage permagnetic synchronous motor as claimed in claim 2, is characterized in that,

with measured current value by current time k obtains through coordinate transform.

4. the current control method of switch flux-linkage permagnetic synchronous motor as claimed any one in claims 1 to 3, is characterized in that, in described step (2), and the p phase winding electric current of next moment k+1

for:

$i_p^{k+1}=i_p^k+\frac{1}{L_p}(u_p^k-R_p{i}_p^k-i_p^k\frac{{\mathrm{dL}}_p}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r-\frac{{\mathrm{d\&psi;}}_{\mathrm{pm},p}}{{\mathrm{d\&theta;}}_r}{\mathrm{\&omega;}}_r)*T_s,$

Wherein, L _pfor p phase winding self-induction, R _pfor p phase winding resistance, θ _rfor motor rotor position angle, ψ _{pm, p}for motor p phase permanent magnetism magnetic linkage, T _sfor switch periods.

5. the current control method of switch flux-linkage permagnetic synchronous motor as claimed in claim 4, is characterized in that, in described step (3), and the current forecasting value i of d axle and q axle _d ^k+1and i _q ^k+1can be expressed as:

$\left[\begin{array}{c}{i}_d^{k+1}\\ i_q^{k+1}\end{array}\right]=\frac{2}{3}\left[\begin{array}{ccc}\cos {\mathrm{\&theta;}}_e& \cos ({\mathrm{\&theta;}}_e-2\mathrm{\&pi;}/3)& \cos ({\mathrm{\&theta;}}_e+2\mathrm{\&pi;}/3)\\ \sin {\mathrm{\&theta;}}_e& \sin ({\mathrm{\&theta;}}_e-2\mathrm{\&pi;}/3)& \sin ({\mathrm{\&theta;}}_e+2\mathrm{\&pi;}/3)\end{array}\right]\&times;\left[\begin{array}{c}{i}_A^{k+1}\\ i_B^{k+1}\\ i_C^{k+1}\end{array}\right],$

Wherein, θ _efor motor synchronous electric angle, i _a ^k+1, i _b ^k+1and i _c ^k+1be respectively A phase, B phase and the C phase winding electric current of next moment k+1.

6. the current control method of the switch flux-linkage permagnetic synchronous motor as described in any one in claim 1 to 5, is characterized in that, in described step (4),

by motor reference rotation velocity and output speed, by pi regulator, obtained.

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