CN102724028A - Alamouti encoding method based on collaborative constellation mapping - Google Patents

Abstract

The invention discloses an Alamouti encoding method based on collaborative constellation mapping, and mainly solves the problems of high block error and worse anti-fading capability of an Alamouti space time code. The method comprises the following steps: dividing the bit stream of a transmitting end into three groups, generating a modulation constellation to map three groups of bits, constructing and combining a collaborative modulation symbol to obtain a transmission symbol, using the Alamouti space time code to transmit the symbol, receiving signals within two symbol periods by a receiving end and constructing a composite receiving signal vector, calculating a zero forcing detection solution vector according to the composite receiving signal vector, using a maximum likelihood estimator to perform first layer judgment on the zero forcing detection solution vector, calculating an European style and performing second layer judgment, and finally performing constellation inverse mapping on the judged result and combining the result to obtain the bit stream. By means of collaborative constellation mapping mode, a better error block function is obtained, the anti-fading capability of the system is enhanced, the communication confidentiality is strengthened at the same time, and the method can be applied to a two-transmit-antenna and one-receive-antenna system.

Description

Alamouti coding method based on cooperative constellation mapping

Technical Field

The invention belongs to the field of wireless communication, relates to an Alamouti coding method, in particular to an Alamouti coding method based on cooperative constellation mapping, and can be used for a two-transmitting one-receiving multi-antenna system.

Background

An important feature of the next generation wireless communication system is that a MIMO system is used, and the channel capacity of the MIMO system can be increased by times compared with the conventional single-input single-output system. MIMO transmission schemes generally fall into two categories: spatial multiplexing and transmit diversity. Multiplexing may improve system capacity, while diversity may improve reliability.

In order to obtain all diversity gains of the MIMO system, Alamouti proposes a space-time block code STBC scheme, which essentially transmits the same information from two antennas after orthogonal coding, and due to the orthogonality of two paths of signals, a receiving end only needs to perform simple linear combination to obtain all diversity gains. Alamouti coding is of great interest due to its simple structure and good performance and has entered the 3GPP standard.

However, in the conventional Alamouti space-time code transmission scheme based on linear combination, signals transmitted by two transmitting antennas are obtained by linear combination of two constellation symbols without cooperative relationship, and the system has poor anti-fading capability and poor block error performance under the scheme.

Disclosure of Invention

The invention aims to provide an Alamouti coding method based on cooperative constellation mapping aiming at the defects of the prior art, so that the anti-fading capability of a system is improved, and the block error rate is reduced.

The technical idea for realizing the invention is as follows: the input information bit stream is divided into three paths through serial-parallel conversion, constellation mapping is respectively carried out, two cooperative constellation symbols are constructed according to the three mapped symbols, then the two cooperative constellation symbols are linearly combined to obtain two sending symbols, and the two symbols are sent out from two antennas in the form of Alamouti space-time codes. The specific implementation process is as follows:

1) the transmitting end divides each L bits in the input information bit stream into 3 groups according to the selected modulation constellation, the bit number of the first group is 1, and the bit number of the second group is L₁Number of bits of the third group is L₂Wherein L is₁>0，L₂>0，L＝L₁+L₂+1；

2) Number of bits L according to second group of bits₁Generating a first modulation constellation Y₁According to the number L of bits of the third group of bits₂Generating the second toneSystem constellation Y₂；

3) Selecting a cooperative constellation X ═ {1, j }, wherein j is an imaginary unit, mapping a first group of bits to a cooperative symbol X in X according to the cooperative constellation X, and according to a first modulation constellation Y₁Mapping the second group of bits to Y₁Of (3) a first modulation symbol y₁According to a second modulation constellation Y₂Mapping the third set of bits to Y₂Second modulation symbol y in (1)₂；

4) According to the cooperative symbol x and the first modulation symbol y₁And a second modulation symbol y₂Two cooperative modulation symbols z are obtained as follows₁And z₂：

$z_1=\frac{y_1}{x},$ $z_2=\frac{y_2}{x};$

5) The two cooperative modulation symbols z are combined₁And z₂Linear combination is carried out to obtain two symbols s to be sent₁And s₂；

6) The two symbols s to be transmitted₁And s₂Sending in the form of Alamouti space-time code: i.e. in the first symbol period, the two antennas transmit the symbol s separately₁And s₂During the second symbol period, the two antennas transmit symbols respectively

And

wherein, represents the conjugate operation;

7) the receiving end receives the transmitted symbols in two consecutive symbol periods, i.e. the signal received by the receiving end in the first symbol period is r₁The signal received by the receiving end in the second symbol period is r₂And from two received signals r₁And r₂Obtaining a composite received signal vector

8) According to the composite received signal vector

Computing zero forcing detection solution vectors

${\hat{z}}_1=H^{-1}\tilde{r},$

Wherein H^-1Is the inverse of the weighted channel matrix H;

9) to zero-forcing detection solution vector

The first layer of judgment is carried out by adopting a maximum likelihood estimation algorithm to obtain

Maximum likelihood estimating symbol vector

To zero-forcing detection solution vector

Multiplying by an imaginary unit j to obtain a zero forcing detection solution vector rotating 90 degrees counterclockwise

Then, the zero-forcing detection solution vector is processed

The first layer of judgment is carried out by adopting a maximum likelihood estimation algorithm to obtain

Maximum likelihood estimating symbol vector ${\hat{u}}_2={[{\hat{u}}_{21},{\hat{u}}_{22}]}^T,$ Superscript T represents a transposition operation;

10) according to the above

Andcomputing zero forcing detection solution vectors

And

maximum likelihood estimating symbol vector

Square of Euclidean distanceAccording to the above

And

computing zero forcing detection solution vectors

And

maximum likelihood estimating symbol vector

Square of Euclidean distance

According to delta₁And delta₂Making a second layer decision if delta₁＜δ₂If so, the decision result is:

otherwise, the judgment result is:

${\hat{y}}_2={\hat{u}}_{22};$

11) according to the judgment result of the cooperative constellation X pairInverse mapping is carried out to obtain a first group of decision bits, and a first modulation constellation Y is obtained₁For the judgment resultInverse mapping is carried out to obtain a second group of decision bits, and the decision bits are modulated according to a second modulation constellation Y₂For the judgment result

And carrying out inverse mapping to obtain a third group of decision bits, and combining the three groups of bits into L bits according to a bit grouping rule.

The invention has the following advantages:

1) the invention utilizes the cooperative signal decomposition principle to carry out cooperative constellation modulation design at the transmitting end, and compared with a linear combination mode, the invention obtains better error block performance under the same frequency spectrum efficiency and enhances the anti-fading capability of the system;

2) the invention can flexibly configure the constellation and is suitable for adaptive modulation;

3) the modulation signal characteristics adopted by the invention are different from the traditional M-QAM signal characteristics, the modulation mode is difficult to identify, and the communication confidentiality is enhanced.

Drawings

FIG. 1 is a flow chart of the present invention;

FIG. 2 is a constellation diagram generated by the present invention;

fig. 3 is a block error rate performance simulation diagram of the present invention.

Detailed Description

The technical process of the present invention will be further described below by way of the accompanying drawings and examples.

Referring to fig. 1, the specific implementation steps of the present invention include:

step 1: the transmitting end divides each L bits of the input information bit stream into 3 groups according to the selected modulation constellation, the bit number of the first group is 1, the bit number of the second group is L₁The number of bits of the third group is L₂。L₁、L₂The calculation rule of (1) is: when L is an odd number:

when L is an even number:

there are several ways of bit grouping here, for example the first way: the first bit is the first group, 2 nd to L th₁+1 bits as a second group, Lth₁+ 2-L bit streams as a third group; the second mode is as follows: the Lth bit is a first group of bits 1 to L₁Bits being of a second group, Lth₁Bit + 1-L-1 is the third group; the third mode is as follows: l th₁+1 bits as the first group, 1 st to L₁Bits being of a second group, Lth₁+ 2-L bit streams are in the third group. The first mode is taken as an example here.

Step 2: number of bits L according to second group of bits₁Generating a first modulation constellation Y₁According to the number L of bits of the third group of bits₂Generating a second modulation constellation Y₂。

2a) Number of bits L according to second group of bits₁Generating a first modulation constellation Y₁The method comprises the following steps:

when L is₁When =2, the first modulation constellation Y₁Is {1+ j, 3-j, -1-j, -3+ j }, j represents an imaginary unit;

when L is₁>0 and L₁Not equal to 2, from the conventional

Selection from constellationsObtaining a first modulation constellation Y from each constellation point₁Is to traverse

Selecting constellation points of the constellation from the constellation points, wherein the constellation points have a Euclidean distance square of 8 with complex number 1+ j as a first modulation constellation Y₁As shown in fig. 2. Wherein FIG. 2 (a) is when L₁When the symbol is =1, the solid points and the hollow points form a 4QAM constellation; FIG. 2 (b) shows that when L is₁When the symbol number is =2, the solid points and the hollow points form a constellation {1+3j, 1+ j, 3-j, 1-j, -1-3j, -1-j, -3+ j, -1+ j }; FIG. 2 (c) is L₁When =3, the solid points and the hollow points form a 16QAM constellation; FIG. 2 (d) is when L₁When the symbol is =4, the solid points and the hollow points form a 32QAM constellation; FIG. 2 (e) shows that when L is₁And when =5, the solid dots and the hollow dots form a 64QAM constellation. The solid dots in fig. 2 constitute a first modulation constellation Y₁The hollow point is Y₁The constellation diagram formed by solid points and hollow points has the invariance of rotating 90 degrees, 180 degrees and 270 degrees after rotating 90 degrees anticlockwise.

2b) Number of bits L according to third group₂Generating a second modulation constellation Y₂The method comprises the following steps:

when L is₂When =2, the second modulation constellation Y₁Is {1+ j, 3-j, -1-j, -3+ j };

when L is₂>0 and L₂When not equal to 2, from

Selection from constellations

Obtaining a second modulation constellation Y from the constellation points₂Is to traverseSelecting constellation points of the constellation from the constellation points, wherein the constellation points have Euclidean distance squared of multiple number 1+ j of 8 as second modulation constellation Y₂The constellation point of (1).

And step 3: selecting a cooperative constellation X ═ {1, j }, wherein j is an imaginary unit, mapping a first group of bits to a cooperative symbol X in X according to the cooperative constellation X, and according to a first modulation constellation Y₁Mapping the second group of bits to Y₁Of (3) a first modulation symbol y₁According to a second modulation constellation Y₂Mapping the third set of bits to Y₂Second modulation symbol y in (1)₂。

And 4, step 4: for the above cooperation symbol x and the first modulation symbol y₁A second modulation symbol y₂Two cooperative modulation symbols z are obtained according to the following formula₁And z₂：

$z_1=\frac{y_1}{x},$ $z_2=\frac{y_2}{x},$

When x is 1, the first modulation constellation Y₁And a second modulation constellation Y₂Remaining unchanged, when x ═ j, the first modulation constellation Y₁And a second modulation constellation Y₂While rotating 90 deg. counterclockwise.

And 5: for the above two cooperative modulation symbolsz₁And z₂Linear combination is carried out to obtain two symbols s to be sent₁And s₂：

$\left[\begin{array}{c}{s}_1\\ {\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">s</figure-callout>}}_2\end{array}\right]=\begin{array}{}\end{array}\left[\begin{array}{cc}l& k\\ l& -k\end{array}\right]\left[\begin{array}{c}{z}_1\\ z_2\end{array}\right]=\mathrm{Pz},$

Wherein $P=\left[\begin{array}{cc}l& k\\ l& -k\end{array}\right]$ In order to normalize the matrix for the power,

E₁is a first constellation Y₁Average energy of medium constellation points, E₂For the second modulation constellation Y₂Average energy of the medium constellation points; z is ═ z₁,z₂]^TFor cooperative modulation symbol vectors, the superscript T denotes the transpose operation.

Step 6: for the above s₁And s₂And sending out according to an Alamouti space-time code form: in the first symbol period, two antennas respectively transmit a symbol s₁And s₂In the second symbol period, two antennas transmit symbols respectivelyAnd

where denotes the conjugate operation.

And 7: the receiving end receives the symbol sent by the sending end in two continuous symbol periods, namely the signal received by the receiving end in the first symbol period is r₁The signal received by the receiving end in the second symbol period is r₂And from two received signals r₁And r₂Constructing a composite received signal vector

7a) In the first symbol period, the signal received by the receiving end is r₁：

r₁＝h₁s₁+h₂s₂+n₁，

Wherein r is₁Representing a signal received by a receiving end in a first symbol period; s₁And s₂Respectively representing the transmission of two antennas at the transmitting end in the first symbol periodSending a signal; h is₁Represents the channel transmission coefficient h between the first transmitting antenna and the receiving antenna at the transmitting end in the first symbol period₂Representing a channel transmission coefficient between a second sending antenna and a receiving antenna of the sending end in a second symbol period; n is₁Representing the noise of the receiving end in the first symbol period, the mean value of the noise is zero, and the variance is sigma²(ii) a gaussian distribution of;

7b) in the second symbol period, the signal received by the receiving end is r₂：

$r_2=-h_1{\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">s</figure-callout>}}_2^{*}+h_2{\mathrm{<figure-callout\; id="1"\; label="s"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png"\; state="\{\{state\}\}">s</figure-callout>}}_1+{\mathrm{<figure-callout\; id="2"\; label="n"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">n</figure-callout>}}_2,$

Wherein r is₂Indicating the signal received by the receiving end in the second symbol period;

and

respectively representing the sending signals of two antennas in a second symbol period by the sending end; h is₁Represents the channel transmission coefficient h between the first transmitting antenna and the receiving antenna at the transmitting end in the first symbol period₂Representing a channel transmission coefficient between a second sending antenna and a receiving antenna of the sending end in a second symbol period; n is₂Representing the noise of the receiving end in the second symbol period, the mean value of the noise is zero, and the variance is sigma²(ii) a gaussian distribution of;

7c) based on the signal r received in the first symbol period₁And the signal r received in the second symbol period₂Obtaining a composite received signal vector

$\tilde{r}=\left[\begin{array}{c}{r}_1\\ {\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">r</figure-callout>}}_2^{*}\end{array}\right]=\left[\begin{array}{cc}{h}_1& h_2\\ {\mathrm{<figure-callout\; id="2"\; label="h"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">h</figure-callout>}}_2^{*}& {-\mathrm{<figure-callout\; id="1"\; label="h"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png"\; state="\{\{state\}\}">h</figure-callout>}}_1^{*}\end{array}\right]\left[\begin{array}{c}{s}_1\\ {\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">s</figure-callout>}}_2\end{array}\right]+\left[\begin{array}{c}{n}_1\\ {\mathrm{<figure-callout\; id="2"\; label="n"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">n</figure-callout>}}_2^{*}\end{array}\right]$

$=\left[\begin{array}{cc}{h}_1& h_2\\ {\mathrm{<figure-callout\; id="2"\; label="h"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">h</figure-callout>}}_2^{*}& {-\mathrm{<figure-callout\; id="1"\; label="h"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png"\; state="\{\{state\}\}">h</figure-callout>}}_1^{*}\end{array}\right]\left[\begin{array}{cc}l& k\\ l& -k\end{array}\right]\left[\begin{array}{c}{z}_1\\ {\mathrm{<figure-callout\; id="2"\; label="z"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">z</figure-callout>}}_2\end{array}\right]+\left[\begin{array}{c}{n}_1\\ {\mathrm{<figure-callout\; id="2"\; label="n"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">n</figure-callout>}}_2^{*}\end{array}\right],$

$=\tilde{H}\mathrm{Pz}+n$

$=\mathrm{Hz}+\tilde{n}$

Wherein,

is a two-dimensional column vector, r₁Indicating the received signal at the receiving end in the first symbol period, r₂Indicating a receiving signal in a second symbol period of the receiving end, wherein the superscript T indicates transposition operation and the index indicates conjugate operation; $\tilde{H}=\left[\begin{array}{cc}{h}_1& h_2\\ {\mathrm{<figure-callout\; id="2"\; label="h"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">h</figure-callout>}}_2^{*}& -{\mathrm{<figure-callout\; id="1"\; label="h"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png"\; state="\{\{state\}\}">h</figure-callout>}}_1^{*}\end{array}\right]$ is a channel matrix between a transmitting end and a receiving end, where h₁Representing the channel transmission coefficient from the first transmit antenna to the receive antenna, where h₂Denotes the channel transmission coefficient from the second transmit antenna to the receive antenna;

is a weighted channel matrix; p is a power normalization matrix;

is zero mean and covariance matrix is σ²I₂Complex gaussian noise vector, σ²Representing the mean power of the Gaussian noise, I₂Representing a 2 x 2 identity matrix.

And 8: according to the composite received signal vector

Computing zero forcing detection solution vectors

${\hat{z}}_1=\left[\begin{array}{c}{\hat{z}}_{11}\\ {\hat{z}}_{12}\end{array}\right]=H^{-1}\tilde{r}$

$=\frac{\sqrt{{\mathrm{<figure-callout\; id="1"\; label="E"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png"\; state="\{\{state\}\}">E</figure-callout>}}_1+{\mathrm{<figure-callout\; id="2"\; label="E"\; filenames="2012102079146100003DEST\_PATH\_IMAGE001.png,2012102079146100003DEST\_PATH\_IMAGE002.png"\; state="\{\{state\}\}">E</figure-callout>}}_2}}{2\left|H\right|}{H}^H\tilde{r},$

Wherein,is a two-dimensional column vector, and is,

and

representing two zero-forcing detection solutions, H representing a weighted channel matrix, | H | representing a determinant of the weighted channel matrix H, superscript H representing a conjugate transpose operation, E₁Is a first modulation constellation Y₁Average energy of medium constellation points, E₂For the second modulation constellation Y₂Average energy of the medium constellation points.

And step 9: to zero-forcing detection solution vector

The first layer of judgment is carried out by adopting a maximum likelihood estimation algorithm to obtain

Maximum likelihood estimating symbol vector

To zero-forcing detection solution vector

Multiplying by an imaginary unit j to obtain a zero forcing detection solution vector rotating 90 degrees counterclockwise

Then, the zero-forcing detection solution vector is processedThe first layer of judgment is carried out by adopting a maximum likelihood estimation algorithm to obtainMaximum likelihood estimating symbol vectorThe superscript T represents the transpose operation, whose maximum likelihood estimation is performed using the following formula:

<math> <mrow> <msub> <mover> <mi>u</mi> <mo>^</mo> </mover> <mn>1</mn> </msub> <mo>=</mo> <mi>arg</mi> <munder> <mi>min</mi> <mrow> <mi>y</mi> <mo>&Element;</mo> <mi>Y</mi> </mrow> </munder> <mo>|</mo> <mo>|</mo> <msub> <mover> <mi>z</mi> <mo>^</mo> </mover> <mn>1</mn> </msub> <mo>-</mo> <mi>y</mi> <mo>|</mo> <mo>|</mo> <mo>,</mo> </mrow> </math>

<math> <mrow> <msub> <mover> <mi>u</mi> <mo>^</mo> </mover> <mn>2</mn> </msub> <mo>=</mo> <mi>arg</mi> <munder> <mi>min</mi> <mrow> <mi>y</mi> <mo>&Element;</mo> <mi>Y</mi> </mrow> </munder> <mo>|</mo> <mo>|</mo> <msub> <mover> <mi>z</mi> <mo>^</mo> </mover> <mn>2</mn> </msub> <mo>-</mo> <mi>y</mi> <mo>|</mo> <mo>|</mo> <mo>,</mo> </mrow> </math>

wherein,

representing zero forcing detection solution vectorsThe maximum likelihood estimation symbol vector of (1);

zero forcing detection solution vector representing 90 degrees of counterclockwise rotationThe maximum likelihood estimation symbol vector of (1); argmin represents the variable value at which the target function takes the minimum value; i | · | | represents a 2-norm operation; y denotes a modulation constellation from the first Y₁And a second modulation constellation Y₂Respectively, wherein Y represents a two-dimensional column vector formed by randomly selecting one constellation point, and Y represents a modulation constellation Y from the first modulation constellation₁And a second modulation constellation Y₂Respectively, and respectively selecting a two-dimensional column vector set formed by any constellation point.

Step 10: according to the above

And

computing zero forcing detection solution vectors

Andmaximum likelihood estimating symbol vector

Square of Euclidean distance

According to the above

And

computing zero forcing detection solution vectors

And

maximum likelihood estimating symbol vectorSquare of Euclidean distance

According to delta₁And delta₂Making a second layer decision if delta₁＜δ₂If so, the decision result is:

otherwise, the judgment result is:

${\hat{y}}_1={\hat{u}}_{21},{\hat{y}}_2={\hat{u}}_{22}.$

step 11: according to the judgment result of the cooperative constellation X pair

Inverse mapping is carried out to obtain a first group of decision bits, and a first modulation constellation Y is obtained₁For the judgment result

Inverse mapping is carried out to obtain a second group of decision bits, and the decision bits are modulated according to a second modulation constellation Y₂For the judgment result

And carrying out inverse mapping to obtain a third group of decision bits, and combining the three groups of bits into L bits in sequence.

The effects of the present invention can be further illustrated by the following simulations:

1. simulation conditions

The method is characterized in that a communication system with 2 antennas configured at a sending end and 1 antenna configured at a receiving end is adopted, a quasi-static Rayleigh flat fading channel is adopted as a channel, the channel coefficient obeys complex Gaussian distribution with the mean value of zero and the variance of 1. The simulated baud rates are 2.5bpcu, 3.5bpcu, 4.5bpcu, and 5.5bpcu, respectively, where bpcu represents the number of bits per channel.

2. Simulation content and results

The block error rate of the present invention and the linear combination scheme are compared in simulation, and the simulation result is shown in fig. 3. As can be seen from FIG. 3, when the Baud rates are 2.5bpcu, 3.5bpcu, 4.5bpcu and 5.5bpcu, the method has better block error performance than that of the linear combination scheme, and obtains the signal-to-noise ratio gain of 0.7dB to 1 dB.

Claims (7)

1. An Alamouti coding method based on cooperative constellation mapping comprises the following steps:

1) the transmitting end divides each L bits in the input information bit stream into 3 groups according to the selected modulation constellation, the bit number of the first group is 1, and the bit number of the second group is L₁Number of bits of the third group is L₂Wherein L is₁>0，L₂>0，L＝L₁+L₂+1；

2) Number of bits L according to second group of bits₁Generating a first modulation constellation Y₁According to the third group ratioNumber of bits L₂Generating a second modulation constellation Y₂；

3) Selecting a cooperative constellation X ═ {1, j }, wherein j is an imaginary unit, mapping a first group of bits to a cooperative symbol X in X according to the cooperative constellation X, and according to a first modulation constellation Y₁Mapping the second group of bits to Y₁Of (3) a first modulation symbol y₁According to a second modulation constellation Y₂Mapping the third set of bits to Y₂Second modulation symbol y in (1)₂；

4) According to the cooperative symbol x and the first modulation symbol y₁And a second modulation symbol y₂Two cooperative modulation symbols z are obtained as follows₁And z₂：

$z_1=\frac{y_1}{x},$ $z_2=\frac{y_2}{x};$

5) The two cooperative modulation symbols z are combined₁And z₂Linear combination is carried out to obtain two symbols s to be sent₁And s₂；

6) The two symbols s to be transmitted₁And s₂Sending in the form of Alamouti space-time code: i.e. in the first symbol period, the two antennas transmit the symbol s separately₁And s₂During the second symbol period, the two antennas transmit symbols respectively

And

wherein, represents the conjugate operation;

7) the receiving end receives the transmitted symbols in two consecutive symbol periods, i.e. the signal received by the receiving end in the first symbol period is r₁The signal received by the receiving end in the second symbol period is r₂And from two received signals r₁And r₂Obtaining a composite received signal vector

8) According to the composite received signal vector

Computing zero forcing detection solution vectors

${\hat{z}}_1=H^{-1}\tilde{r},$

Wherein H^-1Is the inverse of the weighted channel matrix H;

9) to zero-forcing detection solution vector

The first layer of judgment is carried out by adopting a maximum likelihood estimation algorithm to obtainMaximum likelihood estimating symbol vector

To zero-forcing detection solution vectorMultiplying by an imaginary unit j to obtain a zero forcing detection solution vector rotating 90 degrees counterclockwise

Then, the zero-forcing detection solution vector is processed

The first layer of judgment is carried out by adopting a maximum likelihood estimation algorithm to obtainMaximum likelihood estimating symbol vector ${\hat{u}}_2={[{\hat{u}}_{21},{\hat{u}}_{22}]}^T,$ Superscript T represents a transposition operation;

10) according to the above

And

computing zero forcing detection solution vectorsAnd

maximum likelihood estimating symbol vectorSquare of Euclidean distance

According to the above

And

computing zero forcing detection solution vectors

And

maximum likelihood estimating symbol vector

Square of Euclidean distance

According to delta₁And delta₂Making a second layer decision if delta₁＜δ₂If so, the decision result is:

otherwise, the judgment result is:

${\hat{y}}_2={\hat{u}}_{22};$

11) according to the judgment result of the cooperative constellation X pairInverse mapping is carried out to obtain a first group of decision bits, and a first modulation constellation Y is obtained₁For the judgment result

Inverse mapping is carried out to obtain a second group of decision bits, and the decision bits are modulated according to a second modulation constellation Y₂For the judgment result

And carrying out inverse mapping to obtain a third group of decision bits, and combining the three groups of bits into L bits according to a bit grouping rule.

2. The Alamouti encoding method based on cooperative constellation mapping according to claim 1, wherein the number of bits of the second set of bits in step 1) is L₁And the number L of bits of the third group₂The calculation rule of (1) is: when the total number of bits L is odd: $L_1=L_2=\frac{L-1}{2};$ when L is an even number: $L_1=\frac{L}{2}-1.$ $L_2=\frac{L}{2}.$

3. the Alamouti encoding method based on cooperative constellation mapping according to claim 1, wherein said step 2) generates a first modulation constellation Y₁The method comprises the following steps:

when L is₁When =2, the first modulation constellation Y₁Is {1+ j, 3-j, -1-j, -3+ j }, j represents an imaginary unit;

when L is₁>0 and L₁When not equal to 2, from

Selection from constellations

Obtaining a first modulation constellation Y from each constellation point₁Is to traverse

Selecting constellation points of the constellation from the constellation points, wherein the constellation points have a Euclidean distance square of 8 with complex number 1+ j as a first modulation constellation Y₁The constellation point of (1).

4. The Alamouti encoding method based on cooperative constellation mapping according to claim 1, wherein said step 2) generates a second modulation constellation Y₂The method comprises the following steps:

when L is₂When =2, the second modulation constellation Y₂Is {1+ j, 3-j, -1-j, -3+ j }, j represents an imaginary unit;

when L is₂>0 and L₂When not equal to 2, from

Selection from constellations

Obtaining a second modulation constellation Y from the constellation points₂Is to traverse

Selecting constellation points of the constellation from the constellation points, wherein the constellation points have Euclidean distance squared of multiple number 1+ j of 8 as second modulation constellation Y₂The constellation point of (1).

5. The Alamouti encoding method based on cooperative constellation mapping according to claim 1, wherein said step 5) consists in combining two cooperative modulation symbols z₁And z₂Linear combination is carried out to obtain two symbols s to be sent₁And s₂The method is carried out according to the following formula:

$\left[\begin{array}{c}{s}_1\\ s_2\end{array}\right]=\begin{array}{}\end{array}\left[\begin{array}{cc}l& k\\ l& -k\end{array}\right]\left[\begin{array}{c}{z}_1\\ z_2\end{array}\right]=\mathrm{Pz},$

wherein, $P=\left[\begin{array}{cc}l& k\\ l& -k\end{array}\right]$ in order to normalize the matrix for the power,

E₁is a first modulation constellation Y₁Average energy of medium constellation points, E₂For the second modulation constellation Y₂Average energy of the medium constellation points;

for cooperative modulation symbol vectors, the superscript T denotes the transpose operation.

6. The Alamouti encoding method based on cooperative constellation mapping according to claim 1, wherein the composite received signal vector in step 7) is

Comprises the following steps:

$\tilde{r}=\left[\begin{array}{c}{r}_1\\ r_2^{*}\end{array}\right]=\left[\begin{array}{cc}{h}_1& h_2\\ h_2^{*}& {-h}_1^{*}\end{array}\right]\left[\begin{array}{c}{s}_1\\ s_2\end{array}\right]+\left[\begin{array}{c}{n}_1\\ n_2^{*}\end{array}\right]$

$=\left[\begin{array}{cc}{h}_1& h_2\\ h_2^{*}& {-h}_1^{*}\end{array}\right]\left[\begin{array}{cc}l& k\\ l& -k\end{array}\right]\left[\begin{array}{c}{z}_1\\ z_2\end{array}\right]+\left[\begin{array}{c}{n}_1\\ n_2^{*}\end{array}\right],$

$=\tilde{H}\mathrm{Pz}+\tilde{n}$

$=\mathrm{Hz}+\tilde{n}$

wherein r is₁Indicating the received signal of the receiving antenna in the first symbol period, r₂Indicating the received signal of the receiving antenna in the second symbol period; $\tilde{H}=\left[\begin{array}{cc}{h}_1& h_2\\ h_2^{*}& -h_1^{*}\end{array}\right]$ denotes the channel matrix between the transmitting end and the receiving end, where h₁Representing the channel fading coefficient, h, from the first transmit antenna to the receive antenna₂Representing the channel fading coefficient from the second transmitting antenna to the receiving antenna, and representing the conjugate operation;representing the complex Gaussian noise vector, n, at the receiving end₁Representing the Gaussian noise at the receiver end in the first symbol period, n₂Representing the Gaussian noise at the receiving end in the second symbol period;

p is a power normalization matrix.

7. The Alamouti encoding method based on cooperative constellation mapping according to claim 1, wherein the solution vector for zero-forcing detection using maximum likelihood estimation algorithm of step 9) is

And a zero forcing detection solution vector rotated 90 DEG counterclockwise

Carrying out maximum likelihood estimation according to the following formula:

<math> <mrow> <msub> <mover> <mi>u</mi> <mo>^</mo> </mover> <mn>1</mn> </msub> <mo>=</mo> <mi>arg</mi> <munder> <mi>min</mi> <mrow> <mi>y</mi> <mo>&Element;</mo> <mi>Y</mi> </mrow> </munder> <mo>|</mo> <mo>|</mo> <msub> <mover> <mi>z</mi> <mo>^</mo> </mover> <mn>1</mn> </msub> <mo>-</mo> <mi>y</mi> <mo>|</mo> <mo>|</mo> <mo>,</mo> </mrow> </math>

<math> <mrow> <msub> <mover> <mi>u</mi> <mo>^</mo> </mover> <mn>2</mn> </msub> <mo>=</mo> <mi>arg</mi> <munder> <mi>min</mi> <mrow> <mi>y</mi> <mo>&Element;</mo> <mi>Y</mi> </mrow> </munder> <mo>|</mo> <mo>|</mo> <msub> <mover> <mi>z</mi> <mo>^</mo> </mover> <mn>2</mn> </msub> <mo>-</mo> <mi>y</mi> <mo>|</mo> <mo>|</mo> <mo>,</mo> </mrow> </math>

wherein,

representing zero forcing detection solution vectors

The maximum likelihood estimation symbol vector of (1);

zero forcing detection solution vector representing 90 degrees of counterclockwise rotation

The maximum likelihood estimation symbol vector of (1); argmin represents the variable value at which the target function takes the minimum value; i | · | | represents a two-norm operation; y denotes a modulation constellation from the first Y₁And a second modulation constellation Y₂Respectively, wherein Y represents a two-dimensional column vector formed by randomly selecting one constellation point, and Y represents a modulation constellation Y from the first modulation constellation₁And a second modulation constellation Y₂Respectively, and respectively selecting a two-dimensional column vector set formed by any constellation point.

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