CN102468764B - The adjustment gain method of resonant converter and device thereof - Google Patents
The adjustment gain method of resonant converter and device thereof Download PDFInfo
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- CN102468764B CN102468764B CN201010538316.8A CN201010538316A CN102468764B CN 102468764 B CN102468764 B CN 102468764B CN 201010538316 A CN201010538316 A CN 201010538316A CN 102468764 B CN102468764 B CN 102468764B
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
- Y02P80/00—Climate change mitigation technologies for sector-wide applications
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Abstract
The invention provides a kind of adjustment gain apparatus and method thereof of resonant converter, in order to the DC current gain of regulable control resonant converter, to improve the power supply serviceability of circuit.The present invention to this resonant converter, in order to produce one first control signal, controls the primary drive of this resonant converter by a phase shift block assembly whereby, and produces one second control signal, controls the secondary driver of this resonant converter whereby.Wherein this first control signal and this second control signal have a phase shift, in order to control a DC current gain of this resonant converter.The present invention makes it reach optimization by the DC current gain of regulable control resonant converter, and then increases power supply service efficiency, improves the serviceability of power supply.
Description
Technical field
The present invention relates to a kind of adjustment gain method and device thereof, particularly relate to a kind of the adjustment gain method and the device thereof that are applied to resonant converter.
Background technology
Effective manipulation of power supply and application, when developing consumption-orientation and industrial electronics, are one main consider emphasis.And be resonant converter (resonantconverter) for wherein a kind of pattern of power conversion system now, it includes inductor-capacitor (L-C) network of various configuration, in order to mould the waveform of the curtage of given switch element in whole circuit.Various electronic component can be used to design one resonant converter, and synchronous rectifier (synchronousrectifiers, SRs) is then often used to alternating current (AC) to be transferred in the application of direct current (DC).Synchronous rectifier includes the diode and a transistor (being typically a mos field effect transistor MOSFET) that are connected in parallel.When operational applications, when this diode is in a forward bias, then this transistor can be activated to lower pressure drop, so as to reducing loss in circuit.
Under practical working situation, no matter whether use synchronous rectifier, this resonant converter all can meet with a problem, in the scope namely normally worked at this transducer, its DC current gain (nVo/Vin) is nonmonotonicity change, and namely multiple operating frequency may appear in corresponding same gain.Moreover in any case the frequency of change resonant converter, its DC current gain cannot be all null value or the null value that levels off to.This is using the usefulness of reduction resonant converter 101 as the Energy control mechanism in electronic application or consumer device, and this usefulness refers to that obtain one stablizes and the gain reaction of monotonicity (as linear) according to frequency, damages to avoid element; And produce one zero DC current gain, to obtain a smooth-going starting waveform by a no-voltage or the no-voltage that levels off to.
Graph of a relation between its switching frequency and DC current gain when Figure 1A-Fig. 1 C to disclose for resonant converter in its normal range of operation.It should be noted that for typical resonance formula converter application (as LLC resonant converter), being difficult to only reduce its DC current gain to one null value, to reach the requirement smoothly started from no-voltage by increasing switching frequency.For example, Figure 1A discloses the typical DC current gain curve to both underload and heavy duty, x-axis represents the normalizing operation frequency of circuit, for the ratio of the characteristic resonant frequency of frequency of operation and this resonant circuit, y-axis represents DC current gain level, for output voltage Vo takes advantage of the ratio of n (turn ratio of transformer T) and this input voltage vin.For light load operation, in same frequency range, its DC current gain curve 1001 is higher than the curve 1003 of heavy duty person.Along with the increase of switching frequency, this underloaded gain is all greater than a particular value, and as shown in Figure 1A 0.7.Continuous increase frequency significantly can not reduce the net value of DC current gain not to the utmost again to much higher value, and can increase the loss of circuit element and cause circuit to damage (as overheated, burning out).
Figure 1B is the circuit diagram according to an example embodiment, the resonant converter of the one or more parasitic capacitance of its display tool.In the present embodiment, the situation of circuit when parasitic capacitance Cp and Cs is imported the resonant network of this resonant converter Circuits System and rectifier network respectively to reflect real work.Usually, electronic component all contains certain parasitic parameter, as parasitic capacitance, inductance or resistance etc.Coil or the switch element of transformer all include parasitic capacitance.Therefore electric capacity CP1005 and CS1007 can be added in this transducer to reflect situation during circuit real work.
Unfortunately, when light load operation, electric capacity can cause charge pump effect and increase DC current gain.This phenomenon is illustrated in Fig. 1 C, and it discloses the DC current gain curve that example embodiment of the present invention is simulated in the resonant converter adding capacity cell both underload and heavy duty.Charge pump effect, as shown in the curve 1009 in figure, because electric capacity 1005 and 1007 can make more multi-energy be passed to the output of this resonant converter, and causes output voltage to increase.What this phenomenon was more detailed is described in U.S. Patent number the 7th, and 733, in 669, be all incorporated in this and consider and examine.
It should be noted that phenomenon in fig. 1 c, the curve 1009 i.e. corresponding different operating frequency (as frequency f 1 and f2) that the existence due to this charge pump charge pump effect makes the DC current gain curve 1009a originally answering monotone variation under underloading situation become a non-monotonic change has same gain level (as gain=A).Make the control of circuit occur so chaotic, the gain characteristic that cannot expect will cause this transducer unstable and bear the consume of high power supply.
Summary of the invention
In view of restriction and the defect of known technology, the present invention proposes one " the adjustment gain method of resonant converter and device thereof ", make it reach optimization by the DC current gain of regulable control resonant converter, and then increase power supply service efficiency, improve the serviceability of power supply.
This section of outline some feature of the present invention, other features will be described in follow-up paragraph.The present invention is by the definition of additional claim, and it is herein incorporated paragraph as a reference.
Main purpose of the present invention is for providing a kind of adjustment gain method and device of resonant converter, by the DC current gain of regulable control resonant converter, make it thus obtain one according to frequency/phase shifting angle or other changes to stablize and the gain of monotonicity (as linear), and avoid element to damage; One zero DC current gain can be produced simultaneously, to be obtained a smooth-going starting waveform by a no-voltage, power supply service efficiency can be increased simultaneously.
For reaching above-mentioned purpose, a broad embodiment of the present invention is for providing a kind of adjustment gain method being applied to resonant converter, and it comprises step: produce one first control signal, in order to control a primary drive of a resonant converter; And producing one second control signal, in order to control this resonant converter one secondary driver, wherein this first control signal and this second control signal have a phase shifting angle, in order to control a DC current gain of this resonant converter.
For reaching above-mentioned purpose, another broad embodiment of the present invention is for providing an adjustment gain apparatus, and it comprises: a delay circuit; And a control module, be coupled to this delay circuit, and assembly provides and exports this delay circuit to, produces in order to the output voltage according to a resonant converter primary drive that one first control signal removes to control this resonant converter; And produce the secondary driver that one second control signal removes to control this resonant converter; Wherein this first control signal and this second control signal have a phase shifting angle, in order to control a DC current gain of this resonant converter.
For reaching above-mentioned purpose, another broad embodiment of the present invention is for providing an adjustment gain apparatus, and it comprises: a resonant converter, has a primary drive and a secondary driver; And a phase shift block, be coupled to this primary drive and this secondary driver, this phase shift block produces one first control signal to control this primary drive and to produce one second control signal to control this secondary driver; Wherein this first control signal and this second control signal have a phase shifting angle, in order to control a DC current gain of this resonant converter.
The present invention makes it reach optimization by the DC current gain of regulable control resonant converter, and then increases power supply service efficiency, improves the serviceability of power supply.
The present invention must be illustrated by following accompanying drawing and embodiment, makes a clearer understanding.
Accompanying drawing explanation
Figure 1A to Fig. 1 C: it discloses graph of a relation when applying for resonant converter between its switching frequency and DC current gain;
Fig. 2: its resonant converter assembly disclosing example embodiment of the present invention has the block diagram of a phase shift block, in order to regulate the DC current gain of resonant converter;
Fig. 3: it discloses the LLC serial-resonant converter circuit schematic diagram of example embodiment of the present invention, and it controls direct current by switching frequency and increases electric gain;
Fig. 4: it discloses the time-histories schematic diagram of the primary drive of resonant converter and the phase shifting angle of secondary driver in example embodiment of the present invention;
Fig. 5 A: it discloses the DC current gain curve that example embodiment of the present invention is simulated with the means of compensate for parasitic capacitance;
Fig. 5 B and Fig. 5 C: it discloses the phase shift of resonant converter and the graph of a relation of voltage gain in various embodiments of the present invention;
Fig. 6 A: in its announcement example embodiment of the present invention, resonant converter combo one phase shift block is to regulate the circuit diagram of DC current gain;
Fig. 6 B and Fig. 6 C: it discloses in various embodiments of the present invention using the FREQUENCY CONTROL of phase shift block, Dead Time regulable control and phase shifting control ability as the application schematic diagram of individual control module;
Fig. 6 D: it discloses resonant converter and the circuit diagram using different error amplifiers to regulate the phase shift block institute assembly of DC current gain in example embodiment of the present invention;
Fig. 6 E: it discloses the flow chart regulating resonant converter DC current gain in example embodiment of the present invention;
Fig. 7 A to Fig. 7 E: it discloses in various embodiments of the present invention, and resonant converter is in time according to the behavior timing diagram of phase shifting angle between main and secondary driver, and wherein this phase shifting angle exceedes preplanned Dead Time between driver;
Fig. 8 A to Fig. 8 E: it to disclose in this each embodiment resonant converter in time according to the behavior timing diagram of phase shifting angle between main and secondary driver, wherein between this main and secondary driver, phase shifting angle has a preplanned Dead Time, and it is less than the phase shift between driver;
Fig. 9: its phase shift disclosing a resonant converter in example embodiment of the present invention regulates strain stress relation figure.
Description of reference numerals in above-mentioned accompanying drawing is as follows:
1000,1008: DC current gain curve chart
1001,1003,1009,1009a: DC current gain curve
1004: circuit diagram
1005,1007: capacity cell
100: resonant converter application circuit system
101: resonant converter
103: primary drive
105: secondary driver, synchronous rectifier driver
107: phase shift block
201,203: synchronous rectifier
201a-b, 203a-b: main switch element
205: synchronous rectifier, resonant mode inductor
207: magnetization load inductance
209: resonant mode capacitor
211: transformer
213: main coils
215a, 215b: secondary coil
217: rectangular wave input voltage
219: load voltage
225: dotted line
225a: rectangular wave generator
227: dotted line
227a: rectifier network
229: transformer turn ratio
235: resonant network
300: time-histories schematic diagram
301: rectangular wave inputs
303: the pulse of secondary (synchronous rectifier) driver
303a: secondary driver pulse
305,307: primary drive pulse
410: DC current gain curve chart
411,413: DC current gain curve
415:x axle
417: block
500: application circuit
501: resonant converter
503: phase shift block
505: error amplifier
507,509: resistance
511: resonant converter control module
511a: frequency adjustment control inputs
511b: Power MOSFET inputs
513: switch element
515: load impedance
517: smoothing capacitor
519: compensate output signal
521: delay circuit
523: frequency controller
525: signal generator
527: phase-shift controller
529: frequency responds curve
531: phase shift curve
533:x axle
535a, 535b: voltage range
537: phase shift block
538: application circuit
539: error amplifier
540: flow process
541: amplifier circuit
543,545,547: step
600,604,610,614,618: circuit diagram
602,606,612,616,620: graph of a relation
603,613,617:ON state
607:OFF state
609: rise progressively
700,704,708,712,718: circuit diagram
702,706,710,714,720: graph of a relation
703,707,717:ON state
711: rise progressively
715:OFF state
800: chart
801: time of delay
803,807,811: Dead Time
810,820: timing diagram
ASIC: application-specific integrated circuit (ASIC)
Co: smoothing capacitor
Cp: parasitic capacitance
Cr: resonant mode capacitor
Cs: parasitic capacitance
DSP: digital signal processor
EA, EAF: error amplifier
EAD: amplifier circuit
Fea: output signal
Fref: reference voltage
F1, f2: frequency
IQ6, ICo, ILr, IRL: electric current
Lm: magnetization load inductance
Lr: resonant mode inductor
Q1 ~ Q4: switch element
Q5 ~ Q6: synchronous rectifier
Qd: switch
Rd, Rf: resistance
RL: load resistance
Ro: load impedance
RCD: resistor-capacitor diode
VAB: rectangular wave input voltage, supply voltage
Vea: error voltage signal
Vin: supply voltage
VL: load voltage
Vo: export (load) voltage
Vref: reference voltage
VD: supply voltage
S1: the first (mainly) control signal
S2: the second (secondary) control signal
T: transformer
T0 ~ t6: time point
ZVS: efficient zero voltage switch
α: phase shift (delay)
α 1 ~ α 3: phase angle (poor)
Embodiment
Some exemplary embodiments embodying feature & benefits of the present invention describe in detail in the explanation of back segment.Be understood that the present invention can have various changes in different modes, it does not depart from the scope of the present invention, and explanation wherein and accompanying drawing are in itself when the use explained, and is not used to limit the present invention.
Fig. 2 is the block diagram that the resonant converter assembly of example embodiment of the present invention has a phase shift block (phaseshiftmodule), in order to regulate the DC current gain (DCgain) of resonant converter.Resonant converter 101 is a power supply changeover device pattern, for carrying out in the electronic equipment of high-effect electrical source exchange in circuit.When being made up of inductor-capacitor (L-C) network of various assembly, resonant converter 101 uses circuit capacitance and inductance to go to mould and be wholely driven in the electric current of given switch element or the waveform of voltage, this switch element can be as elements such as Mosfet, IGBT.
Synchronous rectifier (SRs), is generally used for efficient operable transducer, if resonant converter 101 grade is to reduce the loss in circuit.Namely Fig. 3 discloses the better of synchronous rectifier 201 and 203 that the present invention is applied to resonant converter Circuits System and practices.
As mentioned above, synchronous rectifier 201 and 203 includes diodes in parallel to transistor typically, namely as a power metal oxide semiconductor field-effect transistor (powerMOSFET).
As shown in each embodiment, resonant converter 101 also comprises one or more main switch elements, if 201a-b (Q2, Q3) and these switch elements of 203a-b (Q1, Q4) can be MOSFET, BJT or IGBT etc.This main switch element driven by primary drive 103, this synchronous rectifier then driven by secondary driver 105, and drive the secondary driver 105 of this synchronous rectifier and drive nothing between the primary drive 103 of this resonant converter 101 main switch element to postpone, make this synchronous rectifier 201 and 203 and corresponding main switch element (as shown in Figure 3, as switch element 201a, during 201b work, synchronous rectifier 201 namely corresponding to synchronous rectifier 201 element that works is 201a-201b, in like manner, corresponding to synchronous rectifier 203 element is 203a-203b) and be activated at one time.
Under this kind of control mode, in synchronous rectifier 201 and 203 is by circuit secondary driver 105 open before, electric current first can flow through the body diode of synchronous rectifier 201 and 203.For reduce produce because electric current flows through synchronous rectifiers body diode consume, synchronous rectifier 201 and 203 slightly can be driven the time of respective switch element 201a-b and 203a-b, to lower consume by the time that secondary driver 105 starts early than primary drive 103.So way introduces a phase difference (phasedifference) in synchronous rectifier 201 between 203 and corresponding main switch element 201a-b and 203a-b.This synchronous rectifier now early than its corresponding main switch element with one diode current flow time/phase place (phase place corresponding to the time of the body diode mainly conducting of switch element corresponding to it of this synchronous rectifier or this period) difference and be activated in advance.
The resonant converter application circuit system 100 of Fig. 2 includes a resonant converter 101, and its assembly has a phase shift block 107.As shown in the figure, this phase shift block 107 is electrically connected to the primary drive 103 of this resonant converter 101 switch element and the secondary driver 105 of this resonant converter 101 synchronous rectifier.When operating, this phase shift block 107 can produce a control signal S1, in order to control or to change the drive singal exported by this primary drive 103; And produce a control signal S2, in order to control or to change the drive singal exported by this secondary driver 105.This phase shift block 107 by the phase shift angle between regulable control signal, and then can change DC current gain.The definition of phase shift angle is and starts this synchronous rectifier 201 and 203 (being driven by secondary driver 105), and starts the phase difference of its corresponding main switch element 201a-b and 203a-b (being driven by primary drive 103).By regulating this phase difference, the DC current gain of this resonant converter can effectively reduce and/or along with frequency monotonicity maintain.
Fig. 3 discloses the LLC serial-resonant converter circuit schematic diagram of example embodiment of the present invention, and it controls direct current by switching frequency and increases electric gain.Although resonant converter can be reached by various execution mode, its operation is all identical in essence and summation is as follows.According to provided voltage Vin, driving switch element 201a-b or 203a-b produced the square pulse wave of one voltage VAB.This square wave voltage VAB is then applied in a resonant circuit, is one in the present embodiment and has the LLC resonant circuit (resonanttankcircuit) that inductance is connected with capacitances in series.Energy is then passed to a transformer T211 first siding ring by this resonant circuit and produces voltage VL219 above, and its transformer turn ratio 229 is n: 1.
For example, this resonant converter 101 circuit leftmost side has a square wave voltage generator 225a in the element of dotted line 225 left.This square baud generator 225a comprises main switch element 201a-b and 203a-b, in order to produce a rectangular wave input voltage VAB217 to this LLC resonant network 235.This resonant converter 101 circuit rightmost side comprises a rectifier network 227a in dotted line 227 right element (transformer load).This rectifier network 227a produces a direct voltage output Vo by rectification in the AC current of transformer T211 second siding ring gained.In addition, this rectifier network 227a also comprises synchronous rectifier 201 and 203.Actual practicing can utilize full-bridge type and/or all-wave resonant circuit, and other certainly also can be used to practice.
For example, when switch element 201a-b conducting, secondary driver 105 starts synchronous rectifier 201, and transformer T211 second siding ring first half 215a passes through synchronous rectifier 201 to load transfer energy.Similarly, during the negative sense input half period, the Lower Half 215b conducting of this second siding ring, to send the negative sense half period to this load, therefore needs the action of this synchronous rectifier 203 to flow through this coil and corresponding switch element 203a-b to allow electric current.
In one embodiment, this rectifier network 227a also comprises an output capacitor (smoothingcapacitor) Co and load resistance RL.
The resonant circuit of this resonant network 235 is named, the resonant capacitance Cr209 be namely connected in series with a resonant inductance Lr205 and the magnetizing inductance Lm207 of transformer T211 for three assemblies comprising LLC circuit.This resonant circuit is then connected to the first siding ring 213 of transformer T211 again, and second siding ring 215a and 215b of this transformer T211 is then connected to this synchronous rectifier 201 and 203 simultaneously.
In this embodiment, this resonant converter 101, comprise a rectangular wave generator 225a and main switch element 201a-b and 203a-b matched, a resonant network 235 and various inductance thereof and electric capacity and a rectifier network 227a, can be applicable to the Power convert etc. of ac/dc or DC-DC.But resonant converter 101 also can be used for AC/DC conversion, DC/DC conversion or high frequency electric source conversion.Moreover, this resonant converter 101 also can practice according to other known change-over circuit assembly and framework, comprises but is not restricted to series resonance, parallel resonance and/or series parallel resonance, LCC frameworks and so on etc., this phase shift block 107 visible is applicable to any circuit structure by it.
In one embodiment, this phase shift block more comprises a feedback circuit, a voltage controlled oscillator (voltage-controlledoscillator, VCO), and a phase-shift controller.
Fig. 4 discloses the time-histories schematic diagram of the primary drive of resonant converter in example embodiment of the present invention and the phase shifting angle (phaseshiftdifferential) of secondary driver.For example, time-histories schematic diagram 300 is the time-histories schematic diagram of the LLC resonant converter of Fig. 3.The voltage VAB being supplied to resonant converter 101 resonant network 235 is a rectangular wave input 301, and its voltage changes by the occasion of replacing to negative value.As previously mentioned, be used for driving switch element 201a-b and 203a-b by primary drive pulse 305 and 307, secondary driver pulse 303 then provides drive singal to synchronous rectifier 203 (drive singal of synchronous rectifier 201 is not disclosed in Fig. 4).In typical operation, the primary drive pulse 305 of secondary (synchronous rectifier) driver pulse 303 and correspondence does not have phase retardation.Synchronous rectifier 201 and 203 is simultaneously open-minded with corresponding main switch.
For example, this primary drive pulse 305 is opened in almost identical time point such as t2, t5 etc. with corresponding synchronous rectifier driver pulse 303, after continuing for some time then in time point as t3, t6 are closed; In t2, the t5 moment, VAB301 rises to its peak value and starts to decline at t3, t6 moment VAB.Therefore, this relation discloses Fig. 3 synchronous rectifier Q6203 and is activated in almost identical time point with corresponding main switch element 203a and 203b (Q1 and Q4), and occurs without delay phenomenon.Similarly, synchronous rectifier Q5201 is unlocked in almost identical time point with corresponding switch element 201a and 201b (Q2 and Q3), and it corresponds to the pulse 307 that time point t0 and t4 occurs.And switch element 201a and 201b (Q2 and Q3) pulse 307 close and between the main driving pulse 305 of switch element 203a and 203b (Q1 and Q4) starts, the phase place (angle) of its correspondence or time-histories can be expressed as α 1=(phase angle of time point t1)-0 ° (Q1 (Q4) of time point t2 starts phase angle).Because during this period of time, be the Dead Time on a brachium pontis between complementary switch Q1 (Q4) and Q3 (Q2), so α 1 is called as Dead Time phase angle.And Q1 (Q4) and Q3 (Q2) are complementary switch, then angle [alpha] 2=-180 °=(phase angle of time point t0)-(phase angle of time point t2), α 3=180 °=(phase angle of time point t4)-(phase angle of time point t2).
When the signal of secondary driver 105 is comparatively early comparatively early opened relative to the signal of primary drive 103, a new phase shifting angle α is imported into.For example, this new phase shifting angle α can be expressed as: be engraved in the difference in phase place or time-histories during the opening of the secondary driving pulse 303 of this synchronous rectifier 203 and the corresponding main driving pulse 305 of corresponding switch element 203a-203b, wherein this secondary driving pulse 303 is signable for pulse 303a in the diagram, and it to be opened but not open-minded in time point t2 in time point t ' comparatively early.Its relational expression is α=(phase angle of time point t ')-0 ° (phase angle of time point t2).Because this LLC resonant converter 101 is one full-bridge type/all-wave series resonant converter, this DC current gain (nVo/Vin) determined by switching frequency usually.But this phase shift angle also can be used for DC current gain to be regulated, and this phase shift degree exceedes or be less than this dead band time phase angle degree α 1 all can affect this gain further.The scope of this phase shift α (angle) can by-180 degree (comprising) to 180 degree (comprising), and in one embodiment, this range can by-180 degree to 0 degree (comprising).For example, this range can be spent by Dead Time phase angle to 0, more particularly, this range by Dead Time phase angle to diode current flow phase angle.Aforesaid specified scope all comprises endpoint value.Moreover in other embodiments, this endpoint value can be approximation, but comprises in essence, only otherwise affect the identity property of its operation.
Fig. 5 A discloses the DC current gain curve that example embodiment of the present invention is simulated with the means of compensate for parasitic capacitance.For example, required monotonicity DC current gain feature is disclosed in the curve 411 in figure.In this case, the DC current gain of light load operation is reduced to null value step by step along with the increase of frequency of operation.This also unlike the DC current gain curve 1009 (referring to Fig. 1 C) of tool charge pump, easily has different frequency of operation with value relatively.It should be noted that result that DC current gain curve 1009 is responded be due to import an additional electric capacity (as parasitic capacitance) and make output loading voltage Vo and non-self null value slowly smooth-going (as linear forms) be promoted to caused by given voltage levvl.Therefore, be necessary in order to the means solved or improve charge pump effect in this resonant converter.
For reaching aforesaid effect, a phase shift block 107 of relatively this resonant converter 101 assembly as disclosed in Figure 2 can make this DC current gain fall when the HF switch application of this resonant converter to subtract.About the enforcement of this phase shift block 107 and application by the circuit diagram 500 that is further disclosed in Fig. 6 A.But in order to clearly demonstrate, Fig. 5 B and Fig. 5 C also discloses the graph of a relation between the phase shift of resonant converter in various embodiments of the present invention and voltage gain.In figure 5b, standardization DC current gain curve 413 is in changing along with the different phase shift phase angle value of x-axis 415.In block 417, the enlarged drawing of gain curve 413 is then disclosed in Fig. 5 C, and it is presented in the limited field of phase angle excursion between-30 degree to+30 degree.The value of this gain curve 413 continues to maintain null value, until phase angle value is to the position of A point and Dead Time phase angle, it is the Dead Time of the primary drive of complementary switch in a corresponding switch bridge (switchbridge) also.After phase place increases above A point, DC current gain just exceedes null value and increases in monotonicity.Therefore, DC current gain can be fallen and be reduced to null value, if when the B point of phase angle in figure (phase shifting angle is 0 degree) moves to A point.
Fig. 6 A to disclose in example embodiment of the present invention resonant converter combo one phase shift block to regulate the circuit diagram of DC current gain.For example, the phase shift block 503 of this application circuit 500 receives the output voltage Vo of this resonant converter 501 as input.The voltage Vo of input is sent to an error amplifier EA505 of this phase shift block, and subsequently again compared with reference voltage (voltage as corresponding required voltage gain).Error voltage signal Vea is produced after two signals that this error amplifier 505 receives.
This error voltage signal Vea is sent to a resonant converter control module 511 as input.This resonant converter control module 511 (the control chip MC33067 as ONsemi) can be efficient zero voltage switch (ZVS) controller, as the application of off-line, ac/dc or DC/DC conversion, it utilizes frequency to adjust the fixing shut-in time or fixing Dead Time carrys out control circuit.The error voltage signal Vea of application can be used as the input of a frequency adjustment control input end 511a of this resonant converter control module 511, in order to adjust switching frequency.Resistance Rf509 then can in order to blanketing frequency response accordingly.Error voltage signal Vea is also sent to a Power MOSFET input 511b of resonant converter control module 511, in order to the Dead Time of regulating circuit breaker in middle element.Vea inputs to 511b by resistance Rd507 and switch element 513.Resistance Ro515 and electric capacity 517 are for filtering.Such frequency adjustment and Dead Time regulate be inter-related.Therefore the adjustment of frequency adjustment and phase shifting angle is also mutually related.
Like this, an output signal 519 produced by this resonant converter control module 511, and this signal will be applicable to the secondary drive singal S2 generating resonant converter 501.S2 can produce the time of delay (can how second (nanoseconds) expression) of a specified rate again between this secondary drive singal S2 and this main drive singal S1 by a delay circuit 521 (can form for resistor-capacitor circuit-diode (RCD) as shown in FIG.).This postpones, and more may be interpreted as phase delay alpha.As shown in the figure, signal S1 and S2 is passed to the primary drive 103 of resonant converter and secondary driver 105 more respectively to drive switch element and the synchronous rectifier of this resonant converter 501 respectively.
Fig. 6 B and Fig. 6 C disclose the application schematic diagram with FREQUENCY CONTROL and the separate phase shift block of phase shifting control in various embodiments of the present invention.For example, Fig. 6 B discloses a phase-shift controller 527, and it provides main switch element Q1 and the Q4 and switch element Q2 and Q3 that output signal to this resonant converter 101.Again, a frequency controller 523 provides output signal with the frequency regulating this resonant converter 101.This phase-shift controller and this frequency controller are mutually independently.Independently frequency controller 523 and phase-shift controller 527 are connected alternately by a signal generator 525.In this assembly, this signal generator 525 is received from the signal of frequency controller 523 output and provides drive singal to synchronous rectifier Q6 and Q5 of this resonant converter 101.Therefore this signal generator 525 produces signal S1 and S2 respectively and controls the main of this resonant converter 101 and secondary driver according to this.Again, be supplied to frequency controller 523 and phase-shift controller 527 in the fed-back output voltage Vo of the rectifier network 227a of this resonant converter 101, and each controller operates its respective function based on its input.
Fig. 6 C disclose example embodiment medium frequency of the present invention control and phase shifting control mutual independent time, frequency and phase shift accordingly result graph of a relation.For example, the voltage feedback signal that represents according to x-axis 533 separately of frequency curve 529 and phase shift curve 531 and changing.Along with feedback voltage signal increases, this frequency curve 529 falls progressively between voltage range 535a and 535b, and is down to low frequency (as 100K) by high frequency (as 400K).Meanwhile, phase shift curve 531 starts to rise progressively, but then all maintains with scope thereafter between identical voltage range 535a and 535b and fix.It should be noted that frequency and the independence of phase shift control with Fig. 6 A be that frequency and phase shift are mutually related and control to produce identical control result.
Other different application assembly of circuit 500, comprise and utilize additional error amplifier to remove regulation output voltage, be also contained in the scope of this example embodiment.For example, Fig. 6 D discloses resonant converter in example embodiment of the present invention and uses different error amplifiers to regulate the circuit diagram of the phase shift block institute assembly of DC current gain.This assembly corresponding diagram 6B and the method shown in Fig. 6 C, wherein FREQUENCY CONTROL and phase shifting control are performed by different controllers and in circuit, obtain identical DC current gain regulating effect.In the present embodiment, circuit 538 is the same with circuit 500 is furnished with a phase shift block 537, and itself and resonant converter 501 assembly are to receive a feedback signal Vo.Output voltage signal (Vo) sequentially inputs to an error amplifier EAF539 of this phase shift block 537 to save operation (work) frequency adjusting this circuit 535 together with reference voltage Vref given with.The output Fea of this error amplifier EAF539, inputs to this resonant converter control module 511 by a resistance Rf.Be with circuit 500 difference, this output signal Fea is also provided to amplifier circuit EAD541, and it is a frequency feedback amplifier, in order to the control lag time.The output signal Dea of amplifier EAD is coupled to a switch Qd again, has signal S1 and S2 of phase shift and the phase shifting angle between conditioning signal between providing according to this mutually.It should be noted that this circuit 538, when frequency adjustment lower than some values as 400K time, output signal Fea is greater than reference value Fref, thus makes amplifier EAD541 be that negative sense is saturated.Like this, the time of delay between signal S1 and S2 and the phase shifting angle caused by it then reduce to minimum value.
And when the output signal Fea of amplifier EAF539 is less than reference value Fref, this amplifier EAD541 outputs signal Dea by according to the conducting degree of the difference control switch Qd between Fea and Fref thus control lag extent time phase.
Fig. 6 E discloses the flow chart regulating resonant converter DC current gain in example embodiment of the present invention.In order to the flow process 540 of activation gain-adjusted, frequency adjustment and Power MOSFET, for example, can the circuit 500 of Fig. 6 A or the circuit 538 of Fig. 6 D perform.In the step 543 of this flow process 540, phase shift block 503 produces one first control signal S1 to control the primary drive of resonant converter 501.In step 545, this phase shift block 503 produces one second control signal S2 to control the secondary driver of resonant converter 501.And this phase shift block 503, in step 547, then feedback assembly (as shown in circuit 500 and 538) of this resonant converter 501 collaborative is run, and is to control a DC current gain of this resonant converter at least partly based on the phase shifting angle of the first control signal and the second control signal.This phase shifting angle or difference can affect the various characteristic of required yield value in order to compensating gain condition and/or other.
In Fig. 7 A to Fig. 7 E announcement various embodiments of the present invention, resonant converter is in time according to the behavior timing diagram of phase shifting angle between main and secondary driver, and wherein this phase shifting angle is less than Dead Time phase angle.For example, the upper section of each accompanying drawing shows electric current in separating preset time in the trend of resonant converter circuit; In the inferior portion of each accompanying drawing is then presented at and separates preset time, the phase shifting angle occurred between the main of resonant converter and secondary driver, particularly demonstrates the oscillogram of each electric current in circuit.
Fig. 7 A is described as follows:
Between time point t0 to t1:
1. in synchronous rectifier Q6 is by circuit 600, secondary driver starts to ON state 603, and as shown in graph of a relation 602, namely synchronous rectifier Q6 is forced On current, as shown in circuit in figure 600.
2. because electric current I Lr flow to switch element Q3 from switch element Q2 through resonant circuit, the electric current I Q6 of synchronous rectifier Q6 then flow to source electrode by drain electrode, causes electric capacity Co to produce discharging current ICo, as illustrated as shown in 602.
Fig. 7 B is described as follows:
Between time point t1 to t2:
1. after switch element Q2 and Q3 is closed to OFF state 607, as illustrated as shown in 606, the electric current I Lr flowing through resonant inductance flows through the body diode of switch element Q1 and Q4, as illustrated waveform 609 as shown in 606, putting electric current I Lr and starting to rise at this moment.
2., because the electric current I Lr flowing through resonant inductance is identical with direction between time-histories t0 ~ t1, electric current I Q6 still flow to source terminal by drain electrode end, and electric capacity Co is discharged, as illustrated as shown in 606.
Fig. 7 C is described as follows:
Between time point t2 to t2 ':
1., as shown in diagram 612, switch element Q1 and Q4 is now in a conducting state 613, and the electric current I Lr flowing through resonant inductance flows through switch element Q1 and Q4, as illustrated as shown in 612.
2. electric current I Lr does not change direction in section at this moment, and the electric current I Q6 therefore flowing through synchronous rectifier Q6 still flows to source electrode from drain electrode, and this electric capacity Co is discharged, as illustrated as shown in 612.
Fig. 7 D is described as follows:
Between time point t2 ' to t3:
1., after time point t2 ', the electric current I Lr flowing through this resonant inductance changes its direction, as illustrated as shown in 616.
2. electric current I Q6 is existing flow to drain electrode from source electrode, and this electric capacity Co is charged, as illustrated as shown in 616.
3., after time point t3, synchronous rectifier Q5 then transfers unlatching ON state 617 to as shown in diagram 616, and its operational process is identical with synchronous rectifier Q6.
Fig. 7 E sums up Fig. 7 A to Fig. 7 D and is described as follows:
During time point t0 ~ t3:
1. the current value flowing through electric capacity and load resistance is summed up ICo+IRL and is equaled to flow through respectively the electric current I Q5 of synchronous rectifier Q5 and Q6 and the current value summation of IQ6.
2. the current waveform produced by current value summation ICo+IRL, as illustrated as shown in 620, it is charging current that the waveform table above x-axis is shown in this time point; And person is discharging current below x-axis.
3., as shown in diagram 620, average discharge current is greater than charging current.Therefore, can reach a compensating effect makes load voltage Vo be discharged to required magnitude of voltage.This regulates the method for resonant converter voltage gain to be reached according to the adjustment of phase delay, Dead Time and/or switching frequency by a phase shift block 107.
In Fig. 8 A to Fig. 8 E announcement various embodiments of the present invention, resonant converter is in time according to the behavior timing diagram of phase shifting angle between main and secondary driver, and wherein between this main and secondary driver, phase shifting angle is greater than a Dead Time phase angle, and is less than 0 degree.Below to be same as the process description discussion of Fig. 7 A-Fig. 7 E.
Fig. 8 A is described as follows:
Between time point t0 to t1:
1. be activated to ON state 703 at synchronous rectifier Q6, as shown in graph of a relation 702, namely synchronous rectifier Q6 is forced On current IQ6, as illustrated as shown in 700.
2. because electric current I Lr to flow to the body diode of switch element Q1 from the body diode of switch element Q4, as illustrated as shown in 700, the electric current I Q6 of synchronous rectifier Q6 flow to source electrode by drain electrode, causes electric capacity Co to produce discharging current ICo, as illustrated as shown in 702.
Fig. 8 B is described as follows:
Between time point t1 to t1 ':
1. being opened into ON state at switch element Q1 and Q4, causing the electric current I Lr flowing through resonant inductance to flow through switch element Q1 and Q4, as illustrated as shown in 706.
2. because the electric current I Lr that at this moment, section flows through resonant inductance does not change direction, so the electric current I Q6 of synchronous rectifier Q6 still flow to source terminal by drain electrode end, and make electric capacity Co produce discharging current ICo, as illustrated as shown in 706.
Fig. 8 C is described as follows:
Between time point t1 ' to t2:
1. the electric current I Lr flowing through resonant inductor changes its direction afterwards at time point t1 ', as illustrated as shown in 710.
2. the electric current I Q6 flowing through synchronous rectifier Q6 will flow to drain electrode from source electrode, and this electric capacity Co is charged, waveform 711 as shown in the drawings.
Fig. 8 D is described as follows:
Between time point t2 to t3:
1., as shown in diagram 714, after switch element Q1 and Q4 is closed to OFF state 715, the electric current I Lr flowing through this resonant inductance flows through the body diode of switch element Q2 and Q3.
2. owing to flowing through the electric current I Lr of inductance as there is identical direction with electric current I Lr between time-histories t1 ' to t2, therefore flow through the electric current I Q6 of synchronous rectifier Q6 still from source and course extremely, and this electric capacity Co is charged, as illustrated those disclosed herein in 714.
3., after time point t3, synchronous rectifier Q5 transfers unlatching ON state 717 to, and its operational process is identical with synchronous rectifier Q6.
Fig. 8 E sums up Fig. 8 A to Fig. 8 D and is described as follows:
During time point t0 ~ t3:
1. the current value flowing through electric capacity and load resistance is summed up ICo+IRL and is equaled to flow through respectively the electric current I Q5 of synchronous rectifier Q5 and Q6 and the current value sum of IQ6.
2. the current waveform produced by current value summation ICo+IRL, as illustrated as shown in 720, it is charging current that the waveform table above x-axis is shown in this time point; And person is discharging current below x-axis.
3., as shown in diagram 720, average charge stream is greater than discharging current.Therefore, can reach a compensating effect makes load voltage Vo charge to suitable level value.This regulates the method for resonant converter voltage gain to be reached according to the adjustment of phase delay, Dead Time and/or switching frequency by a phase shift block 107.
The phase shift that Fig. 9 discloses a resonant converter in example embodiment of the present invention regulates strain stress relation figure, regulates phase shift angle by regulating Dead Time thus affects DC current gain.Wherein the delay circuit 521 Received signal strength S2 of this phase shift block 503 also produces with the dephased input signal S1 of S2 to affect voltage gain.For example, the delay between the S1S2 imported by delay circuit 521 be a fixed value as can be 150ns, this value does not change with circuit work frequency, namely as shown in the horizontal linear 801 in chart 800.In timing diagram 810 and 820, announcement RCD delay (as 100K) under high frequency (if height is to 500K) and low frequency situation is all identical respectively.Differently, the Dead Time characteristic of this resonant converter circuit tapers off in the scope of frequency increments, namely as shown in the straight line 803 in chart 800.Timing diagram 810 describes, and in frequency applications, this Dead Time 807 is less than RCD time of delay 801 (as 50ns < 150ns).Timing diagram 820 describes, and in low frequency applications, this Dead Time 811 is greater than RCD time of delay 801 (as 500ns > 150ns).
Change in frequency of operation, time of delay is fixed, under Dead Time between the switch element Q1 (Q4) of resonant circuit and Q2 (Q3) then carries out the situation regulated shown in foundation chart 800, the phase shift angle between synchronous rectifier Q5 and Q6 of main switch and correspondence is also conditioned.Again, under same frequency is responded, when Dead Time becomes to being similar to time of delay, crosspoint C is equivalent to.An advantage of the present invention is, by the assembling structure of Fig. 6 A or Fig. 6 D, the frequency of resonant converter 101 and phase shift angle can be regulated and controled simultaneously.
The demonstration system that the present invention discusses and technology can provide the short-cut method of the key property of adjustment one resonant converter circuit, on design ap-plication, especially more utilize synchronous rectifier to carry out start switch.By assembly this resonant circuit 101 and a phase shift block 107, this resonant converter is its voltage gain characteristic adjustable then.
Moreover this system can process when light load operation according to the charge pump effect of parasitic capacitance application generation or its inner effect produced.So example embodiment of the present invention can carry out phase shifting control and obtain lower voltage gain and meet design within the scope of change frequency considering.Technological means disclosed above in order to lower voltage or the current stress of each element in circuit, and can promote stability and the reliability of circuit.
Aforesaid advantage all can be overlapped in the assembly for resonant converter, comprises as LLC, LCC, parallel connection, series connection, series parallel resonance and combination thereof, but is not limited to this.Moreover, describe at this and control resonant converter DC current gain to increase the flow process of power supply service efficiency in circuit, more further by software, hardware, firmware or can implement in conjunction with software and/or firmware and/or hardware.Such as, Digital Signal Processing controls can in order to carry out the adjustment of frequency and phase place, whereby the frequency signal that produces in phase shift block 107/503 by comparing triangular signal and reference value signal of this driver and/or controller and/or phase shifting angle signal.By the method, frequency can be regulated by triangular wave count value, and time of delay then changes by changing digital signal processor (digitalsignalprocessor, DSP) the chip internal reference value that compares.And also can application specific integrated circuit (ApplicationSpecificIntegratedCircuit, ASIC) and field programmable gate array (FieldProgrammableGateArray, FPGA) etc. reach in the program described in this.
The technology of the present invention has practicality, novelty and progressive, files an application in accordance with the law.Even if the present invention has been described in detail by the above embodiments and can have been appointed by those skilled in the art and execute craftsman and to think and for modifying as all, but de-as the scope of attached claim institute for protecting.
Claims (22)
1. be applied to an adjustment gain method for resonant converter, it comprises step:
Produce one first control signal, in order to control a primary drive of a resonant converter; And
Produce one second control signal, in order to control this resonant converter one secondary driver, wherein this first control signal and this second control signal have a phase shifting angle, control this resonant converter by this phase shifting angle and realize a stable and DC current gain for monotone variation, this phase shifting angle is start by the synchronous rectifier of this secondary driver drives and the phase difference of the main switch element driven by primary drive starting its correspondence.
2. regulate gain method as claimed in claim 1, wherein the scope of this phase shifting angle is for being more than or equal to-180 degree to being less than or equal to+180 degree.
3. regulate gain method as claimed in claim 1, wherein the scope of this phase shifting angle is for being more than or equal to-180 degree to being less than or equal to 0 degree.
4. regulate gain method as claimed in claim 1, wherein this primary drive drives multiple main switch element, and the method more comprises step:
This main switch element one Dead Time given.
5. regulate gain method as claimed in claim 4, wherein the scope of this phase shifting angle is for being more than or equal to Dead Time phase angle to being less than or equal to 0 degree.
6. regulate gain method as claimed in claim 4, wherein the scope of this phase shifting angle is for being more than or equal to Dead Time phase angle to being less than or equal to diode current flow phase angle.
7. regulate gain method as claimed in claim 4, more comprise step:
Receive an output signal conduct feedback of this resonant converter, to produce this first control signal and this second control signal.
8. be applied to an adjustment gain apparatus for resonant converter, it comprises:
One delay circuit; And
One control module, is coupled to this delay circuit, and provides and export this delay circuit to, in order to produce one first control signal to control a primary drive of this resonant converter; And produce one second control signal with control this resonant converter one secondary driver;
Wherein this first control signal and this second control signal have a phase shifting angle, control this resonant converter by this phase shifting angle and realize a stable and DC current gain for monotone variation, this phase shifting angle is start by the synchronous rectifier of this secondary driver drives and the phase difference of the main switch element driven by primary drive starting its correspondence.
9. regulate gain apparatus as claimed in claim 8, wherein the scope of this phase shifting angle is for being more than or equal to-180 degree to being less than or equal to 0 degree.
10. regulate gain apparatus as claimed in claim 8, wherein the scope of this phase shifting angle is for being more than or equal to a Dead Time phase angle to being less than or equal to 0 degree.
11. regulate gain apparatus as claimed in claim 8, and wherein the scope of this phase shifting angle is for being more than or equal to a Dead Time phase angle to being less than or equal to a diode current flow phase angle.
12. regulate gain apparatus as claimed in claim 8, more comprise:
One error amplifying circuit, receives the output voltage of this resonant converter, and produces the input of an error voltage signal as this control module to control a frequency of this resonant converter.
13. regulate gain apparatus as claimed in claim 12, wherein:
This error voltage signal is more in order to control a Dead Time of this resonant converter.
14. regulate gain apparatus as claimed in claim 12, more comprise:
One frequency feedback amplifying circuit, is connected with this error amplifying circuit and this delay circuit, receives this error voltage signal to produce an output signal in order to control this phase shifting angle.
15. regulate gain apparatus as claimed in claim 8, and wherein this control module produces this first control signal and this second control signal according to an output signal of this resonant converter.
16. regulate gain apparatus as claimed in claim 8, and wherein this control module produces this first control signal and this second control signal according to the signal of a reaction output voltage of this resonant converter.
17. 1 regulate gain apparatus, and it comprises:
One resonant converter, has a primary drive and a secondary driver; And
One phase shift block, is coupled to this primary drive and this secondary driver, and this phase shift block produces one first control signal to control this primary drive and to produce one second control signal to control this secondary driver;
Wherein this first control signal and this second control signal have a phase shifting angle, control this resonant converter by this phase shifting angle and realize a stable and DC current gain for monotone variation, this phase shifting angle is start by the synchronous rectifier of this secondary driver drives and the phase difference of the main switch element driven by primary drive starting its correspondence.
18. regulate gain apparatus as claimed in claim 17, and wherein the scope of this phase shifting angle is for being more than or equal to-180 degree to being less than or equal to 0 degree.
19. regulate gain apparatus as claimed in claim 17, and wherein the scope of this phase shifting angle is for being more than or equal to a Dead Time phase angle to being less than or equal to 0 degree.
20. regulate gain apparatus as claimed in claim 17, and wherein the scope of this phase shifting angle is for being more than or equal to a Dead Time phase angle to being less than or equal to a diode current flow phase angle.
21. regulate gain apparatus as claimed in claim 17, and wherein this phase shift block more receives an output signal of this resonant converter as feedback, to produce this first control signal and this second control signal.
22. regulate gain apparatus as claimed in claim 21, and wherein this phase shift block comprises:
One error amplifying circuit, receives the output voltage of this resonant converter, and produces an error voltage signal; And
One resonant converter control module, is coupled to this error amplifying circuit, and receives this error voltage signal, in order to export this first control signal and the second control signal.
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