CN101615861A - A high-frequency PWM rectification and inverter integrated device and its control method - Google Patents

Abstract

A high-frequency PWM rectification and inverter integrated device and a control method thereof belong to the technical field of power electronics. Including main circuit unit, drive and power amplification unit, PWM unit, central control unit, HMI unit, voltage and current transformer, power parameter measurement unit, power supply unit. The main circuit unit is connected with the voltage and current transformer, drive and power amplification unit; the drive and power amplification unit are respectively connected with the power supply unit and PWM; the PWM is respectively connected with the power supply unit and the central control unit; the central control unit is respectively connected with the power parameter measurement unit and HMI unit connected. During the start-up process of the permanent magnet synchronous generator, the DC current is converted into alternating current. When the permanent magnet synchronous generator is running in a steady state, the high-frequency alternating current generated by the permanent magnet synchronous generator is converted into direct current to realize bidirectional energy flow.

Description

A kind of high-frequency PWM commutation inversion integrated apparatus and control method thereof

Technical field

The invention belongs to electric and electronic technical field, particularly a kind of high-frequency PWM commutation inversion integrated apparatus and control method thereof.

Background technology

The micro-gas-turbine machine power generating system is a kind of new distribution type electricity generation system that develops rapidly in recent years, its output power range generally at 20KW between the 500KW, have cleaning, reliable, multipurpose and realize advantage such as CCHP.In addition, can also be as portable high-performance power supply, for the using electricity system under various particular surroundingss, the specific condition provides high-performance, stable, reliable electric power energy.During system works, by power turbine, drive rotor of permanent-magnetic power generator high speed rotating on air bearing, produce high-frequency alternating current, carry out controlled frequency conversion through power converter, the alternating current of customer requirements is satisfied in output.

Conventional rectification link generally adopts diode not control rectification or Thyristor Controlled rectification.

Diode rectifier circuit is simple, economic and reliable.Therefore its application of eighties of last century is very extensive, but the widely-used problem of also having brought the following aspects of this rectifier:

(1) diode rectification can make the current on line side waveform seriously distort, and causes power factor lower, and highest power factor only may be about 0.8.The consumption meeting of a large amount of reactive powers brings extra burden to electrical network, has not only increased the loss of transmission line, and has seriously influenced power supply quality.

(2) frequency spectrum of diode rectifier input current is analyzed, found to contain abundant low-order harmonic electric current in the input current.

(3) for AC variable-frequency speed regulation system, because the unilateal conduction performance of diode, the regenerated energy of motor braking can't feed back to electrical network.For the safe operation of installing, this part energy must consume by certain approach.In medium and small capacity system, generally adopt the dynamic braking mode, promptly by the built-in or method that adds brake resistance with power consumption in Power Resistor, realize the four quadrant running of motor.Though this method is simple, and following shortcoming is arranged: the waste energy, system effectiveness is low; Resistance heating is serious, influences other part operate as normal of system; Simple dynamic braking can not in time suppress the pump up voltage that fast braking produces, and has limited the raising of performance.

Summary of the invention

For solving the problems of the technologies described above, the invention provides a kind of high-frequency PWM commutation inversion integrated apparatus and control method thereof that can realize the electric energy two-way flow.

Apparatus of the present invention comprise main circuit unit, driving and power amplification unit, PWM control unit, central control unit, human and machine interface unit (HMI), voltage current transformer, electric parameter measurement unit, power subsystem composition.Wherein, main circuit unit comprises device for power switching IGBT, inverse parallel diode, absorber and DC filter capacitors; The PWM control unit comprises digital signal processor (DSP), programmable logic device (CPLD) and RS485 interface circuit, wherein includes the burst pulse limiting module among the CPLD, is used for fault is handled; Central control unit comprises programmable logic controller (PLC) (PLC) unit.

Main circuit unit links to each other with voltage current transformer, driving and power amplification unit; Drive and link to each other with the PWM control unit with power subsystem respectively with power amplification unit; The PWM control unit links to each other with central control unit with power subsystem respectively; Central control unit links to each other with human and machine interface unit (HMI) with the electric parameter measurement unit respectively.

Principle: in the electrokinetic process of permanent magnet synchronous machine, this device is operated in inverter mode, and dc inverter is become alternating current, drives micro fuel engine and starts operation; In the power generation process of permanent magnet synchronous machine, this device is operated in rectification state, and alternating current is reverse into direct current, uses for the user.

A kind of control method of high-frequency PWM commutation inversion integrated apparatus realizes in dsp processor, as shown in Figure 6, may further comprise the steps:

Step 1. starter motor also quickens;

The transmitting supervisory tach signal is given central control unit during step 2.10Hz, and wherein 10Hz is the inverter output frequency;

Whether step 3. detects motor and starts, if start then execution in step 4, otherwise execution in step 1;

The work of step 4. inverter pattern;

Whether step 5. judges rotating speed greater than 36000RPM, if be not more than then execution in step 4, otherwise execution in step 6;

The work of step 6. rectifier pattern;

Step 7. finishes.

The control method of described a kind of high-frequency PWM commutation inversion integrated apparatus, the inverter pattern work described in the step 4, as shown in Figure 7, step is as follows:

Step 1. beginning;

Step 2. judges whether starting command arrives, if arrive then execution in step 3, otherwise execution in step 2;

Step 3. inverter mode of operation: call the SVPWM algorithm;

Step 4. motor raising speed;

Step 5. judges whether to reach 600Hz, if reach then execution in step 6, otherwise execution in step 4, wherein 600Hz is the inverter output frequency;

The control of step 6. inverter constant speed;

Step 7. judges to cease and desist order whether arrive, if arrive then execution in step 8, otherwise execution in step 6;

Step 8. motor reduction of speed;

Step 9. judges whether to arrive to shut down changes, if reach then execution in step 10, otherwise execution in step 8;

Step 10. is shut down envelope pulse subroutine, execution in step 2.

Spatial vector pulse width modulation algorithm SVPWM algorithm described in the step 3 in the control method of inverter mode of operation work, as shown in Figure 8, step is as follows:

Step 1. beginning;

Step 2. pair AD sampling is handled;

Step 3. false judgment and processing;

Step 4. receives the set point of central control unit by SCI;

Step 5. is regularly uploaded sampled value;

Step 6. has judged whether interruption, if having then execution in step 7, otherwise execution in step 2;

Step 7. is removed and is interrupted indicating;

Step 8. is according to the parameters such as form of current frequency configuration SVPWM;

Step 9. is calculated S _a, S _b, S _c

Step 10. is called pwm pulse control method and burst pulse method for limiting;

Step 11. finishes.

The control method of described a kind of high-frequency PWM commutation inversion integrated apparatus, rectifier pattern work in the step 6, as shown in Figure 9, step is as follows:

Step 1.PWM Interrupt Process;

Step 2. is interrupted protection;

Step 3. is removed interrupt identification;

Step 4. direct voltage trace routine;

The sampling of step 5. alternating current side, synchronization signal detection is sampled, and calls the acquisition method of analogue data amount;

Step 6. filtering and data processing are called the filtering method of analogue data amount;

Whether step 7. judges motor speed greater than 36000RPM, if greater than execution in step 8, otherwise execution in step 11;

Step 8. rectifier mode of operation: transfer sliding moding structure direct Power Control algorithm;

Step 9. is called pwm pulse control method and burst pulse method for limiting;

Step 10. program pointer adds 1;

Step 11. is returned.

The described accent sliding moding structure of step 8 direct Power Control algorithm in the work of rectifier mode of operation, as shown in Figure 10, step is as follows:

Step 1. beginning;

Step 2. is called the filtering method (electric current and voltage) of analogue data amount;

Step 3. is carried out abc to a β transformation calculations θ angle, and wherein θ represents the phase angle of voltage;

Step 4, calculating instantaneous power P and Q, the rated output factor;

Step 5. receives the set point reactive power set-point Q of central control unit by SCI ^*=0, calculate active power set-point P ^*: P ^*=Ck ²(V _Dc ^{* 2}-V _Dc ²)+V _Dc* I ₀, V wherein _Dc ^*Be meant rectification side dc voltage set-point, V _DcRectification side dc voltage actual value;

Step 6, timing are uploaded voltage, electric current, power, power factor to central control unit;

Step 7, judged whether interruption, then execution in step 7 has been arranged; Otherwise execution in step 2;

Step 8, removing are interrupted indicating;

Step 9. is according to P, Q, P ^*, Q ^*, the θ angle, the compute switch function S _pWith switch function S _qValue;

Step 10, compute switch function S _a, S _b, S _c

Step 11, call PWM and generate control method;

Step 12, end.

Switch function Sa, Sb, Sc in the described accent sliding moding structure direct Power Control method step 10 are defined as follows:

S _a=1 o'clock, power tube T in the accompanying drawing 1 ₁Connect T ₄Turn-off.S _a=0 o'clock, power tube T in the accompanying drawing 1 ₁Turn-off T ₄Connect.

S _b=1 o'clock, power tube T in the accompanying drawing 1 ₃Connect T ₆Turn-off.S _b=0 o'clock, power tube T in the accompanying drawing 1 ₃Turn-off T ₆Connect.

S _c=1 o'clock, power tube T in the accompanying drawing 1 ₅Connect T ₂Turn-off.S _c=0 o'clock, power tube T in the accompanying drawing 1 ₅Turn-off T ₂Connect.

(S _aS _bS _c) have 8 kinds of vectors: 000,111,100,110,010,011,001,101.

The universal control method of inversion and switching process comprises that acquisition method, the filtering method of analogue data amount, the pwm pulse of analogue data amount generate control method, DSP initial method, system initialization and self check control method, burst pulse method for limiting.

The acquisition method of analogue data amount as shown in Figure 11, step is as follows:

Step 1.AD handling procedure;

Step 2. judges whether to take place AD and interrupts, if take place then execution in step 3, if do not have then execution in step 7;

The clear AD interrupt identification of step 3.;

Step 4. pair sampled result is carried out Filtering Processing;

Step 5. sampling buffer data are carried out the one-period time-delay;

Step 6. is converted to actual value with filter value;

Step 7. is returned;

The filtering method of analogue data amount as shown in Figure 12, step is as follows:

The step 1.AD interruption of sampling;

Step 2. is put AD and is interrupted indicating;

Step 3. copies to the value of sampled result register in the buffering area;

Step 4. is carried out+1 operation and is refreshed in the internal clocking unit during pair with the 20K speedometer;

Step 5. sorting unit that resets, AD starts to prepare next time;

Step 6. is recovered environmental variance;

Step 7. is returned.

Pwm pulse generates control method as shown in Figure 13, and step is as follows:

Step 1. calculates 0;

Step 2. is determined sector number;

Step 3. is calculated pwm pulse time T 1 and T2;

Step 4. is provided with CMPR1, CMPR2, CMPR3;

Step 5. finishes.

DSP initialization flow process as shown in Figure 14, step is as follows:

Step 1. system initialization and self check subprogram;

The initialization of step 2. system clock;

Step 3. time-delay started the main circuit relay after one minute;

Step 4.AD sampling and filtering subprogram;

The initialization of step 5. digital I;

Step 6. is caught the CAPTURE initialization, promptly is to its next step initialization;

Step 7. electrical network phase acquisition, computation of Period and protection;

Step 8. timer and the initialization of PWM generating unit;

Step 9. is interrupted initialization;

Step 10. overall situation is opened interruption;

Step 11. is called rectification, inversion operating mode control method.

System initialization and self check flow process as shown in Figure 15, step is as follows:

Step 1. defconstant, variable;

Step 2. is provided with interruption, sub-interrupt vector table;

Step 3. initialization of register;

Step 4, Installed System Memory self check;

Step 5.I/O port initialization;

The initialization of step 6. task manager;

Step 7.ADC initialization;

Step 8.SCI initialization;

Step 9. finishes.

In system, the major function of CPLD is that fault is handled, and burst pulse restriction and last PWM waveform generation be not because special-purpose drive plate possesses the function of burst pulse restriction, so in hardware designs, the burst pulse limiting element should be arranged.

The determination methods of burst pulse: in the half switch periods instruction in the right side of calculating gained waveform, the action time of two vectors about judgement.As long as less than Tnarrow (burst pulse clock cycle), just there is burst pulse in one of them.

The removing method of burst pulse: when burst pulse produces, adopted 3 kinds of ways that it is eliminated successively, mainly as follows:

(a) if zero vector action time greater than 4Tnarrow, zero vector and rearranging then.

(b) if waveform opposite side vector action time greater than 2Tnarrow, then to opposite side travelling backwards Tnarrow.

(c) if reference vector drops on the overlapping region, then select another zone as with reference to the zone under the vector, then adopt above-mentioned a again, the method for b is carried out the elimination of burst pulse.

Burst pulse limits flow process as shown in Figure 16, and step is as follows:

Step 1, timer zero clearing;

Step 2, judge whether input is high, if be height then execution in step 3, otherwise execution in step 7;

Step 3, timer add 1;

Step 4, judge whether to reach binding hours, if reach then execution in step 5, if do not reach then execution in step 6;

Step 5, output high level;

Step 6, output low level are returned step 2;

Step 7, timer subtract 1;

Step 8, judge whether counter is 0, if be 0 then execution in step 9, otherwise execution in step 10;

Step 9, output low level;

Step 10, output high level return step 2.

Technique effect: this high-frequency impulse modulation (PWM) commutation inversion integrated apparatus can be reverse into the direct current in the energy storage original paper alternating current and provide starting current for the high-speed permanent magnetic synchronous generator in the start-up course of the magneto alternator that is driven by micro fuel engine, and the high-frequency alternating current conversion direct current that when the high-speed permanent magnetic synchronous generator enters steady operation, the high-speed permanent magnetic synchronous generator can be produced, really make energy realize two-way flow, promptly realize the electric energy green conversion.Adopt sliding moding structure direct Power Control method, can respond the conversion of input fast, and insensitive to parameter transformation and disturbance, have good robustness, and physics is made simpler.

Description of drawings

Fig. 1, for high-frequency impulse modulation (PWM) commutation inversion integrated apparatus system architecture diagram of the present invention.

Fig. 2, be the alternating current sample circuit schematic diagram of the embodiment of the invention.

Fig. 3, be the alternating voltage sample circuit schematic diagram of the embodiment of the invention.

Fig. 4, be the direct voltage and the current sampling circuit schematic diagram of the embodiment of the invention.

Fig. 5, be DSP, the CPLD of the embodiment of the invention and the circuit connection diagram of EPM570T100.

Fig. 6, for the control method flow chart of high-frequency PWM commutation inversion integrated apparatus of the present invention.

Fig. 7, the control method flow chart that carries out work for inverter mode of operation of the present invention.

Fig. 8, for SVPWM algorithm flow chart of the present invention.

Fig. 9, the control method flow chart that carries out work for rectifier mode of operation of the present invention.

Figure 10, transfer sliding moding structure direct Power Control method flow diagram for the present invention.

Figure 11, for the acquisition method flow chart of analogue data amount of the present invention.

Figure 12, for the filtering method flow chart of analogue data amount of the present invention.

Figure 13, be that pwm pulse of the present invention generates method flow diagram.

Figure 14, for DSP initialization control method flow chart of the present invention.

Figure 15, for system initialization of the present invention and self check control method flow chart.

Figure 16, be burst pulse restriction flow chart of the present invention.

Figure 17, for software system structure of the present invention.

Figure 18, sliding moding structure direct Power Control block diagram during for main circuit unit rectification of the present invention.

Embodiment

Apparatus of the present invention system architecture diagram as shown in Figure 1.Form by main circuit unit, driving and power amplification unit, pulse modulation unit (PWM), central control unit, human and machine interface unit (HMI), voltage current transformer, electric parameter measurement unit, power subsystem.Wherein, main circuit unit comprises device for power switching IGBT, inverse parallel diode, absorber and DC filter capacitors; The PWM control unit comprises digital signal processor (DSP), programmable logic device (CPLD), RS485 interface circuit; Central control unit comprises programmable logic controller (PLC) (PLC).

Main circuit unit links to each other with voltage current transformer, driving and power amplification unit; Drive and link to each other with the PWM control unit with power subsystem respectively with power amplification unit; The PWM control unit links to each other with central control unit with power subsystem respectively; Central control unit links to each other with human and machine interface unit HMI with the electric parameter measurement unit respectively.

One embodiment of the present of invention adopt the LT308-S7 current sensor of Beijing lime (LEM) company, (LEM) AV100-2000 voltage sensor of selecting for use Beijing lime company to produce; The multi-functional 53U-1211-AD4 voltameter of disk-installed type that the electric parameter measurement unit adopts Japanese Ai Mo company (MSYSTEM) to produce; Central control unit adopts the FX of the production of MIT _2N-32MR chip; Digital signal processor in the PWM control unit (DSP) is selected the TMS320F2812A chip of Texas Instrument, and programmable logic device (CPLD) is selected the EPM570T100 chip of ALTERA company; Driving and power amplification unit adopt the standard drive plate SKHI23/17 that selects the IGBT that is produced by German Xi Menkang (SEMIKRON) company for use; Power subsystem adopts Switching Power Supply; Man-machine interface (HMI) unit adopts the touch-screen PWS6AOOT-P of Hai Tai company.

Wherein, accompanying drawing 2 is an alternating current sample circuit schematic diagram.Ac-side current links to each other with the output of LT308-S7 current sensor, the AC side phase current test side of the alternating current sampling circuit in the output of LT308-S7 current sensor and the PWM control unit links to each other, and alternating current is exported from the Ia-ADC pin through alternating current sampling circuit.

Accompanying drawing 3 is an alternating voltage sample circuit schematic diagram.AC side voltage links to each other with the output of AV100-2000 voltage sensor, the AC side phase voltage test side of the alternating current sampling circuit in the output of AV100-2000 voltage sensor and the PWM control unit links to each other, and alternating voltage is exported from the Ua-ADC pin through the alternating voltage sample circuit.

Accompanying drawing 4 is a voltage and current sample circuit schematic diagram.Wherein, the direct voltage test side of the circuit of direct voltage sampling links to each other with main circuit unit.

Accompanying drawing 5 is the circuit connection diagram of DSP, CPLD and EPM570T100.The AC side that present embodiment adopts adopts three-phase alternating current, comprises 3 road alternating current sample circuits, 3 road alternating voltage sample circuits and 1 road direct voltage sample circuit in the pulse modulation control unit (PWM).Being respectively of three road alternating current sample circuits wherein: alternating current sampling circuit 1, alternating current sampling circuit 2 and alternating current sampling circuit 3; 3 road alternating voltage sample circuits are respectively: alternating voltage sample circuit 1, alternating voltage sample circuit 2 and alternating voltage sample circuit 3.

The Ia-ADC pin of the Ia-ADC pin alternating current sampling circuit 1 of TMS320F2812A chip links to each other; The Ia-ADC pin of the Ib-ADC pin alternating current sampling circuit 2 of TMS320F2812A chip links to each other; The Ia-ADC pin of the Ic-ADC pin alternating current sampling circuit 3 of TMS320F2812A chip links to each other; The Idc-ADC pin of the Idc-ADC pin direct current sample circuit of TMS320F2812A chip links to each other; The Udc-ADC pin of the Udc-ADC pin direct current sample circuit of TMS320F2812A chip links to each other;

The output of LT308-S7 current sensor links to each other with the input of 53U-1211-AD4 voltameter; The output of AV100-2000 voltage sensor links to each other with the input of 53U-1211-AD4 voltameter.

The RS485 interface and the FX of 53U-1211-AD4 voltameter _2NThe RS485 interface of-32MRPLC chip links to each other.

RS485 interface circuit and FX in the pulse modulation control unit (PWM) _2NThe RS485 interface of-32MRPLC chip links to each other.

The TMS320F2812A chip links to each other by DCB with the EPM570T100 chip.The data pin D0 of TMS320F2812A chip～D15 links to each other with the data pin D0～D15 of EPM570T100 chip respectively; The FULT1 of TMS320F2812A chip～FULT8 pin links to each other with the FULT1～FULT8 pin of EPM570T100 chip respectively; The PWM-A1 pin of TMS320F2812A chip links to each other with the PWM-A2 pin with the PWM-A1 pin of EPM570T100 chip respectively with the PWM-A2 pin; The PWM-B1 pin of TMS320F2812A chip links to each other with the PWM-B2 pin with the PWM-B1 pin of EPM570T100 chip respectively with the PWM-B2 pin; The PWM-C1 pin of TMS320F2812A chip links to each other with the PWM-C2 pin with the PWM-C1 pin of EPM570T100 chip respectively with the PWM-C2 pin.

The PWM-A1-Q port of EPM570T100 chip links to each other with the input port A of standard drive plate SKHI23/17; The PWM-A1-Q port of EPM570T100 chip links to each other with the input port B of standard drive plate SKHI23/17; IGBT-fault 00 port of EPM570T100 chip links to each other with the output of standard drive plate SKHI23/17.

Sliding moding structure direct Power Control block diagram when Figure 18 is the rectification of invention main circuit unit.The concrete formula of sliding moding structure Direct Power is as follows:

Symmetrical and when being sinusoidal wave, pref, qref are active power reference value and reactive power reference qref when main circuit unit AC side three, E is the AC side voltage effective value, then

$\left[\begin{array}{c}{i}_a\\ i_b\\ i_c\end{array}\right]=\frac{\sqrt{2}{p}_{\mathrm{ref}}}{3E}\left[\begin{array}{c}\sin \mathrm{\θ}\\ \sin (\mathrm{\θ}-2\mathrm{\π}/3)\\ \sin (\mathrm{\θ}+2\mathrm{\π}/3)\end{array}\right]+\frac{\sqrt{2}{q}_{\mathrm{ref}}}{3E}\left[\begin{array}{c}\cos \mathrm{\θ}\\ \cos (\mathrm{\θ}-2\mathrm{\π}/3)\\ \cos (\mathrm{\θ}+2\mathrm{\π}/3)\end{array}\right]---\left(1\right)$

By following formula as can be known, under certain alternating voltage,, definite three-phase current state is just arranged by setting active power and reactive power.Promptly control the rectification control that active power and reactive power can realize main circuit unit.

With instantaneous meritorious p, instantaneous reactive q is the power controlling models of variable.

$\left\{\begin{array}{c}L\frac{\mathrm{dp}}{\mathrm{dt}}=\frac{3}{2}{U}_m^2-\mathrm{Rp}-\mathrm{\ωLq}-\sqrt{\frac{3}{2}}{U}_m{u}_{\mathrm{rd}}\\ L\frac{\mathrm{dq}}{\mathrm{dt}}=-\mathrm{Rq}+\mathrm{\ωLp}+\sqrt{\frac{3}{2}}{U}_m{u}_{\mathrm{rq}}\end{array}\right.---\left(2\right)$

From formula (2), power controlling models in the time of can getting the main circuit unit rectification:

$\left\{\begin{array}{c}\frac{\mathrm{dp}}{\mathrm{dt}}=\frac{3}{2L}{E}^2-\frac{R}{L}p-\mathrm{\&omega;q}-\frac{1}{L}\sqrt{\frac{3}{2}}E{S}_p{v}_{\mathrm{dc}}\\ \frac{\mathrm{dq}}{\mathrm{dt}}=-\frac{R}{L}q+\mathrm{\&omega;p}+\frac{1}{L}\sqrt{\frac{3}{2}}{\mathrm{ES}}_q{v}_{\mathrm{dc}}\\ \frac{d}{\mathrm{dt}}\left(v_{\mathrm{dc}}^2\right)=-\frac{2}{R_L\mathrm{<figure-callout\; id="2"\; label="C\; v\; dc"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">C</figure-callout>}}{v}_{\mathrm{dc}}^{}{}_2^{}+\frac{2}{C}p\end{array}\right.---\left(3\right)$

Wherein, variable S _p, S _qWith state variable v _DcCoupling, u _Rd=S _pv _Dc, u _Rq=S _qv _DcWherein, E is the effective value of AC side voltage, and ω is the angular frequency of permanent magnet synchronous machine output voltage, S _XFor the two-valued function switch function under the d-q coordinate (X=d, q), v _DcBe dc voltage, L is an inductor rectifier, and C is the DC side filter capacitor, U _RdBe d axle DC voltage component, U _RqBe q axle DC voltage component.

Choosing of sliding-mode surface

Because two external control amount: v are arranged during the main circuit unit rectification _DcAnd q.According to the selection principle and the formula (3) of sliding-mode surface, we can choose sliding-mode surface and are

$\left\{\begin{array}{c}{S}_1=k_1(q^{*}-q)=0\\ {\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">S</figure-callout>}}_2=k_2(v_{\mathrm{dc}}^{*2}-v_{\mathrm{dc}}^2)+\frac{d}{\mathrm{dt}}(v_{\mathrm{dc}}^{*2}-v_{\mathrm{dc}}^2)=0\end{array}\right.---\left(4\right)$

Q wherein ^*Be the set-point of reactive power, v _Dc ^*Be rectification dc voltage set-point, v _DcBe the dc voltage actual value.Formula (3) is reached $\frac{d}{\mathrm{dt}}{v}_{\mathrm{dc}}^{*2}=0$ Be updated in the formula (4), can get

$\left\{\begin{array}{c}{S}_1=k_1(q^{*}-q)=0\\ {\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">S</figure-callout>}}_2=\frac{{\mathrm{<figure-callout\; id="2"\; label="Ck"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">Ck</figure-callout>}}_2}{2}(v_{\mathrm{dc}}^{*2}-v_{\mathrm{dc}}^2)+\frac{v_{\mathrm{dc}}^2}{R_L}-p=0\end{array}\right.---\left(5\right)$

Formula (5) is carried out further abbreviation, can get

$\left\{\begin{array}{c}{S}_1=q^{*}-q=0\\ {\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">S</figure-callout>}}_2={\mathrm{Ck}}_2(v_{\mathrm{dc}}^{*2}-v_{\mathrm{dc}}^2)+v_{\mathrm{dc}}{i}_0-P=0\end{array}\right.---\left(6\right)$

If order

$\left\{\begin{array}{c}{q}^{*}=0\\ P^{*}={\mathrm{Ck}}_2(v_{\mathrm{dc}}^{*2}-v_{\mathrm{dc}}^2)+v_{\mathrm{dc}}{i}_0\end{array}\right.---\left(7\right)$

Can get $\left\{\begin{array}{c}{S}_1=q^{*}-q=0\\ {\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">S</figure-callout>}}_2=p^{*}-p=0\end{array}\right.---\left(8\right)$

Determining of control rate

In native system, switching line has two, according to satisfying condition of broad sense sliding formwork condition, discusses respectively and how to select control rate to make two switching lines satisfy broad sense sliding formwork condition.

At first analyze the control rate of S1:

When S1＞0, promptly during q*＞q, satisfy broad sense sliding formwork condition and then need make $\frac{{\mathrm{dS}}_1}{\stackrel{\&CenterDot;}{d}t}=-\frac{\mathrm{dq}}{\mathrm{dt}}<0,$ Need to increase q so that ${\stackrel{\&CenterDot;}{S}}_1<0,$ The controlled quentity controlled variable that promptly requires to select can increase q.

When S1＜0, promptly during q*＜q, satisfy broad sense sliding formwork condition and then need make $\frac{{\mathrm{dS}}_1}{\mathrm{dt}}=-\frac{\mathrm{dq}}{\mathrm{dt}}>0,$ Need reduce q so that ${\stackrel{\&CenterDot;}{S}}_1>0.$ The controlled quentity controlled variable that promptly requires to select can reduce q.

In like manner can obtain the control rate of S2.

As seen from the above analysis, the fundamental nature of its control rate is that actual value will be followed set-point constantly, so that deviation reduces, this attribute can be realized with the ring control strategy that stagnates.

It is as follows that the SVPWM algorithm is derived, principle as shown in figure 19:

$\left\{\begin{array}{c}L\frac{{\mathrm{di}}_a}{\mathrm{dt}}+{\mathrm{Ri}}_a=e_a-(s_a{v}_{\mathrm{dc}}+u_{\mathrm{NO}})\\ L\frac{{\mathrm{di}}_b}{\mathrm{dt}}+{\mathrm{Ri}}_b=e_b-(s_b{v}_{\mathrm{dc}}+u_{\mathrm{NO}})\\ L\frac{{\mathrm{di}}_c}{\mathrm{dt}}+{\mathrm{Ri}}_c=e_c-(s_c{v}_{\mathrm{dc}}+u_{\mathrm{NO}})\\ C\frac{{\mathrm{dv}}_{\mathrm{dc}}}{\mathrm{dt}}=i_{\mathrm{dc}}-i_L\end{array}\right.---\left(9\right)$

Wherein: $E_{\mathrm{abc}}^s={\left[\begin{array}{ccc}{e}_a& e_b& e_c\end{array}\right]}^T$ Be permanent magnet synchronous machine back-emf space vector; $U_{\mathrm{abc}}^s={\left[\begin{array}{ccc}{u}_a& u_b& u_c\end{array}\right]}^T$ Be permanent magnet synchronous machine output end voltage space vector; $I_{\mathrm{abc}}^s={\left[\begin{array}{ccc}{i}_a& i_b& i_c\end{array}\right]}^T$ Be permanent magnet synchronous machine output current space vector; i _DcBe the DC side electric current; i _LBe load current; v _DcBe dc voltage; C is the DC side filter capacitor; L is an inductor rectifier, u _NOBe the voltage between DC side N point and the motor O point; R is a stator resistance.

Space voltage vector has been described three-phase main circuit unit AC side phase voltage (u _a, u _b, u _c) spatial distribution on complex plane, S in the formula (9) _a, S _b, S _cBe three-phase unipolarity two-valued function switch function.With 2 ³=8 kinds of switch function combinations promptly obtain corresponding three-phase main circuit unit AC side magnitude of voltage for people's formula (1).Three-phase main circuit unit space voltage vector

Distribution may be defined as

$\left\{\begin{array}{c}{\stackrel{\&CenterDot;}{V}}_k=\frac{2}{3}{v}_{\mathrm{dc}}{e}^{j(k-1)\frac{1}{3}\mathrm{\&pi;}}\\ V_{\mathrm{0,7}}=0\end{array}\right.(k=1,...,6)---\left(10\right)$

Formula (10) can be expressed as the switch function form, promptly

${\stackrel{\&CenterDot;}{V}}_k=\frac{2}{3}{v}_{\mathrm{dc}}(S_a+S_b{e}^{j\frac{1}{3}\mathrm{\&pi;}}+S_c{e}^{-\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">j</figure-callout>}\frac{2}{3}\mathrm{\&pi;}})(k=0,...,7)---\left(11\right)$

For any given three-phase fundamental voltage u _a, u _b, u _c, if consider three-phase balanced system, i.e. u _a+ u _b+ u _c=0, then can in complex plane, define space vector of voltage

$\stackrel{\&CenterDot;}{V}=\frac{2}{3}(v_{a0}+v_{b0}{e}^{j\frac{1}{3}\mathrm{\&pi;}}+v_{c0}{e}^{-\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">j</figure-callout>}\frac{2}{3}\mathrm{\&pi;}})---\left(12\right)$

6 moulds are 2v _Dc/ 3 space voltage vector is divided into six sector region I～VI with complex plane, for the voltage vector in arbitrary sector region

All can synthesize by the space voltage vector on these both sides, fan section.If

On complex plane at the uniform velocity rotation, the just corresponding three symmetrical sinusoidal quantities that obtained.In fact, because the restriction of switching frequency and vectorial combination,

Resultant vector can only be a polygon director circle track thereby make vector end points movement locus with the rotation of a certain stepping rate.Obviously, the PWM switching frequency is high more, and polygon director circle track is approaching more circle just.If

When the I district, then

Can by

With

Synthetic, according to the weber balance rule, have

${\stackrel{\&CenterDot;}{V}}^{*}{T}_s={\stackrel{\&CenterDot;}{V}}_1{T}_1+{\stackrel{\&CenterDot;}{V}}_2{\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">T</figure-callout>}}_2+{\stackrel{\&CenterDot;}{V}}_{\mathrm{0,7}}{T}_{\mathrm{0,7}}---\left(13\right)$

Wherein, T ₁And T ₂Be respectively space vector

With

Action time, T _sIt is a sampling period.

Make zero vector

Duration be T _0,7, then

T ₁+T ₂+T _0，7＝T _s (14)

Make V ^*With V ₁Between angle be θ, calculate by the law of sines

$\frac{\left|{\stackrel{\&CenterDot;}{V}}^{*}\right|}{\mathrm{<figure-callout\; id="2"\; label="sin"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">sin</figure-callout>}\frac{2\mathrm{\&pi;}}{3}}=\frac{\left|\frac{T_2}{T_1}{\stackrel{\&CenterDot;}{V}}_2\right|}{\sin \mathrm{\&theta;}}=\frac{\left|\frac{T_2}{T_1}{\stackrel{\&CenterDot;}{V}}_1\right|}{\sin (\frac{\mathrm{\&pi;}}{3}-\mathrm{\&theta;})}---\left(15\right)$

Again because | V ₁|=| V ₂|=2v _Dc/ 3, then simultaneous formula (13), (14), formula (15) are easy to get

$\left\{\begin{array}{c}{T}_1=m{T}_s\sin (\frac{\mathrm{\&pi;}}{3}-\mathrm{\&theta;})\\ {\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="A2009100128460020H1.png"\; state="\{\{state\}\}">T</figure-callout>}}_2=m{T}_s\sin \mathrm{\&theta;}\\ T_{\mathrm{0,7}}=T_s-T_1-T_2\end{array}\right.---\left(16\right)$

In the formula, the m-SVPWM index of modulation, and

$m=\frac{\sqrt{3}}{v_{\mathrm{dc}}}\left|{\stackrel{\&CenterDot;}{V}}^{*}\right|---\left(17\right)$

For the selection of zero vector, main consideration is selected Or It is the least possible that on off state is changed, to reduce switching loss.In a switch periods, make that the zero vector insertion time is T _0,7, if wherein insert

Time be T ₀=kT _0,7, then insert

Time then be T ₇=(1-k) T _0,7, 0≤k≤1 wherein.

Claims (7)

1. A high-frequency PWM rectification and inverter integrated device, characterized in that it includes a main circuit unit, a drive and power amplification unit, a PWM control unit, a central control unit, a human-machine interface unit HMI, a voltage and current transformer, and power parameters Measuring unit and power supply unit, wherein the main circuit unit includes power switching device IGBT, anti-parallel diode, absorber and DC filter capacitor; PWM control unit includes digital signal processor DSP, programmable logic device CPLD and RS485 interface circuit, where CPLD Contains a narrow pulse limiter module, which is used to deal with faults; the central control unit includes a programmable logic controller PLC unit, The main circuit unit is connected with the voltage and current transformer, drive and power amplification unit; the drive and power amplification unit are respectively connected with the power supply unit and the PWM control unit; the PWM control unit is connected with the power supply unit and the central control unit respectively; the central control unit is respectively connected with the power The parameter measurement unit is connected with the human-machine interface unit HMI. 2. The high-frequency PWM rectification and inverter integrated device according to claim 1, characterized in that: the narrow pulse limiting module included in the CPLD handles faults as follows: Step 1. Clear the timer; Step 2, judge whether the input is high, if it is high, execute step 3, otherwise execute step 7; Step 3, add 1 to the timer; Step 4. Judging whether the time limit is reached, if it is reached, execute step 5, if not, execute step 6; Step 5, output high level; Step 6, output low level, return to step 2; Step 7, the timer minus 1; Step 8, judge whether the counter is 0, if it is 0, execute step 9, otherwise execute step 10; Step 9, output low level; Step 10, output high level, return to step 2. 3. The control method of the high-frequency PWM rectification and inverter integrated device according to claim 1, characterized in that it comprises the following steps: Step 1. Start the motor and accelerate; Step 2. Send the monitoring speed signal to the central control unit at 10Hz; Step 3. Detect whether the motor is started, if it is started, go to step 4, otherwise go to step 1; Step 4, work in inverter mode; Step 5. Determine whether the rotational speed is greater than 36000RPM, if it is greater, perform step 4, otherwise perform step 6; Step 6, work in rectifier mode; Step 7, end. 4. The control method of the high-frequency PWM rectification and inverter integrated device according to claim 3, characterized in that: the steps of working in the inverter mode described in step 4 are as follows: Step 1. Start; Step 2, judging whether the start command arrives, if it arrives, execute step 3, otherwise execute step 2; Step 3. Inverter working mode: call the SVPWM algorithm; Step 4, motor speed up; Step 5, judge whether it reaches 600Hz, if it reaches, execute step 6, otherwise execute step 4; Step 6, inverter constant speed control; Step 7, judging whether the stop command arrives, if it arrives, then execute step 8, otherwise execute step 6; Step 8, motor deceleration; Step 9, judging whether to reach the shutdown turn, if so, execute step 10, otherwise execute step 8; Step 10, stop the pulse sealing subroutine, and execute step 2. 5. The control method of the high-frequency PWM rectification and inverter integrated device according to claim 3, characterized in that: the steps of working in the rectifier mode described in step 6 are as follows: Step 1, PWM interrupt processing; Step 2, interrupt protection; Step 3, clear the interrupt flag; Step 4, DC voltage detection program; Step 5, AC current side sampling, synchronous signal detection sampling, calling the method of collecting analog data volume; Step 6, filtering and data processing, calling the filtering method of the analog data volume; Step 7. Determine whether the motor speed is greater than 36000RPM, if it is greater, perform step 8, otherwise perform step 11; Step 8, rectifier working mode: adjusting sliding mode variable structure direct power control method; Step 9, calling the PWM pulse control method and the narrow pulse limiting method; Step 10, add 1 to the program pointer; Step 11, return. 6. The control method of the high-frequency PWM rectification and inverter integrated device according to claim 4, characterized in that: the step of calling the SVPWM control method in step 3 is as follows: Step 1. Start; Step 2, processing the AD sampling; Step 3, error judgment and processing; Step 4, receiving the setting value of the central control unit through the SCI; Step 5, regularly upload the sampling value; Step 6. Determine whether there is an interruption, if so, execute step 7, otherwise execute step 2; Step 7, clear the interrupt occurrence flag; Step 8, setting the parameters of SVPWM according to the current frequency; Step 9, calculating S _a , S _b , S _c ; Step 10. call the PWM pulse control method and the narrow pulse limit method; Step 11, end. 7. The control method of the high-frequency PWM rectification and inverter integrated device according to claim 5, characterized in that: the method of direct power control of the sliding mode variable structure described in step 8 is as follows: Step 1. Start; Step 2, call the filtering method of the analog data volume; Step 3, carry out abc to aβ transform and calculate θ angle; Step 4, calculate instantaneous power P and Q, calculate power factor; Step 5. Receive the setting value of the central control unit through the SCI. The given value of reactive power Q ^* = 0, and calculate the given value of active power P ^* : P ^* = Ck ² (V _dc ^*2 -V _dc ² )+V _dc ×I0, where V _dc ^* is the given value of the DC side voltage at the rectification side, and V _dc is the actual value of the DC side voltage at the rectification side; Step 6. Regularly upload voltage, current, power, and power factor to the central control unit; Step 7. Determine whether there is an interruption, and if so, execute step 7; otherwise, execute step 2; Step 8, clear the interrupt occurrence flag; Step 9, according to P, Q, P ^* , Q ^* , θ angle, calculate the value of switch function S _p and switch function S _q ; Step 10, calculating switch functions S _a , S _b , S _c ; Step 11, calling the PWM generation control method; Step 12, end.

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