CN101136607A - Motor and drive control device therefor - Google Patents
Motor and drive control device therefor Download PDFInfo
- Publication number
- CN101136607A CN101136607A CNA200710146567XA CN200710146567A CN101136607A CN 101136607 A CN101136607 A CN 101136607A CN A200710146567X A CNA200710146567X A CN A200710146567XA CN 200710146567 A CN200710146567 A CN 200710146567A CN 101136607 A CN101136607 A CN 101136607A
- Authority
- CN
- China
- Prior art keywords
- current
- motor
- instruction value
- voltage
- control
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Pending
Links
Images
Abstract
The present invention provides motor drive control device for controlling motor with three phases, wherein the current or induced voltage waveform of the motor is square wave or quasi square wave with n order harmonic, n is equal to 2, 3, 4 etc., the motor drive control device is characterized in that including d-q voltage calculation part for calculating voltage ed and eq as d axis and q axis of back electromotive force separately; d axis instruction calculation part for figuring out current instruction value Idref as current instruction value d axis component according to torque instruction value Tref, reference angular speed omega b and machinery angular speed omega m, under the state of the machinery angular speed omega m is above to the reference angular speed omega b of the motor; q axis instruction current calculation part, for figuring out current instruction value Iqref as current instruction value q axis component according to voltage ed, voltage eq and current instruction value Idref; current control part, for controlling the current of the motor according to the current instruction value Idref and current instruction value Iqref.
Description
The application is that international filing date is on November 27th, 2003, and international application no is PCT/JP2003/015166, and national applications number is 200380104519.4, and denomination of invention is divided an application for the application for a patent for invention of " motor and driving control device thereof ".
Technical field
The present invention relates to the improvement of a kind of motor that is adapted at using in the driven steering device most and control device thereof and the driven steering device that uses them.
Background technology
The motor that uses in the driven steering device was generally permanent magnet synchronous motor (PMSM) in the past, and permanent magnet synchronous motor is driven by the sine-wave current of three-phase.In addition, as the control mode of drive motors, be extensive use of the control mode that is called as vector control.But the miniaturization of driven steering device requires strong, has the tendency of using brushless DC motor as being suitable for the motor of miniaturization.
Under this situation, with reference to Fig. 1 in the past use the motor drive control device of vector control mode of electric machine for electrodynamic steering gear describe.
Its structure is connected with by detecting subtracter 20-1,20-2, the 20-3 of the error between current instruction value Iavref, Ibvref, Icvref and current of electric Ia, Ib, the Ic for the back of calculating portion 100 at the current instruction value of the electric current of controlling motor 1, input is from the PI control part 21 of each error signal of subtracter 20-1,20-2,20-3, input arrives the main path of motor 1 from voltage va, the vb of PI control part 21, the PWM control part 30 of vc with direct current being converted to the inverter 31 that exchanges.Between inverter 31 and motor 1, be equipped with and detect current of electric Ia, current detection circuit 32-1, the 32-2 of Ib, Ic, 32-3, and constitute the feedback control system B that detected current of electric Ia, Ib, Ic is fed back to subtracter 20-1,20-2,20-3 respectively.
Next, current instruction value being calculated portion 100 describes.At first, import: according to the torque instruction value Tref that calculates by the detected torque of not shown torque sensor about its input; Rotational angle theta e by the rotor in the detected motor 1 of the position-detection sensor that is connected with motor 1 11; The electric angle speed omega e that calculates by differential circuit 24.As input, utilize conversion portion 101 to calculate back electromotive force ea, eb, ec the rotational angle theta e of electric angle speed omega e and rotor.Then utilize 3 phases/2 phase transformation portions 102 to convert the d axle to and become component voltage ed, q axle to become component voltage eq, become component voltage ed, q axle to become component voltage eq on this d axle, utilize q axle instruction current to calculate the current instruction value Iqref that the q axle is calculated by portion 108 as input.Wherein, in this case, the current instruction value Idref=0 that establishes the d axle carries out computing.That is, in the output equation formula of motor
Tref×ωn＝3/2(ed×Id+eq×Iq)...(1)
In, when input Id=Idref=0, with
Iq＝Iqref＝2/3(Tref×ωn/eq) ...(2)
Calculate.Advance angle Φ according to current instruction value Iqref that calculates portion 108 from q axle instruction current and advance angle described later control calculates current instruction value Iavref, Ibvref, Icvref.That is, q axle instruction current is calculated portion's 108 inputs and is calculated advance angle Φ and the current instruction value Iqref that portion 107 calculates by advance angle, utilizes 2 phases/3 phase transformation portions 109 to calculate current instruction value Iavref, Ibvref, Icvref.
In addition, (" acos " represents cos to use Φ=acos (ω b/ ω m) or Φ=K functions such as (1-(ω b/ ω m)) in experiment
-1).And the reference angle speed omega b of motor is the motor limiting angle speed when not using weakened field to control drive motors 1.Fig. 2 represents the relation of torque T and motor speed n (angular velocity omega e), and expression does not have the example of the limiting angle speed omega b under the weakened field control situation.
Next, control describes to advance angle.
At motor 1 not during the high speed rotating, promptly the mechanical angle speed omega m of motor 1 be lower than motor reference angle speed omega b during, current instruction value Iavref, Ibvref, Icvref, irrelevant with advance angle Φ, if, then can export the torque that obtains according to torque instruction value Tref according to controlling by the value that 2 phases/3 phase transformation portions 109 calculate according to current instruction value Iqref.That is,, mean that the driver can successfully carry out steering operation as driven steering device.
In addition, at the motor high speed rotating, when promptly the mechanical angle speed omega m of motor is higher than the reference angle speed omega b of motor,, then can not realize higher angular speed than reference angle speed omega b if do not carry out the control that has added advance angle Φ.When the high speed rotating of this motor is replaced driven steering device, under the situation of changing direction for parking or swerving,, driver's the sense that turns to is worsened because motor 1 can not be followed steering operation because of the act of rescue.
Torque control during as the motor high speed rotating has a kind of control mode that is called as weakened field control, as its concrete method advance angle control is arranged.The detailed content of this advance angle control mode is at United States Patent (USP) the 5th, 667, No. 605 specifications (C1), (Vol 43 for IEEE Transaction onIndustrial Electronics for C.C.Chanetal " NovelPermanent Magnet Motor Drivers for Electric Vehicles ", No.2 April 1996 page335, Fig.5) middle record.The advance angle control mode be characterised in that make current instruction value Iqref phase place only leading angle Φ generate the field weakening composition.In Figure 10 (B), when making current instruction value Iqref only during leading angle Φ, produce Iqref * sin Φ as d axle composition, produce Iqref * cos Φ as q axle composition.Here, Iqref * sin Φ plays a role as the field weakening composition, and Iqref * cos Φ plays a role as the torque composition.
In addition, the motor-driven control mode as using in driven steering device, adopt following vector control: the position of rotation according to rotor, produce rotating magnetic field by controller by inverter, come the rotation of drive controlling rotor.That is, vector control by with the predetermined angular arranged spaced on a plurality of magnet exciting coils 1 on the outer peripheral face of rotor, utilize control circuit to switch the excitation of each magnet exciting coil successively according to rotor-position, the rotation of controlling rotor thus drives.
This vector control is opened among the 2001-18822 (A) etc. open for example spy.Fig. 3 is the functional module structure of an example of the drive controlling of the motor 56 of expression under the vector control.
In Fig. 3, be formed with from the instruction current determination section 51 of control command value of decision motor 56, via PI control part 52,2 phases/3 phase coordinates converter sections 53, PWM voltage generating unit 54, inverter 55 command signal main path until motor 56.In addition, configuration current sensor 571,572 between inverter 55 and motor 56, and formation feedback path, in this feedback path, in 3 phases/2 phase coordinates converter sections 59, will convert 2 phases to, and 2 phase current composition Iq, Id will be fed back in the subtraction circuit 581,582 that is provided between instruction current determination section 51 and the PI control part 52 by these current sensor 571,572 detected current of electric.
Utilize this control system, in instruction current determination section 51, the rotational angle theta and the electric angle ω of the rotor that torque instruction value Tref, the position-detection sensor that reception is calculated according to the torque that torque sensor detected detected, decision current instruction value Idref, Iqref.In subtraction circuit 581,582, utilize the 2 phase current composition Iq, the Id that convert 2 phases by the 3 phases/2 phase coordinates converter sections 59 of feedback path to respectively, these current instruction values Idref, Iqref are carried out feedback compensation.That is the error between computing 2 phase current composition Iq, Id and current instruction value Idref, the Iqref in subtraction circuit 581,582.Then, in PI control part 521,522, the signal of duty ratio of calculating expression PWM control with the form of d composition and q composition is as command value Vd and Vq, utilizes 2 phases/3 phase coordinates converter sections 53 to reverse from d composition and q composition and changes 3 phase constituent Va, Vb, Vc into.Then, inverter 55 is carried out PWM control, provide inverter current to motor 56 according to command value Va, the Vb of 3 phases, Vc, thus the rotation of control motor 56.
In addition, the 61st, vehicle speed sensor, the 62nd, induction region decision circuit, the 63rd, coefficient generation circuit, the 64th, basic power-assisted counting circuit, the 65th, restoring force counting circuit, the 66th, electric angle converter section, the 67th, angular speed converter section, the 68th, non-Interference Control correction value portion.
Under the situation of vector control as described above, determine current instruction value Idref, Iqref according to torque instruction value Tref and electric angle ω, rotational angle theta.In addition, after feedback current Iu, Iw with motor 56 convert 3 phase current Iu, Iv, Iw to, convert 2 phase current composition Id, Iq to, then, error in subtraction circuit 582 and 581 between computing 2 phase current composition Id, Iq and current instruction value Idref, the Iqref, this error realizes obtaining command value Vd, Vq for inverter 55 thus based on the Current Control of PI control.And, in 2 phases/3 phase coordinates converter sections 53, command value Vd, Vq are reversed command value Va, Vb, the Vc that changes 3 phases into once more, and control inverter 55, thereby carry out the drive controlling of motor 56.
But the d axle composition and the q axle composition that produce owing to advance angle control make only leading phase Φ of current instruction value Iqref, so the Iqref of the Iqref of d axle * sin Φ and q axle * sin Φ is defined in fixing relation, balance optimization that may not the amount of making.Its result, motor terminal voltage saturation when high speed rotating, current of electric can not instruct by follow current, and it is big that torque ripple becomes, and it is big that motor noise also becomes.Therefore,, when fast steering is operated, feel by steering wheel unusual vibration perhaps to cause motor noise, produce the rough sledding that brings the offending sensation of driver as driven steering device.
In addition, under the situation of vector control as described above, the detection electric current of motor 56 is output as 3 mutually with inverter 55, and feedback control system is 2 phases.Like this, must come drive controlling motor 56 by converting 3 phases to from 2 contraries once more at 2 phases/3 phase coordinates converter sections 53, owing to mixed 2 phases/3 phase transformations and 3 phase transformations mutually/2, so the problem that exists control system integral body to complicate.
In addition, if can keep the linear characteristic of control system, then the control response of the control of motor 56 becomes well, and control becomes easily, also can easily reach controlled target.But, in the drive controlling of motor 56, comprise various nonlinear factors.As making motor-driven produce nonlinear factor, the Dead Time (dead time) of inverter control is for example arranged.Promptly, though use the switch element of FET (Field Effect Transistor field-effect transistor) as inverter, but FET is not desirable switch element, for prevent on the short-and-medium road of last underarm, the FET that makes underarm simultaneously be set be cut-off state during (Dead Time).Utilization has in the current of electric that the FET switch of such Dead Time produces, and comprises the non-linear factor of switch transition state.In addition, the detecting element of detection current of electric and testing circuit etc. also comprise non-linear factor.
In this case, the d-q of the 3 phases/2 phase coordinates converter sections 59 by reponse system conversion, for example the non-linear factor that produces in a phase current Ia is comprised among d shaft current composition Id and the q shaft current composition Iq.Therefore, carry out Current Control according to electric current composition Id, Iq, calculate command value Vd and Vq from PI control part 522,521 to inverter 55, utilize again 2 phases/3 phase coordinates converter sections 53 from d mutually and the q contrary convert to a mutually, b phase and c mutually, calculate command value Va, Vb, the Vc of 3 phases.Thus, the non-linear factor that was included in originally among a phase current Ia is changed by d-q, diffuses among command value Va, Vb, the Vc of inverter 55, is not only a phase, and b also comprises non-linear factor mutually with in the c command value mutually.That is, under the situation of above-mentioned control mode in the past, although with 3 phase drive motors, but with 2 Current Control of coming computing to feed back mutually, will command value Vd, the Vq of decision convert 3 phase command value Va in form to mutually by 2, Vb, Vc control, therefore, non-linear factor diffusion.
So, exist the torque ripple that causes because of above-mentioned Electric Machine Control in the past to become noise big, motor and also become big problem.In addition, when such Electric Machine Control is applied to driven steering device, can not follows steering operation and carry out correct and stable power-assisted, be created in and feel when turning to that vibration or noise become big such problem.
Summary of the invention
The present invention proposes in view of the above problems, the objective of the invention is to, be separated to by the non-linear factor that in Electric Machine Control, is comprised under the state of each phase and control, a kind of torque ripple is little, noise is little motor and driving control device thereof are provided, and, a kind of this motor and driving control device thereof of adopting in driven steering device is provided, makes steering behaviour improve, have the good driven steering device that turns to sense.
The present invention also aims to, a kind of motor drive control device and driven steering device thereof are provided, its motor terminal voltage when the motor high speed rotating is also unsaturated, torque ripple is little, motor noise is little, when swerving operation in driven steering device, noise is little, steering operation can be followed reposefully.
The present invention relates to motor, above-mentioned purpose of the present invention reaches in the following way: at the motor induction voltage waveform is square wave or quasi-retangular wave, and the harmonic components in the time of will carrying out frequency analysis to described square wave or quasi-retangular wave be made as n (=2,3,4 ...) situation under, if P is a number of poles, if ω is an actual speed, then 5% harmonic components n more than or equal to the amplitude composition satisfies
The higher limit of the response frequency of n * P/2 * ω≤Current Control.
In addition, the present invention relates to a kind of motor drive control device that motor with 3 phases is controlled, the electric current of wherein said motor or induction voltage waveform are square wave or the quasi-retangular wave with nth harmonic, n=2,3,4, it is characterized in that having: d-q voltage is calculated portion, and it is calculated respectively as the d axle of back electromotive force and the voltage ed and the eq of q axle composition; D axle instruction current is calculated portion, be higher than at the mechanical angle speed omega m of detected described motor under the situation of reference angle speed omega b of described motor, this d axle instruction current is calculated portion and is calculated current instruction value Idref as the d axle composition of current instruction value according to the mechanical angle speed omega m of the reference angle speed omega b of the torque instruction value Tref of described motor, described motor and detected described motor; Q axle instruction current is calculated portion, and it is according to described voltage ed, and voltage eq and current instruction value Idref calculate the current instruction value Iqref as the q axle composition of current instruction value; Current control division, it controls the electric current of described motor according to described current instruction value Idref and described current instruction value Iqref.
Above-mentioned motor drive control device among the present invention can will be calculated described current instruction value Iqref in described mechanical angle speed omega m, voltage ed, voltage eq, torque instruction value Tref and the current instruction value Idref substitution motor output equation formula.
Can realize above-mentioned purpose of the present invention in the following way more effectively: described current control circuit comprises integral control, and perhaps described motor is a brushless DC motor, perhaps uses the driven steering device of described motor drive control device.
Description of drawings
Fig. 1 is based on the control block diagram of advance angle control in the past.
Fig. 2 is expression as the figure of the reference angle speed of the limiting angle speed of not using weakened field when control.
Fig. 3 is a control block diagram of representing the control mode of vector control in the past.
Fig. 4 is the cross-section structure of expression as an example of the brushless DC motor of controlling object of the present invention.
Fig. 5 is the figure of the principle of expression rotor position detection.
Fig. 6 is the relevant figure of defined declaration with trapezoidal wave electric current (voltage).
Fig. 7 is the figure of an example of expression induction voltage waveform (square wave).
Fig. 8 is the block diagram of an example of the control system of expression brushless DC motor of the present invention.
To be expression control the block diagram of the structure example that relevant current instruction value Idref calculates with weakened field of the present invention to Fig. 9.
Figure 10 is an expression control mode of the present invention and the figure of the vector correlation of the current instruction value Idref of in the past advance angle control mode and Iqref.
Embodiment
Below, with reference to accompanying drawing embodiments of the present invention are described.
In this example, though 3 phase brushless DC motors are described, the present invention is not limited to this, can use the present invention too for other motor.
As shown in Figure 4,3 phase brushless DC motors 1 of the present invention have: columnar shell 2; Rotating shaft 4, its axle center along this shell 2 sets, and is supported by bearing 3a, the 3b of upper and lower end parts with rotating freely; Motor-driven permanent magnet 5, it is fixed on this rotating shaft 4; Stator 6, its inner peripheral surface that is fixed on shell 2 to be surrounding this permanent magnet 5, and be wound with magnet exciting coil 6a, 6b, the 6c of 3 phases.Constituted rotor 7 by rotating shaft 4 and permanent magnet 5.Near an end of the rotating shaft 4 of this rotor 7, be fixed with the annular permanent magnnet 8 that phase-detection is used, this permanent magnet 8 upwards is polarized to the S utmost point and the N utmost point with equal intervals alternately in week.
On the side end face that is equipped with bearing 3b in the shell 2, be equipped with the supporting substrate 10 that constitutes by annular thin sheets by strut 9.On this supporting substrate 10, be fixed with the rotor position detector 11 of decomposer and encoder etc. mutually opposed to each other with permanent magnet 8.Exception as shown in Figure 5, in fact corresponding to the driving timing of magnet exciting coil 6a～6c, upwards suitably is provided with to distance a plurality of rotor position detectors 11 in week at interval.Here, magnet exciting coil 6a～6c sets the outer peripheral face that surrounds rotor 7 for electric angle 120 degree that separate each other, and the coil resistance of each magnet exciting coil 6a～6c all equates.
In addition, rotor position detector is according to the magnetic pole outgoing position detection signal of relative permanent magnet 8.Rotor position detector 11 utilizes its magnetic pole along with permanent magnet 8 this point that changes, the position of rotation of detection rotor 7.According to this position of rotation, vector control phase current instruction efferent 20 described later carries out 2 energisings mutually to 3 phase excitation coil 6a～6c simultaneously, and utilizes and pursue 2 phase excitation modes that switch magnet exciting coil 6a～6c mutually successively, and rotation drives rotor 7.
Then, the drive controlling of motor 1 uses square wave electric current or quasi-retangular wave electric current as current of electric, perhaps uses square-wave voltage or quasi-retangular wave voltage to control as the motor induced voltage.
Here, the control of carrying out when the square-wave voltage that will utilize square wave electric current or quasi-retangular wave electric current or induced voltage or quasi-retangular wave voltage, when comparing with sine-wave current or sine voltage, if current peak or voltage peak are identical, then, therefore can obtain bigger output valve (power) because the effective value of square wave electric current or square-wave voltage becomes big.Its result under the situation of the motor of making identical performance, uses square wave electric current or quasi-retangular wave electric current as current of electric, perhaps uses square-wave voltage or quasi-retangular wave voltage to be pressed with the advantage of the miniaturization that can realize motor as the motor induced electricity.On the other hand, use square wave electric current or quasi-retangular wave electric current or use the control of induced voltage as the square-wave voltage or the square-wave voltage that is as the criterion, compare with the control of using sine-wave current or sine voltage, have the shortcoming that is difficult to reduce torque ripple.
Fig. 6 represents an example of the motor current waveform controlled according to electric current (Id).Fig. 6 (A) expression motor 1 is in lower speed rotation and not according to situation (Idref=0) motor current waveform down of the weakened field control of electric current (Id) control, the motor current waveform under the situation that the weakened field of Fig. 6 (B) expression motor 1 high speed rotating and with good grounds electric current (Id) control is controlled.Fig. 6 (A) is a motor current waveform, and the waveform of corresponding induced voltage is rectangle (trapezoidal) ripple shown in Fig. 7 (A).Relative with the waveform of the induced voltage of Fig. 7 (A), the actual current waveform when Id=0 is shown in Fig. 7 (B) (corresponding with Fig. 6 (A)), Id=10[A] time the actual current waveform be shown in Fig. 7 (C) (corresponding) with Fig. 6 (B).The square wave electric current of indication or square-wave voltage are with square wave (trapezoidal wave) is different completely in the present invention, but have the waveform of the such crest of the such recess of Fig. 6 (A) and Fig. 7 (B) or Fig. 6 (B) and Fig. 7 (C), or comprise the waveform of such current waveform of Fig. 7 (A) (quasi-retangular wave electric current) or voltage waveform (quasi-retangular wave voltage).
Motor of the present invention by n (=2,3,4 ...) curtage of subharmonic drives, the frequency of nth harmonic is smaller or equal to the higher limit (for example 1000Hz) of the response frequency of Current Control.Promptly, induction voltage waveform at motor is square wave or quasi-retangular wave, harmonic components in the time of will carrying out frequency analysis to square wave or quasi-retangular wave is made as under the situation of n (=2,3,4...), with 5% the harmonic components n of following formula (3) expression more than or equal to the amplitude composition:
The higher limit of the response frequency of n * P/2 * ω≤Current Control ... (3)
Wherein P is a number of poles, and ω is an actual speed.
In this case, angular transducer is set, provides current waveform with the function of the induction voltage waveform of square wave or quasi-retangular wave at least.Can make the relevant electrical time constant of motor more than or equal to control cycle, the angle estimation unit can be set, provide motor current waveform with estimation angle from this angle estimation unit.
For 5% harmonic components n, utilize the reasons are as follows that above-mentioned formula (3) sets more than or equal to the amplitude composition.When the harmonic components n that can not respond when current control division reached current instruction value, it showed as the torque ripple of motor.As everyone knows, if the torque ripple of motor is in 10%, then because moment controlling system and can be by steering wheel perception No. the 3298006th, patent (not for example (B2)).Therefore, can determine back electromotive force harmonic components so that torque ripple smaller or equal to 10% of current value (torque).Can't obtain the relation of the harmonic components that comprises in back electromotive force and the electric current by the mode of vector control (perhaps accurate vector control) uniquely, but method is as can be known by experiment: if harmonic components smaller or equal to 5% of amplitude composition, then torque ripple is smaller or equal to 10% of current value (torque).
In addition, in driven steering device, carry out the PWM control of 20KHz usually, still, when frequency was lower than 20KHz, motor noise became problem, when frequency is higher than 20KHz, produced the problem of electromagnetic radiation noise and heating.This by performance as the FET of driver element about, in the PWM of 20KHz control, 1/20 1000Hz is the higher limit of the response frequency of Current Control, in the PWM of 40KHz control, 1/20 2000Hz is the higher limit of the response frequency of Current Control.
For the motor (number of poles P) of this specific character, constitute motor drive control device as shown in Figure 8 in the present invention.That is, motor drive control device of the present invention has: vector control phase current command value is calculated portion 20; According to the current instruction value Iavref, the Ibvref that calculate portion 20 from vector control phase current command value, Icvref with from electric machine phase current Ia, the Ib of current detection circuit 32-1,32-2,32-3, subtraction circuit 20-1,20-2, the 20-3 that Ic obtains each phase current error; Carry out the PI control part 21 of proportional plus integral control.By the PWM control of PWM control part 30, supply with each phase instruction current from inverter 31 to motor 1, the rotation of control motor 1 drives.The represented regional A of with dashed lines constitutes current control division.
In the present embodiment, calculate in the circuit 20 in vector control phase command value, after the current instruction value that utilizes the excellent specific property decision vector control d of vector control, q composition, convert this current instruction value to each phase current command value, and, in FEEDBACK CONTROL portion with entirely mutually control rather than d, q control come closed vector control phase command value to calculate circuit 20.Therefore, owing to utilized the theory of vector control in the stage of calculating current instruction value, so this control mode is called accurate vector control (Pseudo Vector Control is called " PVC control " below).
In addition, the following formation of current control division A of present embodiment: subtraction circuit 20-1, the 20-2, the 20-3 that obtain each phase current error according to each phase current command value Iavref, Ibvref, Icvref and electric machine phase current Ia, Ib, the Ic of motor 1; With with this each phase current error as the input PI control part 21.In addition, between inverter 30 and motor 1, be equipped with current detection circuit 32-1,32-2,32-3, formed each phase current Ia, Ib of current detection circuit 32-1,32-2, the detected motor of 32-3, the feedback circuit B that Ic offers subtraction circuit 20-1,20-2,20-3 as motor current detecting circuit.
In addition, vector control phase current instruction is calculated portion 20 and is had: the conversion portion 101 of calculating portion as each phase back-emf voltage; Calculate 3 phase transformation portions 102 mutually/2 of portion as d axle and q shaft voltage; Calculate the q axle instruction current of the current instruction value Iqref of q axle and calculate portion 103; Calculate the 2 phases/3 phase transformation portions 104 of portion as each phase current instruction; Calculate the d axle instruction current of the current instruction value Idref of d axle and calculate portion 105; Conversion portion 106 by the reference angle speed omega b of torque instruction value Tref conversion motor.The torque instruction value Tref that portion 20 receives the rotor position detection signal and determines according to the not shown torque that torque sensor detected is calculated in the instruction of vector control phase current, calculate the phase instruction value signal of vector control, wherein the rotor position detection signal comprises: the anglec of rotation θ e of the rotor 7 that is detected by rotor position detectors such as decomposer 11 and by calculate the electric angle speed omega e that anglec of rotation θ e obtains in differential circuit 24.Rotor position detector 11 has the function as angular transducer, and it can be replaced as the angle estimation unit.
Torque instruction value Tref is inputed to q axle instruction current to calculate portion 103, conversion portion 106 and d axle instruction current and calculates portion 105, anglec of rotation θ e is inputed to conversion portion 101,3 phases/2 phase transformation portions 102 and 2 phase transformation portions 104 mutually/3, with electric angle speed omega e input to conversion portion 101, q axle instruction current calculates portion 103 and d axle instruction current is calculated portion 105.
In the structure of the motor drive control device of using such PVC control, following drive controlling of carrying out motor 1.
At first, the anglec of rotation θ e and the electric angle speed omega e that calculate in the portion 20 rotor 7 in vector control phase current command value input to conversion portion 101, back electromotive force ea, the eb, the ec that calculate each phase according to the conversion table that is kept in the conversion portion 101.Back electromotive force ea, eb, ec are the square wave or the quasi-retangular wave of nth harmonic, and the frequency of nth harmonic is for multiply by the electric angle speed of motor the value of n gained.When the actual speed with motor was made as ω, the electric angle speed of motor was represented with P/2 * ω.Next, utilize the 3 phases/2 phase transformation portions 102 of the portion of calculating as d-q voltage, back electromotive force ea, eb, ec are converted to the voltage ed and the eq of d axle and q axle composition according to following (4) and (5) formula.
Next, the computational methods as the current instruction value Idref of the d axle of emphasis of the present invention are described.
Will from the reference angle speed omega b of conversion portion 106, from the electric angle speed omega e of differential circuit 24, from the torque instruction value Tref of torque sensor as input, calculate in the portion 105 at d axle instruction current and to calculate d shaft current command value Idref according to following (6) formula.Wherein, Kt is a moment coefficient, and ω b is the reference angle speed of motor, and reference angle speed omega b obtains in conversion portion 106 as input with torque instruction value Tref.
Idref＝-|Tref/Kt|·sin(acos(ωb/ωm))...(6)
About one of the acos (ω b/ ω m) of above-mentioned formula (6), rotary speed at motor is not under the situation of high speed rotating, and promptly the mechanical angle speed omega m at motor 1 is lower than under the situation of reference angle speed omega b, because ω m＜ω b, therefore acos (ω b/ ω m)=0, so Idref=0.But when high speed rotating, when promptly mechanical angle speed omega m was higher than reference angle speed omega b, the value of current instruction value Idref occurred, the control of beginning weakened field.Described suc as formula (6), owing to current instruction value Idref changes according to the rotary speed of motor 1, therefore have good result that can not have the control when carrying out high speed rotating reposefully with disconnecting.
In addition, as another effect, also produce effect at the saturation problem of motor terminal voltage.The phase voltage V of motor generally uses
V＝E+R·I+L(di/dt) ...(7)
Expression.Here, E is a back electromotive force, and R is a fixed resistance, and L is an induction coefficient, motor more high speed rotating then back electromotive force E is big more because supply voltage such as cell voltage fixes, so available voltage range diminishes in the control of motor.The angular speed that reaches this voltage saturation is reference angle speed omega b, and when producing voltage saturation, the duty ratio of PWM control reaches 100%, follow current command value again, and its result, it is big that torque ripple becomes.
But the polarity of the current instruction value Idref that formula (6) is represented is for negative, and the induced voltage composition of the current instruction value Idref relevant with the L (di/dt) of above-mentioned formula (6) is opposite with the polarity of back electromotive force E.Therefore, shown and utilized the effect that reduces the high speed rotating value is big more more back electromotive force E by the voltage of current instruction value Idref induction.Its result, though motor 1 high speed rotating, also can be by the effect of current instruction value Idref, increase can be controlled the voltage range of motor.That is, following effect is arranged: the weakened field that control the realized control by current instruction value Idref, make that the control voltage of motor can be unsaturated, it is big that the scope that can control becomes, and can prevent also during the motor high speed rotating that torque ripple from increasing.
Fig. 9 is the functional module structure of the Circuits System relevant with calculating of above-mentioned current instruction value Idref.In Fig. 9, torque instruction value Tref is inputed to conversion portion 106 and the 105d of moment coefficient portion, the electric angle speed omega e of motor is inputed to mechanical angle calculate the 105a of portion.Mechanical angle is calculated the 105a of portion and is calculated the mechanical angle speed omega m (=ω e/P) of motor by the electric angle speed omega e of motor, and inputs to acos and calculate the 105b of portion.In addition, conversion portion 106 is converted into reference angle speed omega b with torque instruction value Tref, and inputs to acos and calculate the 105b of portion, the 105d of moment coefficient portion with torque instruction value Tref be converted into coefficient Iqb (=Tref/Kt) and input to the 105e of absolute value portion.Acos calculates mechanical angle speed omega m and the reference angle speed omega b of the 105b of portion according to input, calculates advance angle Φ=acos (ω b/ ω m), and inputs to sin and calculate the 105c of portion.Sin calculates the 105c of portion and obtains sin Φ according to the advance angle Φ that is imported, and input to-1 times multiplier 105f, multiplier 105f will calculate the advance angle Φ of the 105c of portion and from the absolute value of the 105e of absolute value portion from sin | and Iqb| multiplies each other, and multiply by-1 again, obtains current instruction value Idref.Calculate current instruction value Idref according to following formula (8), this is the output that d axle instruction current is calculated portion 105.
Idref＝-|Iqb|×sin(acos(ωb/ωm)) ...(8)
Q axle instruction current be will input to according to the current instruction value Idref that calculate above-mentioned formula (8) and portion 103 and 2 phase transformation portions 104 mutually/3 calculated.
On the other hand, calculate in the portion 103 at q axle instruction current, (the current instruction value Iderf of=ω m * P), d axle according to following formula (9) and the represented motor output equation formula of formula (10), calculates the current instruction value Iqref of q axle based on 2 phase voltage ed and eq, electric angle speed omega e.That is, motor output equation formula is
Tref×ωm＝3/2(ed×Id+eq×Iq) ...(9)
Therefore, when with Id=Idref, during this formula of Iq=Iqref substitution (9), then become
Iqref＝2/3(Tref×ωm-ed×Iderf)/eq ...(10)
In addition, can be with the value substitution current instruction value Idref that utilizes formula (8) to calculate.
As the formula (10), be equivalent to derive the such motor output equation formula of power, therefore can easily calculate current instruction value Iqref because current instruction value Iqref is output from motor.In addition, can calculate the optimum current command value Iqref that has obtained balance with the current instruction value Idref that is used to obtain essential command torque Tref.Even therefore when the motor high speed rotating, the terminal voltage of motor is also unsaturated, can control torque ripple minimum.
If the relation of diagram current instruction value Idref of the present invention discussed above and Iqfef is then shown in Figure 10 (A).Relation under Figure 10 (B) expression advance angle control mode situation in the past.
Current instruction value Idref and Iqref are inputed to 2 phase transformation portions 104 mutually/3 of the portion of calculating as each phase current command value, convert current instruction value Iavref, Ibvref, the Icvref of each phase to.That is, suc as formula shown in (12) and the formula (13).Here, subscript, for example a phase current command value that determines by vector control of current instruction value Iavref " avref " expression.In addition, determinant C2 is represented suc as formula (13), is the constant by the anglec of rotation θ e decision of motor.
Used current instruction value Iqref and advance angle Φ in the past, current instruction value Iavref, Ibvref, Icvref calculate in 2 phases/3 phase transformation portions 109 by Fig. 1, and in the present invention, as mentioned above, current instruction value Idref and Iqref as input, are calculated current instruction value Iavref, Ibvref, Icvref by 2 phases/3 phase transformation portions 104.And subtraction circuit 20-1,20-2,20-3 subtract each other each phase current Ia, Ib, Ic and current instruction value Iavref, Ibvref, the Icvref of current detection circuit 32-1,32-2, the detected motor of 32-3, calculate each error.Next, the error of PI control part 21 each phase current of control, calculate the command value of inverter 31, magnitude of voltage va, the vb, the vc that promptly represent the duty ratio of PWM control part 30, PWM control part 30 is according to this magnitude of voltage va, vb, vc, inverter 31 is carried out PWM control, and drive motors 1 thus, produces the torque of expection.
As described above such, motor of the present invention and driving control device thereof can not make the terminal voltage of motor saturated when the motor high speed rotating yet, can control torque ripple minimum.Therefore, applying the present invention to have following good result under the situation of driven steering device: can carry out reposefully and swerve, can not bring inharmonious sensations such as steering wheel vibration to the driver.
The present invention is different with the FEEDBACK CONTROL of passing through d, q control realization of conventional art, and it is different fully only to carry out this point in FEEDBACK CONTROL in each is controlled mutually.Its result, in the prior art, in the process of carrying out the FEEDBACK CONTROL that realizes by in the past d, q control, the non-linear factor that a produces mutually be diffused into b, c each mutually in, existence can not be carried out the problem of correct correction control, and in the present invention, and the non-linear factor of a phase only carries out FEEDBACK CONTROL at a in mutually, can not be diffused into b phase, c mutually in, therefore can correctly proofread and correct control.
By using such PVC control, the non-linear factor that can comprise in will controlling is separated under the state of each phase and carries out Electric Machine Control, and its result can realize the Electric Machine Control that torque ripple is little, noise is little.Thus, be applied under the situation of driven steering device, can be implemented in when stopping and during emergency turn noise little and steadily, the little steering operation of vibration.
In addition, in the above-described embodiments, phase voltage ea, eb, ec have been used, even but be converted into line voltage eab, ebc, eca control, and also can obtain identical result.
As mentioned above, according to motor of the present invention, following effect is arranged: electricity when motor rotates at a high speed The terminal voltage of machine can be unsaturated yet, and the torque fluctuation is little and motor noise is little, and, at electricity In the moving power steering gear, has following good result: can provide a kind of direction of steadily following Swerving of dish, thus inharmonic sensation in steering operation, do not had, electronic the moving that noise is little Power turns to device.
And, according to driven steering device of the present invention, calculate each mutually electricity based on vector controlled Stream command value, Current Feedback Control are used the PVC control of controlling respectively each phase, can provide thus A kind ofly brushless DC motor can be controlled to the electricity that small-sized, torque fluctuation is little, motor noise is little Machine drives control device, and the electric motor driven power steering that a kind of steering operation is steady, noise is little can be provided Device.
And, according to motor of the present invention, because the frequency of nth harmonic is smaller or equal to Current Control The response frequency on limit value, therefore namely use square wave electric current or accurate square wave electric current or rectangle Wave voltage or accurate square wave voltage drive, and also can obtain torque fluctuation and diminish, and is small-sized and make an uproar The motor that sound is little.
According to the present invention, the terminal voltage of motor can be unsaturated yet when the high speed rotation of motor, turns to The square fluctuation is little, motor noise is little, if therefore be applied to driven steering device, then can provide A kind ofly steadily follow swerving of steering wheel, do not have inharmonious sensation in the steering operation, noise Little driven steering device.
In addition, according to driven steering device of the present invention, calculate each mutually electricity based on vector controlled Stream command value, Current Feedback Control are used each the mutually respectively PVC of control control, can provide thus A kind ofly brushless DC motor can be controlled to the electricity that small-sized, torque fluctuation is little, motor noise is little Machine drives control device, and the electric motor driven power steering that a kind of steering operation is steady, noise is little can be provided Device.
Claims (5)
1. motor drive control device that the motor with 3 phases is controlled, the electric current of wherein said motor or induction voltage waveform are square wave or the quasi-retangular wave with nth harmonic, n=2,3,4 ... it is characterized in that this motor drive control device has:
D-q voltage is calculated portion, and it is calculated respectively as the d axle of back electromotive force and the voltage ed and the voltage eq of q axle composition;
D axle instruction current is calculated portion, be higher than at the mechanical angle speed omega m of detected described motor under the situation of reference angle speed omega b of described motor, this d axle instruction current is calculated portion and is calculated current instruction value Idref as the d axle composition of current instruction value according to the mechanical angle speed omega m of the reference angle speed omega b of the torque instruction value Tref of described motor, described motor and detected described motor;
Q axle instruction current is calculated portion, and it is according to described voltage ed, and voltage eq and current instruction value Idref calculate the current instruction value Iqref as the q axle composition of current instruction value;
Current control division, it controls the electric current of described motor according to described current instruction value Idref and described current instruction value Iqref.
2. motor drive control device according to claim 1, it is characterized in that, will calculate described current instruction value Iqref in described mechanical angle speed omega m, voltage ed, voltage eq, torque instruction value Tref and the current instruction value Idref substitution motor output equation formula.
3. motor drive control device according to claim 1 and 2, wherein said motor is a brushless DC motor.
4. a driven steering device is characterized in that, has used claim 1 or 2 described motor drive control device.
5. a driven steering device is characterized in that, has used the described motor drive control device of claim 3.
Applications Claiming Priority (4)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2002345135 | 2002-11-28 | ||
| JP2002345135 | 2002-11-28 | ||
| JP2002354632 | 2002-12-06 | ||
| JP2003376428 | 2003-11-06 |
Related Parent Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CNB2003801045194A Division CN100369375C (en) | 2002-11-28 | 2003-11-27 | Motor and drive control device therefor |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CN101136607A true CN101136607A (en) | 2008-03-05 |
Family
ID=35822599
Family Applications (3)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CNA200710146567XA Pending CN101136607A (en) | 2002-11-28 | 2003-11-27 | Motor and drive control device therefor |
| CNA2007101465684A Pending CN101102090A (en) | 2002-11-28 | 2003-11-27 | Brushless DC motor |
| CNB2003801045194A Expired - Fee Related CN100369375C (en) | 2002-11-28 | 2003-11-27 | Motor and drive control device therefor |
Family Applications After (2)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CNA2007101465684A Pending CN101102090A (en) | 2002-11-28 | 2003-11-27 | Brushless DC motor |
| CNB2003801045194A Expired - Fee Related CN100369375C (en) | 2002-11-28 | 2003-11-27 | Motor and drive control device therefor |
Country Status (1)
| Country | Link |
|---|---|
| CN (3) | CN101136607A (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110754037A (en) * | 2018-05-11 | 2020-02-04 | 日本精工株式会社 | Motor control device and electric power steering device provided with same |
Families Citing this family (15)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP4862043B2 (en) * | 2006-05-26 | 2012-01-25 | 三菱電機株式会社 | Electric power steering device |
| CN101396976B (en) * | 2007-09-25 | 2012-01-04 | 奇瑞汽车股份有限公司 | Electric machine control method and device in hybrid motor |
| CN101557193B (en) * | 2009-04-22 | 2011-01-19 | 华中科技大学 | Vector control method of non-sinusoidal counter-electromotive-force surface AC magnetoelectric machine |
| JP5494662B2 (en) * | 2009-07-24 | 2014-05-21 | トヨタ自動車株式会社 | Rotation angle detector |
| DE102009045822A1 (en) * | 2009-10-20 | 2011-04-28 | Robert Bosch Gmbh | Electronically commutated electric motor with calibrated motor torque constant |
| GB201010443D0 (en) * | 2010-06-22 | 2010-08-04 | Aeristech Ltd | Controller |
| EP2479886B1 (en) * | 2011-01-19 | 2016-11-02 | KIH-utveckling AB | Method for controlling operation of an electric motor in a height-adjustable furniture arrangement |
| JP2013183532A (en) * | 2012-03-01 | 2013-09-12 | Toshiba Corp | Motor control device and control method of the same |
| JP5839111B2 (en) * | 2012-03-22 | 2016-01-06 | 日産自動車株式会社 | Control device for three-phase AC induction motor and control method for three-phase AC induction motor |
| US8896245B2 (en) * | 2012-03-26 | 2014-11-25 | Gm Global Technology Operations Llc. | Methods, systems and apparatus for generating voltage command signals for controlling operation of an electric machine |
| DE102012205728A1 (en) | 2012-04-05 | 2013-10-10 | Robert Bosch Gmbh | Method and device for the electrodynamic braking of a universal motor |
| US9755567B2 (en) * | 2015-11-04 | 2017-09-05 | GM Global Technology Operations LLC | Determination of permanent magnetic flux in an electric machine |
| US10090788B2 (en) * | 2016-03-03 | 2018-10-02 | Robert Bosch Gmbh | Optimal torque ripple reduction through current shaping |
| US10184860B2 (en) * | 2016-04-08 | 2019-01-22 | Infineon Technologies Ag | Control system for power train control |
| CN105959518B (en) * | 2016-05-20 | 2019-02-05 | Oppo广东移动通信有限公司 | Motor control method, device and picture pick-up device |
Family Cites Families (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US5670854A (en) * | 1994-12-14 | 1997-09-23 | Matsushita Electric Industrial Co., Ltd. | Control system for an induction motor |
| JP3137560B2 (en) * | 1995-05-29 | 2001-02-26 | トヨタ自動車株式会社 | Synchronous motor controller |
| JPH1118471A (en) * | 1997-06-17 | 1999-01-22 | Okuma Mach Works Ltd | Controller for synchronous motor |
| JP3396440B2 (en) * | 1999-02-08 | 2003-04-14 | 株式会社日立製作所 | Control device for synchronous motor |
| JP3624737B2 (en) * | 1999-03-23 | 2005-03-02 | 豊田工機株式会社 | Motor control device |
| JP2001103783A (en) * | 1999-09-30 | 2001-04-13 | Toyoda Mach Works Ltd | Motor controller |
-
2003
- 2003-11-27 CN CNA200710146567XA patent/CN101136607A/en active Pending
- 2003-11-27 CN CNA2007101465684A patent/CN101102090A/en active Pending
- 2003-11-27 CN CNB2003801045194A patent/CN100369375C/en not_active Expired - Fee Related
Cited By (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110754037A (en) * | 2018-05-11 | 2020-02-04 | 日本精工株式会社 | Motor control device and electric power steering device provided with same |
| CN110754037B (en) * | 2018-05-11 | 2022-09-16 | 日本精工株式会社 | Motor control device and electric power steering device provided with same |
Also Published As
| Publication number | Publication date |
|---|---|
| CN1717860A (en) | 2006-01-04 |
| CN101102090A (en) | 2008-01-09 |
| CN100369375C (en) | 2008-02-13 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| EP1587210B1 (en) | Motor and drive control device therefor | |
| JP5310815B2 (en) | Motor drive control device and electric power steering device using the same | |
| JP4624619B2 (en) | Method and apparatus for controlling torque in a permanent magnet brushless electronic motor | |
| JP4039317B2 (en) | Electric power steering device | |
| CN101136607A (en) | Motor and drive control device therefor | |
| US6900613B2 (en) | Motor control apparatus | |
| EP1583217B1 (en) | Motor drive-controlling device and electric power-steering device | |
| EP2048061A1 (en) | Electric power steering system | |
| Lee et al. | Torque ripple minimization control of permanent magnet synchronous motors for EPS applications | |
| US20040012367A1 (en) | Apparatus for controlling brushless motor | |
| JP3804686B2 (en) | Motor drive control device and electric power steering device | |
| JP4103430B2 (en) | Electric power steering device | |
| CN117178471A (en) | Motor control device and motor control method | |
| JP2006217795A (en) | Motor and its drive control device | |
| US20050263330A1 (en) | Field-oriented control for brushless DC motor | |
| JP2006136144A (en) | Non-connection type motor, drive controller thereof and electric power steering device using drive controller for the non-connection type motor | |
| WO2022190582A1 (en) | Motor drive device | |
| JP2000217384A (en) | Controller of position-sensorless motor | |
| JPS61132096A (en) | Controller for ac motor | |
| JP4615440B2 (en) | Motor drive control device and electric power steering device | |
| JP2001054297A (en) | Motor control unit and control method | |
| JP2000232798A (en) | Control method for synchronous motor |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| C06 | Publication | ||
| PB01 | Publication | ||
| C10 | Entry into substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| C02 | Deemed withdrawal of patent application after publication (patent law 2001) | ||
| WD01 | Invention patent application deemed withdrawn after publication |
Open date: 20080305 |