CN101123412A - Integrated method for vector control of induction electromotor frequency conversion under voltage and direct toque control - Google Patents
Integrated method for vector control of induction electromotor frequency conversion under voltage and direct toque control Download PDFInfo
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Abstract
The invention relates to a comprehensive method of vector control and direct torsion control of frequency conversion voltage regulation of induction motors. The invention is primarily to solve the technical difficulty of prior vector control method that time response velocity fails to meet the control requirement of the system. According to the technical proposal of the invention, the comprehensive method comprises the following steps. Firstly, a vector control mathematical model and a difference equation of induction motors are established; expectation values are got through calculation of the difference equation; the difference between the expectation value and the value detected in real time is the control quantity; the control quantity controls a switch of an isolated gate bipolar triode (IGBT) so as to complete combination control of the magnetic field and the torsion. Secondly, a direct torsion control model of induction motors is established; amplitude value of the torsion and size and direction of a rotating magnetic field of an induction motor are calculated in real time; the calculated values are compared with expected torsion and flux to get the difference to adjust the interval controlled and limited by the direct torsion, and to offset lag in response of vector control in nonconstant torsion or nonlinear operation area.
Description
Technical Field
The invention relates to a vector control and direct torque control comprehensive method for frequency conversion and voltage regulation of an induction motor.
Background
As is known, the vector control method of an induction ac motor basically adopts a control concept of dc-converting an ac motor, i.e. decomposing a stator current of the ac motor into i-axis currents stationary relative to a rotor for controlling a resultant magnetic field d And i of q-axis current for torque control q . I.e. the stator current as a control variable. The resultant magnetic flux is controlled by controlling the stator current excitation component and the electromagnetic torque is controlled by the stator current torque component. For this purpose, the coordinate system calculation for converting the ac current U, V, W system into the dc current d, q system is performed in real time. Since the vector control method needs to continuously perform relatively complicated coordinate system transformation, the time response is feasible when the motor or the system runs in a linear segment. However, when the system has impact load or under the influence of transient fluctuation of the motor, the vector control method cannot meet the control requirement of the system on the time response speed.
Disclosure of Invention
The invention aims to solve the technical difficulties and provide a comprehensive method for vector control and direct torque control of variable-frequency voltage regulation of an induction motor by combining vector control and direct torque control.
The technical scheme adopted by the invention for solving the technical difficulties is as follows: the comprehensive method of vector control and direct torque control for frequency conversion and voltage regulation of induction motor includes the following steps:
a) Establishing vector control mathematical model and difference equation of induction motor, and implementing the method according to the following difference equationConverting the three-phase U, V and W alternating current and voltage of the stator to the d axis and the control torque q axis of the synthetic magnetic field, and calculating to obtain a desired value i d(t) 、i q(t) Expected value and i obtained by real-time detection 1d(t) 、i 1q(t) Subtracting the difference value to obtain a Pulse Width Modulation (PWM) regulation control quantity, and controlling the switch of an Insulated Gate Bipolar Transistor (IGBT) by the control quantity so as to complete the comprehensive control of the magnetic field and the torque under the linear running state that the constant torque or the detection voltage/the motor stator frequency (V/f) is equal to a constant;
in the formula: i.e. i d(t) 、i q(t) Is the calculated expected value; I.C. A 1 Is the stator current; theta x Is the angular displacement of the rotor advanced by the rotating magnetic fieldω x The slip ratio of the rotating magnetic field leading rotor of the induction motor can be obtained by the following formula:
L 2 ＝l 2 +M，l 2 is the self-inductance of the rotor of the motor, M is the mutual inductance between the rotor and the stator, r 2 Is the resistance of the rotor circuit and,
in the formula: omega r ＝ω s -ω s ω x ，ω s Is the speed of the rotating magnetic field, ω s =2 π f, according to ω f, in constant torque operation s Obtaining a relation proportional to the voltage; omega r Is the angular velocity of the rotor rotation;
b) Establishing direct torque control model of induction motor, and calculating torque T of induction motor in real time by the following formula m And a rotating magnetic field of \58388 s Magnitude and direction, and desired torqueComparing with expected magnetic flux to obtain a difference value which is a Pulse Width Modulation (PWM) regulation control quantity, and controlling the on-off of an Insulated Gate Bipolar Transistor (IGBT) by the control quantity so as to complete interval regulation limited by direct torque control and compensate the reaction hysteresis of vector control in a non-constant torque or non-linear operation region;
P r ＝E 2N I 1 cosθ s (4)
in the formula: p r Is the power transmitted by the stator to each phase of the rotor; n is a radical of an alkyl radical s Is the rotational speed of the rotor per minute, which can be based on ω, in constant torque operation s Calculating; e 2N Is prepared from \58388 s The induced voltage; i is 1 Is the stator current; theta.theta. s Is the power factor angle of the motor, and can acquire the stator voltage E and the current I in real time 1 Obtaining; voltage E 2N By measuring E at the stator end 1N Then subtract I 1 r 1 Voltage drop over the capacitor;
in the formula: \58388 s Is a resultant rotating magnetic field generated by the stator and the rotor together; omega r Is the angular velocity of rotation of the rotor, which may be determined by
Calculating; e d Is a direct current power supply before PWM inversion; k is a constant, depending on the physical structure of the motor;
c) The power factor is adjusted by controlling the Insulated Gate Bipolar Transistor (IGBT) switches to adjust the reactive power distribution.
Because the invention adopts the technical scheme, compared with the background technology, the invention has the following effects:
(1) The control method of the invention keeps the stability, continuity, accuracy and reliability of the traditional vector control, and particularly can display the superior functions of the motor when the motor runs in a constant torque state.
As known from electromechanics, when the motor runs in a constant torque state, the relative speed of the combined rotating magnetic field and the rotor is constant, namely the magnetic flux relatively cut between the rotating magnetic field and the rotor is constant, namely \58388isconstant. In addition, the voltage of the stator of the motor is V =4.44fN \58388
Wherein f is the system or motor frequency, N is the number of turns of each phase of the stator winding, v ^ f when the magnetic flux is 58388and the constant number is v ^ f,
ω is the motor angular frequency, ω =2 π f,
n is the motor rotor rotational speed.
Where p is the number of N, S poles,
f∝ω∝n。
therefore, when the stator voltage is operated at a constant torque, v ^ f ^ ω ^ n is determined.
(2) The control method of the invention adopts the technical scheme of combining direct torque control and vector control, so when the motor runs deviating from constant torque or runs in a nonlinear way, the control method avoids the complicated transformation of vector control. For any non-linear V/f relationship, direct control of torque and field can be used to respond rapidly to changes in the motor and system.
(3) Since the core of the direct torque control method is to control the torque and the magnetic field, the control method can compensate the problem of magnetic field lag in low-frequency operation by quickly adjusting the magnetic field in low-frequency operation. Therefore, the frequency converter can stably run even when the frequency approaches zero.
(4) The control method of the invention also changes the distribution of the reactive power of the system by adjusting the magnetic flux and controlling the switch, thereby continuously adjusting the power factor of the system and ensuring that the power factor meets the power factor requirements under various load states.
(5) Due to the combination of the control method and the topological structure of the main circuit of the switch, the frequency converter can output a perfect sine wave without harmonic waves. The waveform can meet the requirement of IEEE 519-1992 on the Total Harmonic Distortion (THD) index.
(6) From omega r ＝ω s -ω s ω x The invention adopts a control loop without a rotating speed sensor, so that the control method has the advantages of high control speed, simple hardware equipment and easy control in the control process.
Drawings
FIG. 1 is a block schematic diagram of the control method of the present invention;
FIG. 2 is a single phase equivalent circuit diagram of a three phase induction motor;
FIG. 3 is a vector diagram of current and voltage for an induction motor stator circuit;
FIG. 4 shows a DC voltage E controlled by PWM irregularities d A circuit for converting AC current to AC current A, B and C;
FIG. 5 is magnetic flux \58388 s A hexagonal movement trajectory diagram;
FIG. 6 is a schematic view of the flux operation with flux control intervals and flux zero indications;
FIG. 7 is a magnetic flux \58388directtorque control method s And torque T m Schematic of the interaction of (a);
FIG. 8 is a circuit diagram of a current voltage with PWM technology controlling IGBT switches to provide reactive power to an inductive load to produce a capacitive fundamental current to vary power factor;
fig. 9 is a waveform diagram of current and voltage for changing power factor by controlling the IGBT switch to supply reactive power to inductive load to generate capacitive fundamental current by PWM technique.
Detailed Description
The present invention will be described in further detail with reference to the drawings and examples.
As shown in fig. 1, the method for integrating vector control and direct torque control of frequency conversion and voltage regulation of an induction motor in the present embodiment includes the following steps:
a) Establishing vector control mathematical model and differential equation of induction motor, and implementing d-axis and control torque q-axis of stator three-phase U, V and W alternating current and voltage to synthetic magnetic field according to the following differential equationConverted and calculated to a desired value i d(t) 、i q(t) Expected value and i obtained by real-time detection 1d(t) 、i 1q(t) Subtracting the difference value to obtain a Pulse Width Modulation (PWM) regulation control quantity, and controlling the switch of an Insulated Gate Bipolar Transistor (IGBT) by the control quantity so as to complete the comprehensive control of the magnetic field and the torque under the linear running state that the constant torque or the detection voltage/the motor stator frequency (V/f) is equal to a constant;
in the formula: id(t) 、i q(t) is the calculated expected value; i is 1 Is the stator current; theta x Is the angular displacement of the rotating magnetic field leading the rotorω x The slip ratio of the rotating magnetic field leading rotor of the induction motor can be obtained by the following formula:
L 2 ＝l 2 +M，l 2 is the self-inductance of the rotor of the motor, M is the mutual inductance between the rotor and the stator, r 2 Is the resistance of the rotor circuit and,
in the formula: omega r ＝ω s -ω s ω x ，ω s Is the speed of the rotating magnetic field, ω s =2 π f, according to ω f, in constant torque operation s Obtaining a relation proportional to the voltage; omega r Is the angular velocity of the rotor rotation;
in FIG. 1
Is a flux compensation ring. Because when the magnetic flux changes, even let i d The change cannot immediately follow the change in the magnetic flux, and in order to compensate for this hysteresis component, it is necessary to add a d-axis current component i proportional to the rate of change in the magnetic flux d 。
b) Establishing direct torque control model of induction motor, and calculating torque T of induction motor in real time by the following formula m And a rotating magnetic field of \58388 s The magnitude and direction are compared with the expected torque and the expected magnetic flux to obtain a difference value which is a Pulse Width Modulation (PWM) regulation control quantity, and the control quantity controls the on-off of an Insulated Gate Bipolar Transistor (IGBT), so that the interval regulation limited by direct torque control is completed, and the reaction hysteresis quality of vector control in a non-constant torque or non-linear operation area is compensated;
P r ＝E 2N I 1 cosθ s (4)
in the formula: p r Is the power transmitted by the stator to each phase of the rotor; n is s Is the rotational speed of the rotor per minute, which can be based on ω in constant torque operation s Calculating; e 2N Is prepared from \58388 s The induced voltage; i is 1 Is the stator current; theta.theta. s Is the power factor angle of the motor, and can acquire the stator voltage E and the current I in real time 1 Obtaining; voltage E 2N By measuring E at the stator end 1N Then subtract I 1 r 1 Voltage drop over the capacitor;
in the formula: \58388 s Is a resultant rotating magnetic field generated by the stator and the rotor together; omega r Is the angular velocity of rotation of the rotor, which may be determined by
Solving; e d Is a direct current power supply before PWM inversion; k is a constant, depending on the physical structure of the motor;
c) The power factor is adjusted by controlling the Insulated Gate Bipolar Transistor (IGBT) switches to adjust the reactive power distribution.
As shown in FIG. 2, each element in FIG. 2 is represented as a stator resistor r 1 Stator leakage flux \58388 1 The mutual inductance magnetic flux between the stator and the rotor is 58388and the leakage magnetic flux of the rotor is 58388 2 And a resistance r 2 S (which means that the rotor absorbs the active power transmitted from the stator), where S is the slip ratio, and can be determined by
And (6) obtaining.
R in FIG. 2 1 、x 1 (stator resistance, reactance), r 2 、x 2 (rotor resistance, reactance), x m (field winding reactance).
From electromechanics, total flux \58388infig. 3 s Equal to \58388 1 And, 58388a. Namely:
s ＝ 1 + (6)
similarly, the total torque represented by three-phase power may be represented by:
according to the electromechanics, n in the formula (3) s Is the rotational speed of the rotor per minute, which can be based on omega in constant torque operation s And (4) obtaining. And omega s As mentioned above, the speed of the rotating magnetic field, in constant torque operation, is in accordance with ω s Proportional to the voltage.
Formula medium power P r Is the active power absorbed between point 4 and N, which is identical to the active power flowing between points 2 and N. Because of the reactive component X 1 、X m No active power is consumed, so the following equation can be obtained:
P r ＝E 2N I 1 cosθ s (4)
in the formula:
P r is the power transmitted by the stator to each phase of the rotor.
E 2N Is prepared from \58388 s The induced voltage.
I 1 Is the stator current.
θ s Is the power factor angle of the motor and can acquire the stator voltage E and the current I in real time 1 And (4) obtaining.
Voltage E 2N Is not directly measurable, but its value is easily calculated by measuring E at the stator end 1N Then subtract I 1 r 1 The voltage drop over.
Magnetic flux \58388infig. 3 s Is a sum voltage E 2N Proportional to and 90 degrees behind it, and the stator current voltage and I are marked in the vector diagram of FIG. 3 1 r 1 The relationship between them. Can collect the stator current I in real time 1 Determines the power factor cos theta from the sum voltage E s (ignoring I here) 1 r 1 ). Thus, equations (3) and (4), i.e., the important parameter T in the direct torque control method, can be obtained m The solution is obtained.
Another important parameter in the direct torque control method \58388 s The following equation can be used to solve:
as shown in fig. 1:
s the resultant rotating magnetic field is generated by the stator and rotor together.
ω r Is the angular velocity of the rotor rotation. From the foregoing, can be
And (4) obtaining. Only when the motor runs at a lower frequency or the load fluctuates to cause a weak magnetic area deviating from constant torque, the voltage and the rotating speed omega of the motor s Is in non-linear relation, but can be processed in linear relation in a very small time interval, thereby achieving the purpose of quickly estimating omega s And ω r The purpose of (1).
E d Is a direct current power supply before PWM inversion.
k is a constant, depending on the physical structure of the motor. Such as the type of motor and the number of turns per N, S electrode of the stator.
Since \58388 s Is a vector, and is represented by formula (7) \58388 s The values are scalar, facilitating real-time tracking in the direct torque control method, \58388 s May be calculated in the vector diagram of fig. 3, \58388 s And (5) calculating the direction at any time.
As shown in fig. 4, this is a voltage controlled by a direct current E d Applying a DC voltage E by a random PWM method d And converting into three-phase A, B and C alternating-current voltage circuits. The position of the switches 1, 2, 3, 4, 5, 6 can be controlled at any time by controlling the PWM. Thereby generating a constantly changing magnetic flux \58388 s And torque T m 。
Magnetic flux \58388 s Can be controlled at the value of 58388 B ＜ s ＜ A Within the interval (c). Similarly, torque T m Can be controlled at T B ＜T m ＜T A Within the interval (c).
According to the six-pulse converter of fig. 4, different magnetic field directions can be obtained by different switch combinations. For example: for the direction of the magnetic field of A (+)Obtained by connecting the winding A to a DC power supply E d The positive terminal of (a). And connecting the windings B and C to E d The negative terminal of (a). And so on for others.
As shown in fig. 5, a resultant magnetic field of a three-phase induction motor at any time \58388 s Is controlled by switching the DC voltage E with PWM conversion as shown in FIG. 4 d After inversion, the magnetic fluxes generated by three-phase alternating current A, B and C windings are generated correspondingly. Through the winding and E d Can generate magnetic flux in six directions, A (+), A (-), B (+), B (-), C (+), C (-). Assuming that the stator flux of a three-phase machine is confined within the dashed circles, the flux in each direction is directed from the center of the pattern to a certain direction. To complete a magnetic flux rotation a single revolution at least six times the direction of the magnetic flux needs to be changed.
It is assumed that the starting direction of the magnetic flux is defined by the vector 0-1, i.e., A (+). By applying this combination of C (+) superimposed on 1 point, flux \58388 s Will travel from point 1 to point 2. Namely, magnetic flux \58388 s Move from point 1 to point 2.After reaching point 2, the superposition combination with A (-) is continued, then flux \58388 s Will continue to travel toward point 3. After reaching point 3, continuing to apply the B (+) magnetic flux combination to make the magnetic flux \58388 s Move to point 4 and so on.
As shown in fig. 6, a magnetic flux control interval is provided to control magnetic flux \58388 s The adjustment range of (2). The flux is regulated in the range of 1.0-1.1PU in this figure, the bold points in the figure indicating that the flux is frozen in or at zero in the air at a certain moment, i.e. at that moment, the speed at which the flux runs is zero. The direct reason for this is that the dc power E is supplied at a certain moment d And the connected three-phase alternating current winding coil is in short circuit. It is clear that the more freezing points, the slower the flux speed and vice versa, during a cycle of flux operation.
As shown in fig. 7, a magnetic flux control region \58388isprovided in this figure A 、 B And a torque control region T A 、 T B . Suppose magnetic flux \58388 s At a certain pointAt one time, is operating in the position shown in the figure. It is clear that \58388 s Specific minimum flux \58388 B Is small. To make \58388 s Entering the magnetic flux control region, the DC power supply E is required to be connected d The connection mode with the alternating current winding is controlled. As mentioned above, there are seven ways to control the magnetic flux trajectory, namely, A (+), A (-), B (+), B (-), C (+), C (-), and the special point, zero point. Thus, selection A (+) will produce a flux directed to the right of the horizontal, and selection C (+) will produce a flux at 120 to the horizontal, instantaneously connecting the three-phase stator winding to the DC power source E d A short circuit will create a zero and so on. It is important how to determine the trajectory of the magnetic flux travel for the purpose of controlling both flux and torque. Obviously, in the positions shown in the figure, a (-), B (+) and zero are not suitable choices because they would cause magnetic flux \58388 s Remain unchanged or become smaller. Further analysis, it is readily noted that A (+), B (-), C (+), C (-) are possible choices. However, how to determine the optimum choice depends on this instantaneous torque T m Of the position of (a). If T is m ＜T B C (+) was chosen because it can oppose the magnetic flux \58388 s A movement in a significant direction is generated which increases the torque of the motor. However, if T m ＞T A Then A (+) is selected. This option is firstly that the magnetic flux enters the regulation range, while it also has a braking effect on the motor torque. Finally, if T m Just at T A And T B Then B (-) is the best choice. Since it not only brings the magnetic flux into the control zone but also generates a slight torque in the direction of rotation to the motor. If T is m Is significantly greater than T A C (-) is selected, thereby generating a strong torque brake. If the slip S of the motor is large, the zero point can also be selected at this time to reduce the slip to approach the desired interval. No matter the magnetic flux \58388 s The above control method is applicable to any position of the magnetic flux trajectory circle.
It can be seen that T m 、 s Is made ofIs a measured value of T A 、T B 、 A 、 B Is a desired value of the control method.The allowable error value of the two is the control quantity of the corresponding IGBT switch in figure 1.
As shown in fig. 8, the circuit adjusts various loads, such as inductive or capacitive loads, to adjust the power factor. The IGBT power electronic switch shown in fig. 8 is on in the first half cycle and off in the second half cycle after the voltage zero crossing. Thus, the switch characteristic can be used to sense the inductive load Z L Reactive power is delivered, which is equivalent to having a power capacitor in series in the circuit. Thereby changing the reactive power distribution across the load and thereby adjusting the power factor of the load. In the same way, the power factor of the capacitive load circuit can be adjusted by using the concept that the IGBT in the first half period is turned off and the IGBT in the second half period is turned on. Thus, the control method of the present invention has powerful power factor regulating function no matter whether the load is in inductive or capacitive load state. The above processes are all realized by monitoring the load property in real time by the control method and then sending a PWM trigger pulse to operate the corresponding IGBT switch by the controller.
Claims (1)
1. A vector control and direct torque control integrated method for frequency conversion and voltage regulation of an induction motor is characterized in that: comprises the following steps:
a) Establishing vector control mathematical model and differential equation of induction motor, converting three-phase U, V and W alternating current and voltage of stator to d axis and control torque q axis of synthetic magnetic field according to the following differential equation, and calculating to obtain desired value i d(t) 、i q(t) Expected value and i obtained by real-time detection 1d(t) 、i 1q(t) The difference is the Pulse Width Modulation (PWM) adjustment control quantity which controls the switch of an Insulated Gate Bipolar Transistor (IGBT) so as to complete the linear operation state that the constant torque or the detection voltage/the motor stator frequency (V/f) is equal to the constantComprehensively controlling the magnetic field and the torque;
in the formula: i all right angle d(t) 、i q(t) Is the calculated expected value; I.C. A 1 Is the stator current; theta.theta. x Is the angular displacement of the rotor advanced by the rotating magnetic fieldω x The slip ratio of the rotating magnetic field leading rotor of the induction motor can be obtained by the following formula:
L 2 ＝l 2 +M，l 2 is the self-inductance of the rotor of the motor, M is the mutual inductance between the rotor and the stator, r 2 Is the resistance of the rotor circuit and,
in the formula: omega r ＝ω s -ω s ω x ，ω s Is the speed of the rotating magnetic field, ω s =2 π f, in constant torque operation, according to ω s Obtaining a relation proportional to the voltage; omega r Is the angular velocity of the rotor rotation;
b) Establishing direct torque control model of induction motor, and calculating torque T of induction motor in real time by the following formula m And the rotating magnetic field, \58388 s The magnitude and direction are compared with the expected torque and the expected magnetic flux to obtain a difference value which is a Pulse Width Modulation (PWM) regulation control quantity, and the control quantity controls the on-off of an Insulated Gate Bipolar Transistor (IGBT), so that the interval regulation limited by direct torque control is completed, and the reaction hysteresis quality of vector control in a non-constant torque or non-linear operation area is compensated;
P r ＝E 2N I 1 cosθ s (4)
in the formula: p r Is the power transmitted by the stator to each phase of the rotor; n is s Is the rotational speed of the rotor per minute, which can be based on ω in constant torque operation s Calculating; e 2N Is prepared from \58388 s The induced voltage; i is 1 Is the stator current; theta s Is the power factor angle of the motor and can acquire the stator voltage E and the current I in real time 1 Obtaining; voltage E 2N By measuring E at the stator end 1N Then subtract I 1 r 1 Voltage drop over the capacitor;
in the formula: \58388 s Is a resultant rotating magnetic field generated by the stator and the rotor together; omega r Is the angular velocity of rotation of the rotor, which may be determined by
Calculating; e d Is a direct current power supply before PWM inversion; k is a constant, depending on the physical structure of the motor;
c) The power factor is adjusted by controlling the Insulated Gate Bipolar Transistor (IGBT) switches to adjust the reactive power distribution.
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| CN101604953B (en) * | 2009-07-13 | 2011-09-07 | 山西合创电力科技有限公司 | Comprehensive method for vector or direct torque control of permanent magnetism type wind generator |
| CN101844725B (en) * | 2010-05-25 | 2012-07-25 | 上海明东集装箱码头有限公司 | Line voltage drop automatic detection control method for hoisting machinery |
| CN101844725A (en) * | 2010-05-25 | 2010-09-29 | 许伟新 | Line voltage drop automatic detection control method for hoisting machinery |
| CN102055401A (en) * | 2011-01-10 | 2011-05-11 | 武汉市菱电汽车电子有限责任公司 | System and method for controlling indirect torque of single regulating loop of three-phase induction motor |
| CN102315816B (en) * | 2011-08-02 | 2013-10-23 | 江苏大学 | Arbitrary level real-time sector subdivision direct torque control method |
| CN102315816A (en) * | 2011-08-02 | 2012-01-11 | 江苏大学 | Arbitrary level real-time sector subdivision direct torque control method |
| CN102434441A (en) * | 2011-11-15 | 2012-05-02 | 青岛天信电气有限公司 | Explosion-prevention constant pressure liquid supply device of emulsion pump station |
| CN102710205A (en) * | 2012-06-13 | 2012-10-03 | 深圳市汇川技术股份有限公司 | Orientation control system and method for asynchronous motor |
| CN102710205B (en) * | 2012-06-13 | 2016-05-25 | 深圳市汇川技术股份有限公司 | The directed control system of asynchronous machine and method |
| CN103840736A (en) * | 2012-11-28 | 2014-06-04 | 西门子工厂自动化工程有限公司 | Load torque control system and load torque control method |
| CN103840736B (en) * | 2012-11-28 | 2016-11-30 | 西门子工厂自动化工程有限公司 | Load torque control system and load torque control method |
| CN103973173A (en) * | 2013-01-30 | 2014-08-06 | 尼得科电机有限公司 | Sensorless motor braking system |
| CN103973173B (en) * | 2013-01-30 | 2016-09-28 | 尼得科电机有限公司 | Electric machine without sensor brakes |
| CN106680631A (en) * | 2016-12-30 | 2017-05-17 | 深圳市英威腾电气股份有限公司 | Transformer loading method, apparatus and system |
| CN113872470A (en) * | 2021-09-01 | 2021-12-31 | 河北汉光重工有限责任公司 | Dual-mode composite control method of brushless direct current motor |
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