CN100556072C - Carrier synchronization production method and device thereof in a kind of terrestrial digital television system - Google Patents

Abstract

The invention discloses a kind of carrier synchronization method and device thereof of digital television ground broadcast transmission system receiver, belong to field of digital information processing, it is characterized in that: at first after frame synchronization, receiving terminal is local to be produced and the corresponding to PN sequence of reception frame head, correlated results according to former and later two subsequences in the frame head carries out Frequency Estimation, and the frequency offset signal of gained is through carrying out frequency correction after the loop filtering.Carrier synchronization scheme of the present invention is fit to the frame structure of single carrier and two kinds of patterns of multicarrier, and possesses characteristics such as synchronizing speed is fast, noise robustness good, suitable hardware realization.

Description

Carrier synchronization production method and device thereof in a kind of terrestrial digital television system

Technical field

The invention belongs to the digital information transmission field, be specifically related to the carrier synchronization method in a kind of terrestrial digital television system.

Background technology

The development of television industries and cause is carried out round the center that rises to of rating quality and service ability always, and Digital Television is as new generation TV tech, and its rating quality increases substantially; Simultaneously, development space has been created in more other service that is adopted as of digitizing technique.The development of Digital Television will be of great importance to the development of whole electronic information industry.

Ground system of digital television broadcast is the important component part in the radio and television system.It is with digital video broadcast-satellite system and cable digital TV broadcast system and other auxiliary system is collaborative mutually provides comprehensive audient to cover, and is part and parcel in the comprehensive nerve of a covering of China's radio and television.China has 2/3rds TV user will lean on wireless coverage; In addition, under situations such as natural calamity, war, than cable TV and satellite television, terrestrial television is the transmission system with fast quick-recovery radio and television covering power.

At present, the international standard of terrestrial DTV mainly contains three: what European DVB tissue proposed is DVB-T (Digital Video Broadcasting-Terrestrial, the digital video broadcast-terrestrial) standard of core technology with COFDM; What U.S. Major Leagues tissue proposed is ATSC (AdvancedTelevision System Committee, the Advanced Television Systems Committee) standard of core technology with 8VSB; What Japan proposed is ISDB-T (Integrated Services Digital Broadcasting, the Integrated Services Digital Broadcasting) standard of core technology with BST-OFDM.ATSC is a kind of single-carrier system, and DVB-T and ISDB-T belong to multicarrier system.

Simultaneous techniques is that any one communication system all needs the practical problem that solves, and its performance is directly connected to the performance of whole communication system.We can say there is not synchronized algorithm accurately, just can not carry out reliable transfer of data, it is the prerequisite of information reliable transmission.Recover in order to carry out reliable carrier wave, U.S. ATSC system has added a little pilot signal at distance lower sideband boundary.Carrier recovering method at present more to be adopted this pilot signal elder generation automatic frequency tracking, and the phase-locked Phase Tracking of carrying out in arrowband is similar with traditional analog phase-locked look method again.Make frequency retrieval approach ideal, the cost of paying is the signal power that increases 0.3dB, as shown in Figure 1.And European DVB-T transmission system adds special structure in time domain, it is Cyclic Prefix, and at frequency domain insertion continuous pilot and scattered pilot signal, the position of continuous pilot in each COFDM symbol all fixed, in the 8k pattern, insert 177 continuous pilot, in the 2k pattern, inserted 45 continuous pilot.The position of scattered pilot is different in different COFDM symbols, but is loop cycle with four COFDM symbols, as shown in Figure 2.Europe COFDM adopts typical OFDM frequency estimating methods, in order to guarantee bigger capture range and to optimize tracking performance, is divided into slightly synchronous and thin estimation synchronously.Be unit with the subcarrier spacing generally, carrier wave frequency deviation is divided into integer part and fractional part.Carry out earlier that thick i.e. integer-times frequency synchronization is reduced to less than subcarrier spacing half with several times initial subcarrier spacing frequency deviation synchronously, carry out then carefully estimating that synchronously the thin Frequency Synchronization of promptly little several times is come the fractional part of frequency offset of correct residual.Frequency offset correction adopts a tracking loop that has frequency detector to realize usually.

But be subjected to the influence of channel multi-path when adopting Cyclic Prefix to carry out Frequency Estimation easily, also with regularly relevant, the frequency correction time is longer simultaneously, and reception reliability is relatively poor.

Summary of the invention

The objective of the invention is to propose a kind of is the frame synchronization head with the PN sequence; protect the device of this method of methods and applications of filling at interval simultaneously as the symbol of signal frame; adopt the mode of spread-spectrum correlation reception to carry out carrier synchronization at receiving terminal; thereby improved the performance and the reception reliability of terrestrial digital television system effectively, in the communication environment of complexity, obtained better net synchronization capability.

Carrier synchronization method provided by the invention may further comprise the steps:

1, the digital television signal that receives is divided into frame head and frame two parts, wherein frame head length is P symbol; Frame head is divided into preceding M symbol, a middle L symbol and three parts of M the symbol in back more arbitrarily;

2, calculate local PN sequence and reception frame head PN sequence respectively at the correlated results of M symbol of a preceding M symbol and back, obtain correlated results R respectively ₁And R ₂

3, to local PN sequence with receive the correlated results R of frame head PN sequence at M the symbol in back ₂Get conjugation, obtain R ₂ ^*

4, with the R as a result of step 3 ₂ ^*With R as a result in the step 2 ₁Multiply each other, obtain multiplied result γ;

5, the argument of getting γ obtains arg[γ], and multiply by (2M+L)/(2 π (M+L)), obtain the estimated value of frequency deviation;

6, the frequency offset estimating value of gained in the step 5 is carried out frequency correction to received signal after loop filtering.

Above-mentioned signal frame is made up of frame head and frame two parts, and frame head is the PN sequence, and frame is single-carrier modulated data symbol or multi-carrier OFDM modulated data symbol.

Three parts that signal frame PN frame head is divided have following relation: 2M+L=P.

Carrier synchronization device in a kind of terrestrial digital television system is by forming as the lower part: correlator A, correlator B, controller, get conjugation means, multiplier A, get the argument device, loop filter, digital controlled oscillator and multiplier B.Described correlator A is identical with correlator B, and described multiplier A is identical with multiplier B; Correlator A and correlator B all have two input signals, are respectively to receive PN sequence and local PN sequence; Controller has two output signals, controls correlator A and correlator B respectively; The input signal of multiplier A is result and the correlator B output of correlator A and the result who gets conjugation; Get the output signal that is input as multiplier A of argument device, output signal is the estimated value of frequency deviation; Loop filter is a low pass filter, and its input signal is the output of getting argument, loop filter output control digital controlled oscillator; The input signal of digital controlled oscillator is the output signal of loop filter, is output as frequency correction signal; Multiplier B has two input signals, is respectively the frame data symbol that receives and the output of digital controlled oscillator, and output signal is for through the frame data symbol after the frequency correction.

The signal frame frame head PN sequence that receives is carried out related operation respectively at a preceding M symbol and M symbol in back and local PN sequence in correlator A and correlator B, obtain correlated results R respectively ₁And R ₂, the relevant initial moment and the finish time of correlator A and correlator B are controlled by controller; The output R of correlator B ₂Carrying out obtaining R behind the conjugate operation through getting conjugation means ₂ ^*R afterwards ₁And R ₂ ^*In multiplier A, multiply each other, obtain multiplied result signal γ; Signal γ is through getting the argument arg[γ that the argument device is tried to achieve signal γ], and argument multiply by the estimated value that obtains frequency deviation behind (2M+L)/(2 π (M+L)); The estimated value of this frequency deviation carries out after the low-pass filtering digital controlled oscillator being controlled through loop filter, the output of digital controlled oscillator is the frequency offset correction signal, at last signal frame frame data symbol and the frequency offset correction signal that receives multiplied each other in multiplier B, promptly obtain through the frame data symbol after the frequency correction.

Described controller is made up of counter, comparator A and comparator B; Counter begins counting in the initial moment of the frame head of each signal frame by 0, when the count results of counter is in 0 to M scope, comparator A output high level signal is given correlator A enable signal as correlator A, on the contrary comparator A output low level signal; When the count results of counter at (M+L) to the scope of (2M+L) time, comparator B output high level signal is given correlator B enable signal as correlator B, on the contrary comparator B output low level signal.

The invention has the advantages that:

(1) adopt the PN sequence to carry out estimating carrier frequencies, owing to adopted the mode of correlation reception at receiving terminal, thus can access higher signal to noise ratio after relevant, thus the precision of carrier synchronization improved;

(2) adopt the PN sequence to carry out estimating carrier frequencies, because the PN relevant peaks, the structure with respect to Cyclic Prefix under multipath channel can obtain better receptivity;

(3) carrier synchronization method of the present invention adapts to two kinds of modulation systems of single carrier and multicarrier and different frame structures, and it is simple to possess realization, the characteristics that net synchronization capability is stable.

Description of drawings

Fig. 1 is the position of ATSC system pilot in frequency spectrum;

Fig. 2 is a DVB-T system pilot tone schematic diagram;

Fig. 3 is the multi-frame structure figure of terrestrial DTV standard;

Fig. 4 is the frame assumption diagram of terrestrial DTV standard;

Fig. 5 is a PN frame head structure chart;

Fig. 6 is the signal processing flow figure of carrier synchronization of the present invention;

Fig. 7 is a carrier synchronization structure drawing of device of the present invention;

Fig. 8 is the structure chart of carrier synchronization device middle controller of the present invention;

Fig. 9 is the average of frequency offset estimating of the present invention;

Figure 10 is the variance of frequency offset estimating of the present invention.

Among the figure: 1. correlator A 2. correlator B 3. controllers 301. counters

302. comparator A 303. comparator B 4. get conjugation means 5. multiplier B 6. and get the argument device

7. loop filter 8. digital controlled oscillators 9. multiplier B

Embodiment

The present invention is described in further detail below in conjunction with drawings and Examples.

Figure 3 shows that the multi-frame structure figure of the multilevel hierarchy in the terrestrial DTV standard.Wherein signal frame is the most basic construction unit, and signal frame is made up of frame head and frame two parts, and wherein frame head has three kinds of patterns, i.e. PN420, PN595 and PN945, respective signal frame length 555.56 μ s, 578.7 μ s and 625 μ s respectively.

Superframe is the set of one group of signal frame, and the time span of superframe is 125ms under the different frame head mode, so under three kinds of patterns of PN420, PN595 and PN945, a superframe contains 225,216 and 200 signal frames respectively.

Dividing frame definition is the set of one group of superframe, and time span is 1 minute, so a branch frame contains 480 superframes.

Day, frame definition was the set of a component frame, and time span is 24 hours, contains 1440 branch frames.

Fig. 4 has shown the composition diagram of three kinds of signal frame structure of terrestrial digital television system more specifically.

It is in order to adapt to different applied environments and coverage that terrestrial digital television system provides the purpose of three kinds of different frame head modes.PN420 can provide the protection interval of 55.6 μ s; be adapted at setting up in the city scope regional single frequency network; PN945 can provide the protection interval of 125 μ s; be fit to set up national single frequency network on a large scale; and PN595 can provide the protection interval of 78.7 μ s; simultaneously because PN595 simple in structure, and do not have the recycling-guard characteristic, can utilize time-domain equalizer to resist the characteristic of channel at receiving terminal.

Figure 5 shows that the generation type of the PN frame head of a kind of cyclic extensions in the native system.And PN420 and PN945 have adopted the PN frame head of cyclic extensions in state's mark system.The elementary cell of the PN frame head of cyclic extensions is complete m sequence, and back C bit of m sequence is put into the front of m sequence, preceding D bit of m sequence is put into the back of m sequence, thereby has formed the form of cyclic extensions.It is the form of the truncated sequence of 1023 m sequence that PN595 has then adopted length.

Figure 6 shows that the concrete steps and the flow process of carrier synchronization method provided by the invention.

Suppose that the signal that receiver inlet receives is:

$r_0\left(t\right)=s_0\left(t\right)e^{j(2\mathrm{\π\Δ}{f}_{c}t+\mathrm{\θ})}+n_0\left(t\right)=s_0\left(t\right)e^{j(2\mathrm{\π\Δ}{f}_c\·T_{\mathrm{PN}}t/T_{\mathrm{PN}}+\mathrm{\θ})}+n_0\left(t\right)=s_0\left(t\right)e^{j(2\mathrm{\π}{\mathrm{\ϵ}}_{c}t/T_{\mathrm{PN}}+\mathrm{\θ})}+n_0\left(t\right),$

N wherein ₀(t) be that bilateral power spectral density is N ₀Additivity Gauss's self noise (AWGN) of/2, s ₀(t) be the targeting signal (being the PN sequence length) of transmission.θ is a carrier phase, Δ f _cBe carrier frequency offset to be estimated, ε _c=Δ f _cT _PNBe normalized carrier wave frequency deviation, T _PNBe the shared time width of frame head PN.

At receiving terminal to r ₀(t) with f _sSample and obtain received signal sample sequence r[k], have:

$r\left[k\right]={\mathrm{\σ}}_{s}c\left[k\right]e^{j(2\mathrm{\π}{\mathrm{\ϵ}}_{c}k/P+\mathrm{\θ})}+n\left[k\right],$

Wherein: σ _sThe expression transmit signal power; P represents PN symbolic number that frame head comprises; C[k] symbol after the mapping of expression PN sequence, c[k]=δ (1+j), δ=± 1, by each bit decision of PN frame head, if this bit value is 0, δ=+ 1 then, if bit value is 1, δ=-1 then.

Before the carrier synchronization, receiving terminal at first partly carries out related operation to the PN frame synchronizing signal that receives, and correlation length is a PN sequence length.By adjusting the phase place of local PN sequence and reception PN sequence, make local sequence and receiving sequence phase alignment.

The digital television signal that receives is divided into frame head and frame two parts, and wherein frame head length is P symbol; Frame head is divided into preceding M symbol, a middle L symbol and three parts of M the symbol in back more arbitrarily, following relational expression: 2M+L=P is wherein arranged.

Calculate local PN sequence respectively and receive the correlated results of frame head PN sequence, obtain R at M symbol of a preceding M symbol and back ₁And R ₂:

${\mathrm{<figure-callout\; id="1"\; label="R"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">R</figure-callout>}}_1=\left(\underset{k=0}{\overset{M-1}{\mathrm{\&Sigma;}}}{c}^{*}\right[k\left]r\right[k\left]\right),{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">R</figure-callout>}}_2=\left(\underset{k=0}{\overset{M-1}{\mathrm{\&Sigma;}}}{c}^{*}\right[k+(L+M)]\&CenterDot;r[k+(L+M)\left]\right),$

Again to local PN sequence and the correlated results R of reception frame head PN sequence at M the symbol in back ₂Get conjugation, obtain R ₂ ^*Afterwards with R ₂ ^*With R ₁Multiply each other, obtain γ:

$\mathrm{\&gamma;}=\left(\underset{k=0}{\overset{M-1}{\mathrm{\&Sigma;}}}{c}^{*}\right[k\left]r\right[k\left]\right)\&CenterDot;{\left(\underset{k=0}{\overset{M-1}{\mathrm{\&Sigma;}}}{c}^{*}\right[k+(L+M)]\&CenterDot;r[k+(L+M)\left]\right)}^{*}={{\mathrm{\&sigma;}}_s}^2\frac{\sin (\mathrm{\&pi;}{\mathrm{\&epsiv;}}_{c}M/P)}{\sin (\mathrm{\&pi;}{\mathrm{\&epsiv;}}_c/P)}{\mathrm{<figure-callout\; id="2"\; label="e\; j"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">e</figure-callout>}}^j,$

Wherein, L is two mark spaces between the subsequence, and relational expression P=2M+L is arranged.

The argument of getting γ obtains arg[γ], and multiply by (2M+L)/(2 π (M+L)), obtain the estimated value of frequency deviation:

ε _c＝arg[γ]P/(2π(M+L))＝arg[γ](2M+L)/(2π(M+L))

The sampled signal that receives at first multiplies each other with the multiple sinusoidal signal of digital controlled oscillator output, promptly input signal is carried out the phase place rotation, and the gained signal obtains the estimation estimated value ε of carrier wave frequency deviation after frequency offset estimating _c, ε _cCarry out the controlled quentity controlled variable that low-pass filtering obtains digital controlled oscillator through loop filter, this controlled quentity controlled variable is controlled the frequency of the multiple sinusoidal signal of digital controlled oscillator output again, thereby finishes the closed loop adjustment process of carrier frequency offset.Adopt the desirable second order loop in the analog PLL loop filter to realize at native system intermediate ring road filter.

The frequency offset estimating variance is:

$\mathrm{var}\left({\widehat{\mathrm{\&epsiv;}}}_c\right)=\frac{{\mathrm{<figure-callout\; id="2"\; label="P"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">P</figure-callout>}}^2}{4{\mathrm{\&pi;}}^2{M}^3}\{\frac{{\mathrm{\&sigma;}}_{\mathrm{<figure-callout\; id="2"\; label="n"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">n</figure-callout>}}^2}{{\mathrm{\&sigma;}}_{\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">s</figure-callout>}}^2\sin c^2({\mathrm{\&epsiv;}}_{c}M/P)}+\frac{1}{2M}\&CenterDot;{\left[\frac{{\mathrm{\&sigma;}}_{\mathrm{<figure-callout\; id="2"\; label="n"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">n</figure-callout>}}^2}{{\mathrm{\&sigma;}}_{\mathrm{<figure-callout\; id="2"\; label="s"\; filenames="C20071011979600151.png,C20071011979600161.png"\; state="\{\{state\}\}">s</figure-callout>}}^2\sin c^2({\mathrm{\&epsiv;}}_{c}M/P)}\right]}^2\}.$

By the variance of frequency offset estimating as can be seen, the variance of estimation will reduce with the raising of signal to noise ratio, and the length of variance and M is inversely proportional to, and the variance of the big more estimation of M is more little.This point can obtain very easily explaining from spreading gain.Because the outstanding autocorrelation performance of PN sequence, when M was big more, the signal to noise ratio of relevant peak value afterwards was with respect to the improvement maximum of input signal-to-noise ratio, so M is long more, and relevant signal to noise ratio afterwards is high more, and estimated result is also stable more.

Because, so the scope of frequency offset estimating is:

$\left|\mathrm{\&Delta;}{f}_c\right|<\frac{2M+L}{2(M+L)T_{\mathrm{PN}420}}=(1-\frac{L}{2(M+L)})\&CenterDot;\frac{1}{T_{\mathrm{PN}420}},$

Provable above-mentioned Nonlinear Transformation in Frequency Offset Estimation is not estimated for having partially.

Above-mentioned signal frame is made up of frame head and frame two parts, and frame head is the PN sequence, and frame is single-carrier modulated data symbol or multi-carrier OFDM modulated data symbol.

Figure 7 shows that the structure chart of carrier synchronization device of the present invention, by correlator A 1, correlator B 2, controller 3, get conjugation means 4, multiplier A 5, get argument device 6, loop filter 7, digital controlled oscillator 8 and multiplier B9 form.

PN sequence that receives and local PN sequence are carried out related operation respectively in correlator A1 and correlator B2, obtain R respectively ₁And R ₂, the relevant initial moment and the finish time of correlator A 1 and correlator B 2 are controlled by controller 3; The output R of correlator B 2 ₂Through getting conjugation means 4 after carrying out conjugate operation to R ₂ ^*R afterwards ₁And R ₂ ^*In multiplier A 5, multiply each other, thereby obtain γ; Signal γ is through getting the argument arg[γ that argument device 6 is tried to achieve γ], and multiply by the estimated value that obtains frequency deviation behind (2M+L)/(2 π (M+L)); This frequency offset estimating value carries out controlling digital controlled oscillator 8 after the low-pass filtering through loop filter 7, the output of digital controlled oscillator 8 is the frequency offset correction signal, frame data symbol and frequency offset correction signal with the signal frame that receives multiplies each other in multiplier B9 at last, promptly obtains through the frame data symbol after the frequency correction.

Figure 8 shows that the structure chart of carrier synchronization device middle controller 3 of the present invention.Controller 3 is made up of counter 301, comparator A 302 and comparator B 303.Counter 301 begins counting in the initial moment of the frame head of each signal frame by 0, count down to the end of this signal frame.The count results of 302 pairs of counters 301 of comparator A is judged, when the count results of counter 301 is in 0 to M scope, comparator A 302 output high level signals are given correlator A 1 enable signal as correlator A 1, comparator A 302 output low level signals in the time of not in 0～M scope.1 of correlator A carries out correlation computations during for high level at enable signal.Comparator B 303 judges the count results of counter 301 equally, when the count results of counter 301 at (M+L) to the scope of (2M+L) time, comparator B 303 output high level signals are given correlator B 2 enable signal as correlator B 2, comparator B 303 output low level signals in the time of not within this scope.2 of same correlator B carry out correlation computations during for high level at enable signal.

Figure 9 shows that method of the present invention in frame head PN420 pattern, signal to noise ratio is at 15dB, 20dB and 25dB, under the 64QAM constellation mapping pattern, and the average of the Nonlinear Transformation in Frequency Offset Estimation during parameter M=210.Under this parameter, the result of frequency offset estimation range Theoretical Calculation is ± 18kHz.Transverse axis is the actual size of the frequency deviation of received signal among Fig. 8, and the longitudinal axis is for adopting the resulting Nonlinear Transformation in Frequency Offset Estimation value of carrier frequency bias estimation provided by the invention.The result of the scope of frequency offset estimating and Theoretical Calculation is consistent as seen from Figure 9.And has good linear relationship between frequency offset estimating value and the actual value.Because during M=210, relevant after the signal to noise ratio of peak value with respect to input signal-to-noise ratio the spreading gain of 23dB is arranged, so noise is very little to result's influence of frequency offset estimating, the estimation average under the different noises all point-blank.

Figure 10 shows that the normalization variance of the Frequency Estimation of method of the present invention under the parameter identical with Fig. 9, wherein transverse axis is the actual value of frequency deviation, and the longitudinal axis is the normalization variance of frequency offset estimating.As can be seen from Figure 10, the variance of frequency offset estimating under the given signal to noise ratio condition all 10 ^-5Below the magnitude, satisfy the needs of real system, and along with the raising of signal to noise ratio, the variance of frequency offset estimating reduces thereupon also, is consistent with the trend of Theoretical Calculation.

Claims (4)

1, the carrier synchronization method in a kind of terrestrial digital television system is characterized in that, may further comprise the steps:

Step 1: the digital television signal that receives is divided into frame head and frame two parts, and wherein frame head length is P symbol; Frame head is divided into preceding M symbol, a middle L symbol and three parts of M the symbol in back more arbitrarily, and P, M, L satisfy following relation: 2M+L=P;

Step 2: calculate local PN sequence respectively and receive the correlated results of frame head PN sequence, obtain R respectively at M symbol of a preceding M symbol and back ₁And R ₂

Step 3: to local PN sequence and the correlated results R of reception frame head PN sequence at M the symbol in back ₂Get conjugation, obtain R ₂ ^*

Step 4: with the R as a result of step 3 ₂ ^*With R as a result in the step 2 ₁Multiply each other, obtain γ;

Step 5: the argument of getting γ obtains arg[γ], and multiply by (2M+L)/(2 π (M+L)), obtain the estimated value of frequency deviation;

Step 6: the frequency offset estimating value of step 5 gained is carried out frequency correction to received signal after loop filtering.

2, the carrier synchronization method in a kind of terrestrial digital television system according to claim 1, it is characterized in that: described signal frame is made up of frame head and frame two parts, frame head is the PN sequence, and frame is single-carrier modulated data symbol or multi-carrier OFDM modulated data symbol.

3, a kind of carrier synchronization device that is applied in the described a kind of terrestrial digital television system of claim 1, it is characterized in that, by correlator A, correlator B, controller, get conjugation means, multiplier A, get the argument device, loop filter, digital controlled oscillator and multiplier B form;

Described correlator A is identical with correlator B, and described multiplier A is identical with multiplier B; Controller has two output signals, controls correlator A and correlator B respectively; Correlator A carries out related operation to signal frame frame head PN sequence at a preceding M symbol and local PN sequence, and correlator B carries out related operation to signal frame frame head PN sequence at M symbol in back and local PN sequence; The result of correlator A and correlator B export and get the input signal of the result of conjugation as multiplier A by getting conjugation means; Get the argument device after multiplier A output signal is got argument, this argument is on duty to obtain the estimated value of frequency deviation behind coefficient (2M+L)/(2 π (M+L)), wherein frame head length is P symbol, frame head is divided into preceding M symbol, M symbol of a middle L symbol and back arbitrarily, and satisfies relational expression: 2M+L=P; Loop filter is a low pass filter, and its input signal is the output of getting argument, loop filter output control digital controlled oscillator; The input signal of digital controlled oscillator is the output signal of loop filter, is output as frequency correction signal; Multiplier B has two input signals, is respectively the frame data symbol that receives and the output of digital controlled oscillator, and output signal is for through the frame data symbol after the frequency correction.

4, the carrier synchronization device in a kind of terrestrial digital television system according to claim 3, it is characterized in that: described controller is made up of counter, comparator A and comparator B; Counter begins counting in the initial moment of the frame head of each signal frame by 0, when the count results of counter is in 0 to M scope, comparator A output high level signal is given correlator A enable signal as correlator A, on the contrary comparator A output low level signal; When the count results of counter at (M+L) to the scope of (2M+L) time, comparator B output high level signal is given correlator B enable signal as correlator B, on the contrary comparator B output low level signal.

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