CH695516A5 - Optoelectronic distance measurement in construction and plant engineering industry, involves modulating optical intensities of main and reference emitters, and determining phase of resultant signals simultaneously - Google Patents

Abstract

Optical intensities of main emitter (1) and reference emitter (2) are modulated simultaneously at different frequencies (f1,f2). The resultant frequencies are converted to intermediate frequency (IF) components (fzF1,fzF2). The separation of the phase information between the IF component is determined based on modulated difference frequencies (f1-f2) and IF difference frequencies (f(zF1)-f(zF2)). An independent claim is also included for optoelectronic distance measurement device.

Description

CH 695 516 A5

description

The invention relates to a method for optoelectronic distance measurement according to the preamble of patent claim 1 and a device based on the method according to the preamble of claim 12.

The optoelectronic measurement of distances up to 100 m with accuracies of a few millimeters has become important for numerous applications, especially in the construction industry and in plant construction. The dynamics of such distance measuring systems should be as high as possible in order to be able to process both very weak and strong signals. This eliminates the need to use defined targets on the object whose distance is to be determined from an observation location. The possibility of direct distance measurement on certain surfaces, i. without the use of target brands, it enables reduced production times and cost savings, in particular in the sectors mentioned, while at the same time reducing manufacturing tolerances.

Methods and apparatus for accurate optoelectronic distance measurement are known. In most cases, as in the case of the invention, a preferably sinusoidally intensity-modulated beam of a light source, in particular a laser diode, is directed onto a measurement object. The intensity-modulated light backscattered by the measurement object is detected by a photodiode. The distance to be measured results from the phase shift of the sinusoidally modulated light intensity backscattered by the measurement object in relation to the emitted light intensity of the light source.

A major problem with highly accurate distance or phase measurement systems of the type mentioned is the elimination of temperature and age-dependent parasitic phase changes of the light source, ie in particular in the laser diode transmitter and / or in the photodiode receiver. To address this problem, various methods are known.

A possibility described in the publication EP 0 701 702 B1 is the use of a mechanically switchable reference path. In this case, an intensity-modulated laser beam is first directed at the measurement object during a first measurement and, via a tiltable mirror, at a second reference distance measurement directly onto the photoreceiver. By subtracting the measured phases, temperature and aging effects of the components should be eliminated. However, since the measurement of the distance and the reference distance varies widely with different optical reception powers, a resulting measurement error is not eliminated. A major disadvantage of this concept is also the use of movable mechanical components, which limits the reliability and life of the entire measuring system.

Other known distance measuring devices described in DE 19 643 287 A1 of the type in question here operate with a reference photoreceiver and a main photoreceiver. In this case, part of the intensity-modulated laser light is directed onto the test object and from there to the main photoreceiver and a separate part separated from the laser light beam directly onto the reference photoreceiver. Since the reference photoreceiver is constantly illuminated during a measurement, no moving mechanical switch is required. In this concept, although the phase response of the laser diode transmitter is eliminated, but not the time-varying phase behavior of the receiving components, which is generally different for the measuring and Referenzmesszweig. In addition, even with distance measuring devices of this type with very different reception powers in both branches to be expected, resulting in further phase errors.

In another type of known opto-electronic distance measuring device (see US-4,403,857), which forms the starting point for the invention to be described here, two laser emitters and two photodiode receivers are used to eliminate said phase errors. As stated in detail in the preambles of claims 1 and 12, in this device, a part of the intensity modulated power of a main light emitter is directed directly to the measurement object, from where it comes as a scattered light on a main photoreceiver. Another part of this transmission power is routed via a well-known first reference distance to a reference photoreceiver. Furthermore, a reference light emitter is present whose output power is also intensity-modulated and from a part passes through a second reference distance to the main photoreceiver, while another part is passed over a third reference distance directly to the reference photoreceiver.

The main and the reference light emitter are activated in chronological succession via an electronic switch. This measurement principle does not require mechanical switches. In addition, temperature and aging-related phase changes are completely eliminated both in the transmitting and in the receiving unit. Since, however, the measurements with the signals of the main and reference light emitter are to be expected to result in considerable differences in received power, the resulting phase errors are not eliminated even in the concept underlying this known distance measuring device. Receive power-dependent phase errors are particularly noticeable in avalanche photodiodes (APD), which are preferred because of other advantages as the main receiver. At high gains, saturation effects gradually increase with increasing power, which makes the avalanche gain dependent on the received power. Thus arises in addition a power-dependent phase rotation when receiving high-frequency modulated optical radiation. In addition, with the received power, the generated charge in the junction of the APD varies, thereby affecting the junction width and hence the junction capacitance. With the junction capacitance, the phase behavior of the low-pass filter formed by it changes. With high APD amplification factors, a phase rotation of usually greater than 5 ° can thereby be produced with a reception power variation of two orders of magnitude.

The invention is therefore based on the object of providing a distance measuring method and a method according to this method.

2

CH 695 516 A5

specify ren operating device with which a highly accurate distance measurement urse and which is completely independent of temperature, aging and receiving power-dependent phase errors. Mechanical or electronic switches should be dispensed with and the overall measuring time for obtaining reliable measurement results should be significantly shortened.

The invention is in a method for optoelectronic distance measurement according to the preamble of claim 1, characterized in that the light intensities of the main and the reference emitter are modulated simultaneously with different frequencies, wherein the signal supplied by the main and by the reference receiver signal mixtures, respectively each containing a signal component with the intensity modulation frequency of the main emitter and a signal component with the intensity modulation frequency of the reference emitter, each converted into an intermediate frequency range containing two frequency components, wherein the one frequency component is formed with the signal of the reference emitter and the other frequency component with the signal of the main emitter , And that for the comparative signal evaluation, the separation of the phase information contained in the two simultaneously incurred intermediate frequency signals due to the different frequencies in Zwischenfreque Nzbereich and the different modulation frequency for the intensity modulation of the main and reference beam takes place.

Advantageous developments of this inventive distance measuring method are defined in dependent dependent claims.

An inventive apparatus for optoelectronic distance measurement with the features of the preamble of claim 12 is inventively characterized by a device by which the light emitted from the main or by the reference emitter light beams are intensity modulated simultaneously with different frequencies.

Advantageous embodiments of this inventive distance measuring device are also defined in further dependent claims.

Similar to the distance measuring method described in US Pat. No. 4,403,857, two light emitters, in particular lasers and two photodiode receivers, are also used in the subject matter of the invention. In contrast to this known method, however, according to the invention, the first light source, which is preferably sinusoidally intensity-modulated with a first modulation frequency f1, is directed onto the surface of a measurement object of the first light transmitter designated as the main emitter. The backscattered from there, also intensity-modulated light passes, for example via a receiving optics on the designated as the main receiver second photoreceptor. At the same time, part of the modulated light of the main emitter is led directly over a first reference path onto the second photoreceiver designated as the reference receiver. The reference emitter is also preferably sinusoidally intensity modulated with a second modulation frequency. A portion of its modulated optical radiation reaches the main receiver via a second known reference path and in particular via a scattering medium, while another portion of its modulated optical radiation reaches the reference receiver via a third reference path.

Both receivers are acted upon by both emitter signals simultaneously, so that in contrast to the distance measuring method described in said US Patent no switch is needed and the measurement time is significantly shortened. The photoreceivers convert the detected modulated optical powers into photocurrents, which are then converted into voltages, preferably with transimpedance amplifiers.

The two signal voltages thus obtained are then converted by using a locally generated frequency by associated mixer in suitable intermediate frequency ranges and then evaluated by analog-to-digital conversion of a signal evaluation for error-free determination of signal delay time-dependent phase shift and thus the distance.

The invention and advantageous details are explained below with reference to the drawings.

Fig. 1 shows a principal, preferred construction of a distance measuring device based on the method according to the invention, and

Fig. 2a, b illustrates a obtained at the output of the amplifier 16 in Fig. 1 intermediate frequency signal mixture in the time domain (Fig. 2a) and in the frequency domain (Fig. 2b).

The principle arrangement of a method according to the invention operating distance measuring device includes a main emitter 1 designated first light emitter, in particular laser and a reference emitter 2 designated second light emitter, preferably also a laser, and designated as a reference receiver 4 first photoreceiver or one as a main receiver 5 designated second photoreceptor. The main emitter 1 can be a comparatively high-performance edge-emitting laser diode (EEL) whose emission wavelength, as indicated in the drawing, is for example A, = 650 nm. As the reference emitter 2, a laser diode preferably radiating at a different wavelength, for example = 850 nm, is used e.g. a VCSEL (Vertical Cavity Emitting Laser Diode) is used. The choice of different wavelengths for main and reference emitter allows a further explained below optical filtering, so that can be reduced by optical crosstalk possible problems. The reference receiver 4 is preferably a PIN photodiode, while the main receiver 5 is preferably an avalanche photodiode.

For the invention of crucial importance is the idea that emitted by the main emitter 1 and the reference emitter 2 radiant power at a given measurement time with different frequencies,

3

CH 695 516 A5

preferably sinusoidal, to modulate in intensity. Thus, for example, the main emitter 1 (initially) with the frequency f and the reference emitter 2 (first) with the frequency f intensity modulated. The two modulation frequencies f f are obtained via two frequency synthesizers 21 and 22 driven by a common oscillator 20. The feeding of the modulation signals with the frequencies f1 or f into the excitation circuit (not shown) for the main emitter 1 or the reference emitter 2 takes place, for example, via high-frequency amplifiers 23 and 24.

The output from the main emitter 1, bundled main light beam 30 passes, for example, guided in a tube 40, first on a beam splitter 3, the main light beam 30 in two parts, namely a main portion 32 which is passed to the measurement object and in a branched Divides portion 33, which passes over a first known reference distance D ^ with the interposition of a diffuser 51 or scatter on the reference receiver 4. By means of the diffuser 51 or the scatter, a homogeneous modulation phase distribution in the beam cross-section in front of the reference receiver 4 is ensured. As a beam splitter 3, a semipermeable mirror, a prism, a glass plate, an integrated-optical beam splitter, a diffractive element, for. As a hologram or the like. Be provided. It should be emphasized, however, that the beams of the reference lines do not necessarily have to be collimated or deflected by beam splitters. Alternatively, for example, a volume dispersion or direct illumination of the receiver z. B. be provided via a diffuser. The backscattered from the remote measurement object portion 36 of the measuring beam 32 passes through a collection optics 37 on the main receiver 5. The output from the reference emitter 2 usually collimated reference light beam 31 is also divided by means of a beam splitter 6 into two parts, wherein a first portion 34 via a second Reference distance D and via a diffuser 51 or Scatter on the reference receiver 4 passes, while a second portion 35 over a third reference distance D3, preferably via a matched to the wavelength of the reference emitter optical filter 41, first a scattering medium (scatter) 11 and then applied as a scattered portion together with the backscattered by the measurement object portion 36 of the main light beam to the main receiver 5. As Scatter 11 can be used in principle any scattering material. Even the housing wall would be suitable. However, to control the stray light output of the reference emitter light, the scattering wheel of the scatter material should be tuned to the receiving system. Since a strong power attenuation of the reference emitter beam 35 is desired due to the generally weak measurement signals, a scattering material with a low spreading degree is advantageous, at least for the main receiver 5, e.g. black paper, black velvet or the like

The very advantageous use of the scattering 11 in connection with the invention is based on the following observation: In the beam cross section of laser diodes, the modulation phase is not homogeneous, d. that is, different points of the beam cross section have different phases with respect to the modulated light intensity. Homogeneous backscattering measures a phase averaged over the beam cross section. In the case, however, that certain areas of the beam are faded out of the target surface, e.g. For example, if one part of the light spot strikes black, absorbing areas and another part on white, strongly backscattering areas of the measuring object surface, then the middle phase changes and a measurement error arises, which depends on the unknown object surface. This error can not be compensated. In most cases, however, all points of the spot are scattered back almost equally.

In a guided directly to the main receiver 5 reference emitter beam 35, only a small portion of the beam cross-section would be detected due to the small APD area. The thus measured phase of the beam then usually does not represent the mean modulation phase. In addition, the phase distribution in the beam cross section is not constant over time and also depends on the temperature. The use of the scatter 11 ensures that signal components are detected from all points of the beam cross section. Thus, a middle phase is measured, which has a more constant behavior than a punctiform phase. In this case, errors caused by punctual measurement of the phase of the reference emitter beam 35 are reduced by the scatter 11.

To avoid such phase errors, it may also be advantageous to irradiate the beams 33 and 34, respectively, via a scatter, e.g. the diffuser 51, before the modulated light is detected by the reference receiver 4.

Another object of the scatter 11 is a strong power attenuation of the reference emitter light. Due to the very weak measuring signals, the receiving system is very sensitive. The power attenuation prevents the APD in the main receiver 5 from being overdriven. In addition, this reduces the increased shot noise with the detected light output.

It is important to tune the scattering power of the reference emitter light to the system, so that an optimized signal-to-noise ratio (SNR) is achieved. If the scattered light power is too strong, large shot noise and therefore weak measurement signals can be expected to cause a poor SNR when determining the phase of the measurement signal. On the other hand, if the stray light powers are too low, the SNR will be poor in determining the phase of the reference emitter signal. Since this phase is also included in the distance measurement, this can lead to a measurement error. It is therefore appropriate to find a suitable compromise. For a distance measuring range of up to 100 m, it has been found in experiments that the detected scattered light power of the reference emitter beam 35 should be approximately as large as a power scattered back from a distance of 15 m of the measuring beam 36. In this case, a backscatter coefficient of the measuring object surface of 0.5 and Homogeneous backscatter assumed. For a power of the measuring beam 32 of 1 mW and a diameter of the receiving optics 37 of 50 mm, this corresponds to a detected scattered light power of the reference emitter beam of 0.7 nW.

The photocurrents supplied by the reference receiver 4 and the main receiver 5 are first converted via transimpedance amplifiers 9 and 10 into corresponding measuring voltages Xg and X1, respectively.

In principle, instead of the transimpedance amplifiers also impedance-controlled (50 ohms) HF

4

CH 695 516 A5

Power amplifiers are used. They usually have worse noise properties and lower reinforcements. However, instead of the transimpedance amplifier 9, it may be useful for the reference branch to use an impedance-controlled HF power amplifier for cost reasons, whereby a strong signal can be achieved with a lower current consumption.

The signal originating from the reference receiver 4 xg then acts on a measuring signal input of a first mixer 7, while the signal xi reaches the measuring signal input of a second mixer 8. The two mixers 7 and 8 are controlled by the same local frequency oscillator 20 (mother oscillator) via frequency synthesizer 12 with a frequency f, which is selected so that also in the output side resulting intermediate frequency range a signal mixture with the frequency components f and f is present. Here, it is important that the modulation frequencies of the main emitter 1 and the reference emitter 2 are different and the main receiver 5 and the reference receiver 4 both provide a signal mixture consisting of two signals with the frequencies fi and f. This signal mixture is converted in the two signal branches with the usually identically constructed mixers 7 and 8 and the local oscillator signal of the frequency f in the aforementioned intermediate frequency range. A direct mixing via avalanche photodiodes (APD) is conceivable in this context, wherein the APD operating high voltage is superimposed directly on the signal of the local oscillator 12. Due to the resulting modulation of the avalanche amplification, this local oscillator signal is mixed with the received signal so that the APD output current contains the two IF signal components f and fZF2. On high frequency amplifier and high frequency mixer can thus be dispensed with. However, avalanche photodiodes should be used for both the reference and the measuring branch in this case. The first intermediate frequency signal with the frequency f is then derived from the received signal at the first modulation frequency f and the intermediate frequency signal at the second intermediate frequency f from the received signal at the second modulation frequency f. After a low-pass filtering 13 or 14 for eliminating the signal components with the sum frequencies f + f and f + f and for noise signal reduction and amplification 15 or 16, the suitably amplified IF signals x4 and x3 with analog-to-digital converters 17 or 18 sampled. The first and the second intermediate frequency f or fZF2 and the sampling times of the analog-to-digital converters 17, 18, ie the measurement window, are expediently selected such that an integer number of periods of the two intermediate frequency signals f-f lie in the measurement window of the digital scan , This avoids a so-called leakage effect, which occurs in the case of the digital, cfisKreten Fourier transformation (DFT) if the frequency components are not in the frequency grid of the DFT; e.g. 40 kHz and 60 kHz for a 1 kHz frequency raster, d. h., the distance of the discrete frequency values is 1 kHz and the associated measurement window 1/1 kHz = 1 ms.

A digital Fourier transform 19 of the sampled composite signal x3 of the main receiver branch 42 and independently of the sampled composite signal x of the reference receiver branch 43, e.g. for a given measurement window as indicated above, the phases of the light signal components 33 and 36 of the main emitter 1, the phases of the light signal components 34 and 35 of the reference emitter 2 and temperature, aging and receiving power-dependent phase shifts, which are caused in the main and reference receiver branch. Thus, four phase relationships are determined.

Since the phase can be clearly measured only in an interval from 0 to 27t, but the measuring distance produces in most cases significantly larger phase shifts, according to a significant improvement of the basic idea of the invention - to achieve a clear distance measurement and the To improve measurement results - in a second measurement cycle, the modulation frequencies changed, ie, the main emitter 1 is now modulated with the frequency f and the reference emitter 2 with the frequency f sinusoidal intensity. The measurement procedure described above is repeated with these new settings, so that for these new settings there are four more phases for the sampled signal mixes in the main and reference receiver branches.

Since the signal of the main emitter 1 and the signal of the reference emitter 2 in the branch of the main receiver 5 and in the branch of the reference receiver 4 - as shown - the same components through, is formed by forming phase differences with respect to the main receiving branch 42 and in the reference receiving branch 43rd measured signal phases completely eliminates the phase behavior of the respective receiver components. In addition, the reception ratios are constant, since the main and reference emitter signals pass through the respective reception branches at the same time. The separation of the signals is - as also shown - on the different modulation frequencies f and f receive power-dependent phase errors are thus also eliminated.

By further formation of phase differences in addition, the phase behavior of the main emitter 1 and the phase behavior of the reference emitter 2 is eliminated, so that ultimately only a constant phase difference remains, which by the specified device-internal path differences of the reference distances D, D and D3 and by the ( in Fig. 1 unspecified) measuring distance Dq of both emitter signals outside the two laser emitter is formed. The measuring distance Dq thereby influences the phase of the modulated and detected by the main receiver 5 light portion 32 of the main emitter 1, the reference distance D1, the phase of the modulated and detected by the reference receiver light portion 33 of the main emitter 1, the reference distance Dg the phase of the modulated and detected by the reference receiver light portion 34th of the reference emitter 2 and the reference distance D3, the phase of the modulated and detected by the reference receiver light portion 35 of the reference emitter 2. Since the device internal maturities on the reference distances D, D and D outside the emitter 1, 2 are known and constant, the Di can be measured -

12 3

determine punching. It is therefore determined with the inventive method completely independent of the phase behavior of the transmitting and receiving units.

The phase difference determination by Fourier transformation will be explained below. The principal form of the IF signals sampled by the analog-to-digital converters 17 and 18 can be seen in FIG. 2a. In Dia5

CH 695 516 A5

gram acc. Fig. 2b is also the signal mixture shown in the frequency domain.

First, the main emitter 1 emits radiation at the modulation frequency f and the reference emitter 2 radiation at the modulation frequency f The mixer input signals xi and Xg each have two sinusoidal signal components with the frequencies fi and f of the modulation. We have x, = x,., Cos (2rt /, f + + <p "e (f,) - 2izf, 2Da / c) + XI2 cos (2jr / ^ f (p + <pHgifz) ~ / c)

(1)

[0035

^ HS ^ RS

<PHE <PHE

^ RE

^ RE

with xä = Xv cos (2rt £ f + (pHS (f,) + <pReß) - 2Kfp, i c)

(2)

+ x22 cos (2; r V Jr (pRS {fi) + <pRe (f.) - 2-xfzDz / c)

Sum of the initial phase of the synthesizer (21) and phase shift of the driver (23) and main emitter (1) at the frequency f; dependent on temperature and age;

Sum of the initial phase of the synthesizer (22) and phase shift of the driver (24) and reference emitter (2) at the frequency f; temperature and age dependent.

Sum of phase shift of the main receiver (5) and the transimpedance amplifier (10) at the frequency f; dependent on temperature and age;

Sum of phase shift of the main receiver (5) and the transimpedance amplifier (10) at the frequency f; dependent on temperature and age;

Sum of phase shift of the reference receiver (4) and the transimpedance amplifier (9) at the frequency f; dependent on temperature and age;

Sum of phase shift of the reference receiver (4) and the transimpedance amplifier (9) at the frequency f; dependent on temperature and age;

Dq measuring distance;

D1 constant and known device internal distance;

Dg constant and known device internal distance;

Dg constant and known device internal distance;

c speed of light in air.

By mixing (multiplying) the signals from Eq. (1) and (2) with the local oscillator signal of frequency f followed by low-pass filtering

= * 3.t cos (2 ^ 4fi ^ + 9hs ^ -) + ~ 2 ^ 2D0 i c)

+ x3 2 cos (2rt4p2f + 9ss (f2) + 9HS (U) + 9zfz (Ïzfî) ~ 2t: / 3D3 / c)

xt - X4i1cos (2tc ^ p / + <pHS (/ i) - -p / c)

+ x22 cos (2tifali + (pRS (f2) + <PzË (f2) + <PZfJfZF2) ~ 2Tzf2D2 / c)

(3)

(4)

with the intermediate frequencies

4fi = | ^ i-4oÌ (5)

and

^ ZF2 =% ~ Lo I (6)

By the low-pass filtering, the signal components with the sum frequencies fi + f and f + f, which also arise through the non-linear mixing process, are eliminated. Also, the noise level is thereby reduced.

They mean:

tpzF3 (fzFi) sum of the initial phase of the synthesizer (12), initial sampling phase of the ADC (18) and phase shift of the low pass (14) and amplifier (16) at the frequency f;

(pZF3 (fzF2) Sum of the initial phase of the synthesizer (12), the initial sampling pnase of the ADC (18) and the phase shift of the low-pass filter (14) and amplifier (16) at the frequency f;

<p (f) sum of the initial phase of the synthesizer (12), the initial sampling pnase of the ADC (17) and the phase shift of the low-pass filter (13) and the amplifier (15) at the frequency f;

(pZF4 (fzF2) Sum of the initial phase of the synthesizer (12), initial sampling pnase of the ADC (17) and phase shift of the low pass (13) and amplifier (15) at the frequency f

The IF signals x3 and x4 are now sampled synchronously with the A / D converters 17, 18. By discrete Fourier transform of the sampled signals x3 and x4 in block 19, the signal components with the various intermediate signals

6

CH 695 516 A5

separate the frequencies f and fZF2.

In Fig. 2a, a signal mixture x3 in the time domain with the frequency components f = 40 kHz and fZF2 = 60 kHz is shown as an example. On the right, the magnitude of the Fourier-transformed signal mixture is plotted against the intermediate frequency f. There are two sharp signal peaks at the respective intermediate frequencies. At other frequencies, the values of the spectrum are nearly zero. The different heights of the peaks result from different amplitudes of the signal components in the time domain (0.7 V, 0.4 V).

The values of the transformed signal mixture in the frequency domain are complex, i. H. it sits down according to

X3 (fZF) = Re {X3 (/>)} + j ■ \ m {X3 (fZF)}

composed of a real and an imaginary part. In Fig. 2, the amount is shown. From the complex values at the respective frequencies f and fZF2, by means of the Arctan

\ X, (fz.) \ = ^ 'L m2 [Xz {fZF}} - ¥ \ e2 [X 3 (fZF)}

t. From the complex We

Function according to

= ancten (lfn {X3 (fZF1)} / Re {X3 (/ tari)})

and

ç2 (fz) = arctane (lm {X3 (f ^)} / Re {X3 (fZF2)})

the phases of the separated signal components are determined at the intermediate frequencies f and fZF2 to be considered. For x3 from Eq. (3) they are

• Pi (^ i) = • PhsCT) + 9ne (fì) ^ PzfsÌ ^ ZFÌ) ~ / c (7)

and

9 2 (f2) = <pRs (f2) + 9HE (4) + 9ZFZ (fzn) "2B ^ s lc- (8)

For the phases of xd from Eq. (4) follows accordingly

93 $) = 9 ns i) + 9 ae (%) "*" 9 ~ 2y ^ 2, / c (9)

and

<P4 (4) = <Pw (4) + 9'fl € (4) + <P2F4 (4Fa) -27t4D2 / c. (10)

It is particularly advantageous if in the next step, the modulation frequencies fi and f are reversed, so that now the main emitter 1 radiation with the modulation frequency f and the reference emitter 2 emits radiation with the modulation frequency fi. As will be described further below, this measurement with exchanged modulation frequency considerably improves the unambiguous distance measurement. According to the procedure described above, now the phases

= Phs (4) + 9hs (^) 9ZF3 ^ ZF2) ~~ 271 ^ 2 ^ 01C (11)

92 (4) = 9rs (Q + <Pne (f,) + 9zfîVzf <) -Sxfpa! G (12)

93 (4) = 9 HS V *) + 9RS ih) + 9zf * {fan) - 2 «4 2DJc (13)

tpM) = 9rsWÌ) + 9R £ [fi) + 9zF * (fzFi) ~ 2ïc ^ D2 / c (14)

measured. By making the differences, the gin follows. (7-14)

<P, ß) - <PaVi) «9nsW) ~ 9ne (ft) + 9iFz (fzpù ~ 9zf ^ zfì) ~ 2j ^ 2D0 / c + 2jl /, 2Dt / C (15)

92 (4) ~ 9Af2) = 9he ^ 2) -9RS (f2) + 9zfi {fzF2) -9zFt (fzF2) ~ 2rc4D3 / c + 2% f2D2 lc (16)

9 ^ 2) ~ 9s (fi) = 9Hs (fz) ~ 9Rs (fz) + 9zf ^ zfz) ^ 9zFi (fzF2) ~~ 2izfz2DQ 1 c + 2nf22D, / c (17)

9f) ~ = 9hs (f - \) ~ 9r £ {ft) + 9zf3 (fzf \) ~ 9zf * {fzf \) ~ 2-'kfß3! C + 2tzfidî / c, (18) and by subtraction of the gin. (15) and (18) or (16) and (17) finally results

7

CH 695 516 A5

A (p {ft) = 2it / t12D,! C -2tzIDz! C + 2% fp3! c- ~ 2nf, 2D0! c + 2% n (19)

A <p (f2) = ~ 2Tif22D t c + 2% f2D2! c- 2izf2D3 Ì c + 2 izf22D0! c- 2 un. (20)

Since the phase can be clearly measured only in an interval of 0 to 2 k, but the measuring distance produces in most cases much larger phase shifts, is in the Gin. (19) and (20) introduce the integer number n of the full periods, which determines the total phase rotation in addition to the residual phase term. From the latter two gin. (19) and (20), the distance Dq to be measured and the period number n can now be unambiguously determined, since the device-internal distances D, Dg and D3 are constant and can be determined beforehand by measurement. The two modulation frequencies fi and f should be close enough together for both gin. (19) and (20) gives the same number of periods n. This ambiguity of the measuring distance is the reason for the permutation of the modulation frequencies according to the teaching of patent claim 2, since the additional measurement with exchanged frequencies provides the additional value of Eq. (19) independent equ. (20). These two independent equations yield unique values for n and Dq even for large measurement distances Dq.

Another advantage of the additional measurement with frequency interchange is that - like the Gin. (17) and (18) - achieve complete elimination of the main and reference receptor phases (<p (f), ^ h ^ '<pHE (fi)' ^ HE ^ y) and the phases of the IF region. This elimination is done by subfractionation of the gin. (15) and (18) or the Gin. (16) and (17).

The phase differences on the left side of the Gin. (19) and (20) result from the phase measurement.

On the one hand, a smaller frequency difference fi-f (for example a few 100 kHz, at f1 = 900 MHz) is desirable in order to unambiguously determine the integer period number n at large measuring distances (eg> 100 m) n in Gin. (19) and 20)). On the other hand, larger noise-related measurement errors occur at low frequency differences, so that n may be determined incorrectly under certain circumstances.

For even more accurate measurement of large distances and simultaneous accurate determination of the period number n, it is advantageous to use a second frequency pair f and f for intensity modulation in another measurement passage, which differs, for example, by 10 MHz from fi and f. With this frequency pair, the above-mentioned procedure is carried out again, in which case no interchanging is required, since the exact distance measurement is performed with the frequency pair f1 and f. Due to the larger frequency difference (for example, f-f1 = 10 MHz), a possible measurement error is further reduced, and the integer period number n can now be determined unambiguously even for very large measurement distances. By this measure specified in claim 18, the uniform moderate, small change in the intensity modulation frequencies can be used for weak, noisy measurement signals an error-free distance and period number determination by using a further frequency pair, e.g. f - 10 MHz, f - reach 10 MHz. In addition, by changing the measurement frequencies, optimal operating points can be found which lead to optimum signal-to-noise ratios. Due to bandpass filter tolerances, these optimum frequencies may differ slightly from device to device.

The method according to the invention and the distance measuring device based thereon are characterized in particular by the following advantages:

- All phase errors are eliminated by simultaneous measurement of the reference and the main emitter signal. Thus, all temperature, aging and receiving power-dependent phase errors of both the transmitting and the receiving unit are completely eliminated.

- The measurement accuracy is significantly improved.

- The reliability of the measurement results is significantly better.

- The meter is largely maintenance-free, since no mechanical switches or the like are needed.

- The simultaneous measurement of the reference and the main emitter signal reduces the measuring time and increases the measuring accuracy.

Claims (28)

claims 1. A method for optoelectronic distance measurement, in which - One of a main light emitter (1) emitted, intensity-modulated main light beam on the one hand to a remote measurement object whose distance (Dq) is to be measured from an observation site, and the light scattered there (36) via a receiving optics (37) on a photo Main receiver (5) passes, and on the other hand, a branched-off portion (33) of the main light beam is simultaneously passed over a first known reference distance (D ^ on a photoreference receiver (4); a reference light emitter also emitted by a reference light emitter (2), also intensity modulated, on the one hand via a second known reference distance (Dg) to the photo reference receiver (4) and on the other hand a part (35) of the reference light beam over a third reference distance (D) to the main receiver (5) arrives, and in which - The signals supplied by the main and the reference receiver of a comparative signal evaluation for obtaining an error-compensated measuring signal are supplied, characterized in that - The light intensities of the main and the reference emitter (1, 2) with different frequencies (f fg) are intensity-modulated simultaneously, 8th CH 695 516 A5 - The signal mixtures supplied by the main and the reference receiver, each containing a signal component with the intensity modulation frequency of the main emitter and a signal component with the intensity modulation frequency of the reference receiver, are each converted into an intermediate frequency range containing two frequency components, the one frequency component with the signal of the reference emitter and the other frequency component is formed with the signal of the main emitter, and that - For the comparative signal evaluation, the separation of the phase information contained in the two simultaneously incurred Zwischenfrequenzsi- signals carried out due to the different frequencies in the intermediate frequency range and the different modulation frequencies for the intensity modulation of main and reference light beam. 2. The method according to claim 1, characterized in that to achieve a clear distance measurement and to improve the measurement result time sequentially several measurement operations are performed, the modulation frequencies for the intensity modulation according to a fixed scheme on the one hand interchanged and on the other hand alike changed. 3. The method according to claim 1, characterized in that the output power of the main and / or the reference emitter is / are varied to adapt to different dynamic requirements of each measurement process. 4. The method according to claim 1, characterized in that as the main light emitter (1) and for the reference light emitter (2) lasers are used. 5. The method according to claim 4, characterized in that the fundamental wavelengths (A, A) of the two lasers (1, 2) are chosen differently. 6. The method according to claim 5, characterized in that by optical filtering crosstalk between the main beam and the reference beam associated Lichtsignalwegen is reduced. 7. The method according to claim 4, characterized in that for the comparative signal evaluation, the phases of the signal components of the signal mixtures (42; 43) in the intermediate frequency range by digital Fourier transform with evaluation of the real and imaginary parts of the Fourieriertierformierten in the frequency range signal mixtures at the respective intermediate frequencies (f and f) are determined. 8. The method according to claim 7, characterized in that the digital Fourier transformation to be supplied signal mixtures are first subjected to a low-pass filtering. A method according to claim 8, characterized in that the intermediate frequencies (f and f) and the sampling times of an analog-to-digital conversion (17, 18) preceding the Fourier transform, i. the measurement window of the respective digital sampling, be selected so that an integer number of periods of both the signal component with the first intermediate frequency (f) and the signal component with the second intermediate frequency (f) are in the measurement window of the digital scan. 10. The method according to claim 1, characterized in that on the main receiver (5) reaching proportion of the reference light beam is first diffusely reflected or scattered and only as Streuanteil together with the incident on the receiving optical Rückstreuanteil the main light beam to the main receiver (5) is passed. 11. The method according to claim 1 or 10, characterized in that on the reference receiver (4) reaching portions of the reference and main light beam initially diffusely reflected or scattered and passed only as scattered light components on the reference receiver (4). 12. An apparatus for performing the method according to claim 1 for optoelectronic distance measurement with two light emitters (1, 2) whose respective light beam is intensity-modulated, the light beam of the first light emitter designated as "main emitter" (1) being alignable on the one hand to a remote measurement object whose distance (D) is to be measured from an observation location, and on the other hand, a separated radiant passes through a first reference distance (D ^) to one of two photoreceptors (4, 5) referred to as the "reference receiver" (4) and the light beam of the second light emitter designated as the "reference emitter" (2) via a second reference distance (D) to the reference receiver (4) and a beam portion separated therefrom via a third reference distance (D) to the second photoreceiver (5) designated as the "main receiver", which also receives the portion of the light beam from the backscattered from the object to be measured Main emitter (1) is applied, - Each one the reference receiver (4) and the main receiver (5) associated signal mixer (7, 8), which convert the receiver signal mixtures in an intermediate frequency range, and with - An evaluation device (17 to 19) for determining the measuring distance (Dq) from the output signals of the two mixers (7, 8), characterized by a device by means of which the light beams emitted by the main emitter or by the reference emitter (1, 2) can be intensity-modulated simultaneously with respectively different frequencies (f fg). 13. The apparatus according to claim 12, characterized by a respective between the reference receiver (4) and the main receiver (5) and the respective associated mixer arranged transimpedance amplifier (9, 10) or impedance-controlled RF amplifier. M.Vorrichtung according to claim 12, characterized in that as the main receiver (5) and / or as a reference receiver (4) each have an avalanche photodiode is used. 15. The apparatus according to claim 14, characterized in that the avalanche photodiode of the main receiver (5) 9 CH 695 516 A5 and / or the avalanche photodiode of the reference receiver (4) are used as a direct mixer, wherein the respective received signal mixture is converted directly into the intermediate frequency range by modulation of the avalanche gain via a local oscillator signal (f10) generated by a local oscillator (20, 12). 16. The apparatus according to claim 15, characterized in that a LC oscillator is used as a local oscillator whose vibration-determining elements are mainly formed by the capacity of the avalanche photodiode of the main receiver and / or by the capacity of the avalanche photodiode of the reference receiver. 17. The apparatus according to claim 12, characterized in that as the main receiver (5) and / or as a reference receiver (4) each have a PIN photodiode is used. 18. The apparatus according to claim 12, characterized by an in the intensity modulation means (20-22) existing means for mutually exchanging the frequencies (f fg) of the reaching on the respective light emitter modulation signals. 19. The apparatus according to claim 18, characterized by a device for time-sequential mutual exchange of the frequency of reaching the respective light emitter intensity modulation signals. 20. The apparatus according to claim 18, characterized by a device for gieichmässig changing the frequency of the respective light emitter (1, 2) reaching intensity modulation signals. 21. Device according to one of the preceding claims 12 to 20, characterized by a in the beam path of the reference emitter (2) on the main receiver (5) reaching beam portion arranged scatterer (11), of which a scattered light component of the reference emitter (2) originating beam component to the main receiver (5). 22. Device according to one of the preceding claims 12 to 21, characterized by a in the beam path of the reference emitter (2) and the main emitter (1) on the reference receiver (4) reaching beam portions arranged scatterer (51), of the scattered light components of the reference emitter (2) and main emitter (1) coming beam portions on the reference receiver (4). 23. The device according to claim 12, characterized by means for varying the Lichtabgabeleistung the reference emitter (2) and / or the main emitter (1). 24. The device according to claim 12, characterized in that the optical wavelengths (A, A) of reference and main emitter are different. 25. The apparatus according to claim 24, characterized in that in the beam path from the main emitter (1) to the reference receiver (4) tuned to the output from the main emitter wavelength optical filter (52) and / or in the beam path from the reference emitter (2) to the main receiver (5 ) is arranged on the output from the reference emitter wavelength tuned optical filter (41). 26. The device according to claim 24, characterized in that the main emitter (1) or the reference emitter (2) is an edge emitting laser diode, a VCSEL or a light emitting diode is / are. 27. The apparatus according to claim 12, characterized by a common used for transmitting and receiving part of the master oscillator (20). 28. The apparatus of claim 12 or claim 15, characterized in that the two mixers (7, 8 and 4, 5) from the same local oscillator (20, 12) are controlled, the frequency (f) is selected so that in the intermediate frequency range in each case a signal mixture is present whose signal components with both intermediate frequency components (f fZF2) contains the phases at the modulation frequencies (ff). 10