CA2705969A1

CA2705969A1 - Optical receiver with monolithically integrated photodetector

Info

Publication number: CA2705969A1
Application number: CA 2705969
Authority: CA
Inventors: Anthony Chan Carusone; Tony Shuo-Chun Kao; Hemesh Yasotharan
Original assignee: University of Toronto
Current assignee: University of Toronto
Priority date: 2010-06-04
Filing date: 2010-06-04
Publication date: 2011-12-04

Abstract

An optical receiver includes a photodetector for detecting incoming optical data signals and an amplifier for providing signal gain and current to voltage conversion.
The detection signal generated by the photodetector can include a distortion component caused by an operating characteristic of the photodetector. A signal compensating circuit can reconstruct the received optical data signal by effectively canceling the distortion component. For this purpose, the signal compensating circuit can include a decision feedback equalizer implemented using at least one feedback filter matched to the operating characteristic of the photodetector causing the signal distortion so as to reproduce the distortion component for cancellation. Use of a control module can also configure the optical receiver in real time to account for other operating and environmental conditions of the optical receiver. Data rates in excess of 5Gbps can be realized in monolithic CMOS
photodetectors when the signal compensating circuit is properly matched.

Description

TITLE: OPTICAL RECEIVER WITH MONOLITHICALLY INTEGRATED PHOTO
DETECTOR
FIELD
[0001] Embodiments of the present invention relate generally to optical receivers, and more particularly to compensated optical receivers having monolithically integrated photodetectors.

INTRODUCTION

[0002] Optical receivers can be utilized in various different applications, such as local-area networks (LAN) and fiber-to-the-home (FTTH) interconnects, as well as in interfaces for optical storage systems, such as CD-ROM, DVD and Blu-Ray Disc.
In these applications, a photodetector can be used to convert incoming optical data signals into electrical detection signals for further processing, such as decoding, amplification, equalization, and compensation. In some types of optical data systems, the photodetector can be housed in a separate chip or as a standalone component and connected to other signal processing elements in the optical data system using bond wires or other connections. Although this solution allows for the use of high quality and high data rate photodetectors, extra overhead and assembly cost associated with the photodetector, as well as electrostatic discharge (ESD) problems and other parasitics associated with the bond wires can be some of the resulting drawbacks.

[0003] In other optical data systems, the photodetector can be monolithically integrated with other signal processing components on a single semiconductor substrate and implemented, for example, using standard integrated circuit (IC) technologies, such as complementary metal oxide semiconductor (CMOS), Silicon Germanium (SiGe) and mixed bipolar CMOS (BiCMOS) processes. Light detection in CMOS technology can be performed using a pn junction fabricated in the substrate, for example by appropriate doping of the semiconductor, and operated with a reverse bias voltage to create a depletion region. When an incoming optical data signal is received at the photodetector, electron-hole pairs (i.e., charge carriers) generated by the incident photons can be collected in an anode coupled to the depletion region for intensity measurement and optional post-detection processing in order to reconstruct the transmitted optical data signal.
Because the photodetector is monolithically integrated on the semiconductor, use of bond wires is minimized and overhead is reduced. Other advantages common to integrated devices, such as low cost and manufacturability, are also realized.

SUMMARY

[0004] In accordance with one aspect, there is provided an optical receiver comprising a photodetector, an amplifier and a signal compensation circuit.
The photodetector can generate a detection signal representative of an optical data signal received at the photodetector and having a distortion component caused by an operating characteristic of the photodetector. The amplifier can amplify the detection signal to generate an amplified detection signal. The signal compensation circuit can generate a reconstructed data signal from the amplified detection signal and can comprise a decision feedback equalizer matched to the operating characteristic of the photodetector, so that the distortion component of the detection signal is substantially suppressed in the reconstructed data signal.

[0005] The operating characteristic of the photodetector can comprise a diffusion current induced in the photodetector by the optical data signal.

[0006] The decision feedback equalizer can comprise a summer, a non-linear element and at least one filter. The summer can be configured to generate a compensated detection signal by subtracting a feedback compensation signal from the amplified detection signal. The non-linear element can be coupled to the summer to generate the reconstructed data signal from the compensated detection signal. The at least one filter can be coupled between the non-linear element and the summer in a feedback compensation loop to generate the feedback compensation signal based on the reconstructed data signal and can be configured to model the operating characteristic of the photodetector, so that the feedback compensation signal substantially reproduces the distortion component of the detection signal.

[0007] The non-linear element can comprise a signal quantizer, but alternatively can comprise a high-pass filter and a hysteretic comparator coupled to the high pass filter.

[0008] The decision feedback equalizer can comprise a plurality of filters coupled between the non-linear element and the summer in parallel in the feedback compensation loop, each filter configured to provide a respective portion of the feedback compensation signal. Each filter can be a single-pole continuous-time filter.
Alternatively, the plurality of filters can comprise at least one digital filter and at least one continuous-time filter, the at least one digital filter configured to compensate fast distortion components and the at least one continuous-time filter configured to compensate slow distortion components. In such cases, each at least one continuous-time filter can be a single-pole filter and the at least one digital filter can comprise a higher-order finite impulse response filter.
The decision feedback equalizer can comprise between three and five filters arranged in parallel in the feedback compensation loop.

[0009] The signal compensation circuit can further comprise a control module for configuring the decision feedback equalizer to match the operating characteristic of the photodetector by adjusting at least one parameter of the decision feedback equalizer. The at least one parameter of the decision feedback equalizer can comprise a time constant or a gain value for the at least one feedback filter.

[0010] The control module can comprise a dc extractor, a dc reference generator, a summer and a filter controller. The dc extractor can measure a dc component of the compensated detection signal. The dc reference generator can generate a reference dc component of the compensated detection signal. The summer can be configured to generate a compensation error signal representative of uncompensated distortion in the compensated detection signal by comparing the measured and reference dc components of the compensated detection signal. The filter controller can be configured to generate control values based on the compensation error signal used to adjust the at least one parameter of the decision feedback equalizer.

[0011] The dc reference generator can comprise a peak detector for generating an envelope signal representative of a pulse height of the optical data signal, and a scaler coupled to the peak detector for scaling the amplitude signal according to a bit distribution of the optical data signal to generate the reference dc component of the compensated detection signal.

[0012] The decision feedback equalizer can comprise at least one continuous-time filter implemented by a controllable RC-network. In that case, the filter controller can be configured to apply control signals to the RC-network based on the compensation error signal used to vary effective resistance and capacitance values of the RC-network.

[0013] The optical receiver can further comprise an equalizer coupled between the amplifier and the signal compensation circuit for providing high-frequency signal boosting.

[0014] The optical receiver can further comprise an ac coupling circuit coupled between the photodetector and the amplifier for suppressing low frequency components of the detection signal.

[0015] The photodetector can be a spatially modulated light detector, in which case the optical receiver can further comprise a subtractor downstream of the photodetector configured to generate the detection signal by subtracting a pair of differential detection signals generated by the spatially modulated light detector.

[0016] The photodetector can be integrated monolithically within the optical receiver on a semiconductor substrate. The optical receiver can be implemented in CMOS, SiGe and BiCMOS.

[0017] The optical receiver can have a bandwidth of at least 5 Gbps.
BRIEF DESCRIPTION OF THE DRAWINGS

[0018] A detailed description of various embodiments is provided herein below with reference, by way of example, to the following drawings, in which:

[0019] FIG. 1 is a schematic diagram of an optical receiver;

[0020] FIG. 2 is a graph of a typical response for the photodetector illustrated in FIG.
1 when implemented using CMOS, SiGe or BiCMOS;

[0021] FIG. 3 is a schematic diagram of the signal compensation circuit illustrated in FIG.1 in which the non-linear element includes a signal quantizer;

[0022] FIG. 4 is a schematic diagram of the signal compensation circuit illustrated in FIG.1 in which the non-linear element includes a filter and a hysteretic comparator;

[0023] FIG. 5 is a schematic diagram of the signal compensation circuit illustrated in FIG.1 in which a plurality of feedback filters is included;

[0024] FIG. 6A is a schematic diagram of the signal compensation circuit illustrated in FIG.5 in which each feedback filter is a continuous-time filter;

[0025] FIG. 6B is a schematic diagram of the signal compensation circuit illustrated in FIG.5 in which at least one of the plurality of feedback filters is a digital filter and at least one of the plurality of feedback filters is a continuous-time filter;

[0026] FIG. 7 is a schematic diagram of a digital finite impulse response filter that can be used to implement at least one of the plurality of feedback filters illustrated in FIG. 5;

[0027] FIG. 8 is a schematic diagram of a continuous-time finite impulse response filter that can be used to implement at least one of the plurality of feedback filters illustrated in FIG. 5;

[0028] FIG. 9 is a schematic diagram of a digital infinite impulse response filter that can be used to implement at least one of the plurality of feedback filters illustrated in FIG. 5;

[0029] FIG. 10 is a schematic diagram of a continuous-time infinite impulse response filter that can be used to implement at least one of the plurality of feedback filters illustrated in FIG. 5;

[0030] FIG. 11 is a schematic diagram of the signal compensation circuit illustrated in FIG.1 in which a control module for configuring the signal compensation circuit to match the photodetector is included;

[0031] FIG. 12 is a schematic diagram of the signal compensation circuit illustrated in FIG.1 in which an alternative control module is included; and [0032] FIG. 13 is a schematic diagram of the signal compensation circuit illustrated in FIG.1 in which an alternative control module is included.

[0033] It will be understood that reference to the drawings is made for illustration purposes only, and is not intended to limit the scope of the embodiments described herein below in any way. For convenience, reference numerals may also be repeated (with or without an offset) in the figures to indicate analogous components or features.

DETAILED DESCRIPTION OF EMBODIMENTS

[0034] Although CMOS and other IC photodetectors may conveniently minimize use of bond wires and reduce overhead, these types of photodetectors tend to generate significantly distorted detection signals due to their particular mechanisms of detection.
Light photons incident on the photodetector are absorbed either in the depletion region of the photodetector or deep into the underlying substrate depending on the penetration depth of the photon. Charge carriers generated within the depletion region are transported to the photodetector anode relatively quickly through carrier drift in the presence of the reverse biased electric field applied to the pn junction. However, those charge carriers generated deep in the underlying substrate are transported through carrier diffusion until they reach the depletion region, after which point carrier drift again becomes the dominant mode of transport to the anode. Compared to the drift velocity of electrons and holes in the presence of an electric field, diffusion tends to be an extremely slow transport process.

[0035] The penetration depth of 850-nm light, common in many present optical data systems, is much greater than the depletion regions typically found in many standard IC
technologies, which can be about 1-2 m below the surface. For example, CMOS, as well as many SiGe and BiCMOS, manufacturing processes create depletion regions of these or approximately these dimensions. Consequently, most photons of light in photodetectors fabricated using these IC processes are absorbed deep in the underlying silicon substrate where the resulting carriers are generated. These carriers slowly diffuse to the depletion region of the pn junction for transport to the detector anode. The slow diffusion mechanism tends to limit the available data rates of CMOS, SiGE, and BiCMOS
photodetectors to only a few hundreds of Mbps, assuming no form of downstream signal compensation is performed, because the long tail of the diffusion currents associated with one detection signal can interfere with and distort subsequent detection signals. For many present optical systems operating at data rates on the order of Gbps, the maximum available data rate of the CMOS, SiGE, or BiCMOS photodetector may be unacceptably slow. Accordingly, without some form of signal compensation, it may be preferable instead to use a standalone photodetector (which may be fabricated using other technologies that do not generally suffer from the same data rate limitations).

[0036] Several approaches are available to eliminate the negative effects of the slow diffusive carriers in order to improve the speed of monolithically integrated photodetectors.
For example, applying an extremely high reverse bias voltage to the pn junction, perhaps even higher than the available power supplies, can improve detector performance by extending the thickness of depletion region. By making the depletion region thicker so that more of the incident photons are absorbed within the depletion region, as opposed to the underlying silicon substrate, many of the diffusive carriers can be eliminated altogether and replaced with comparatively faster drift carriers. Generally higher data rates can therefore be achieved. However, this approach can seriously impact the reliability of the detector, for example, by creating a risk of the photodetector entering reverse breakdown resulting in large reverse currents and, hence, overheating. Another approach to limiting the effects of slow diffusive current is to introduce an electrically insulating layer between the photodetector and the carriers generated deep in the semiconductor substrate, thereby shielding the anode from the slow diffusive carriers. Generally, this approach is only partially effective, and may require additional fabrication steps that increase the overall cost of manufacture.

[0037] Another approach to the elimination of slow diffusive carriers involves the use of a spatially modulated light (SML) detector comprising alternately covered and exposed photodiodes. When light is incident on the surface of the SML detector, carriers generated in the depletion regions of the exposed diodes are almost immediately collected, while carriers generated deep in the silicon substrate underlying the exposed photodiodes will slowly diffuse to the surface. No carriers of either kind are generally created in the covered photodiodes. However, if the spatial distribution of the covered and exposed photodiodes is balanced, the slow diffusive carriers generated in the exposed photodiodes can have approximately equal probability of reaching the depletion regions of either the exposed or the covered photodiodes. The electron current measured at the covered photodiodes can then approximately represent the component of the electron current measured at the exposed photodiodes that is due to slow diffusion. Subtracting these two currents effectively cancels the slow diffusive carriers.

[0038] It is evident, however, that this approach can severely limit the sensitivity of the SML detector due to the portion of optical data signal incident on the covered photodiodes not being measured. A low-noise transimpedance amplifier can therefore be required in SML type photodetectors. For example, it may be necessary for the transimpedance amplifier to be capable of amplifying detection currents of as low as a few microamperes, with good signal to noise ratio and common mode rejection in order to limit the extent of sensitivity degradation in the photodetector. As the performance requirements of the low-noise transimpedance amplifier can drive up cost and overall complexity, use of an SML detector may not always be appropriate either.

[0039] As described herein, an optical receiver can be provided in which a signal compensation circuit comprising a decision feedback equalizer can be used to increase the effective data rate of monolithically integrated photodetectors. The decision feedback equalizer can be configured, for example by inclusion of a control module, to match one or more operating characteristics of the photodetector, so that a feedback compensation signal modeling a distortion component of the photodetector detection signal is generated by the signal compensation circuit. The feedback compensation signal can be generated using a plurality of feedback filters, each matched to a different characteristic part of the distortion component, so that the feedback compensation signal is synthesized piece by piece. An amplified detection signal can then be compensated by canceling the distortion component using the feedback compensation signal, thereby allowing the optical data signal to be reconstructed with the distortion component substantially suppressed. As an example, the control module can configure the decision feedback equalizer to almost fully compensate for the slow diffusive carriers typical of CMOS, SiGE and BiCMOS
photodetectors. Temperature effects and other operating or environmental conditions of the optical receiver can also be compensated using real-time, feedback control in the control module. Data rates of 5Gbps or more can then be realized using integrated photodetectors.

[0040] Referring initially to FIG. 1, there is illustrated a schematic diagram of an optical receiver 20. The optical receiver 20 comprises photodetector 22 coupled to amplifier 24, optionally, by way of ac coupling 26. Amplifier 24 is also coupled to signal compensation circuit 28, optionally, by way of equalizer 30. Thus, the photodetector 22 and the amplifier 24 can be directly coupled together in some cases, as can the amplifier 24 and signal compensation circuit 28 in some cases. The optical receiver 20 can be implemented in each of CMOS, SiGe and BiCMOS processes on a single semiconductor substrate, so that the photodetector 22 is monolithically integrated with the amplifier 24 and compensation circuit 28. However, it should be appreciated that the compensation circuit 28 could also be used in alternative configurations of the optical receiver 20 as well, such as configurations in which the photodetector 22 is implemented as a standalone device.

[0041] Photodetector 22 is exposed to optical data signal 32, which is transmitted to the photodetector for example through a fiber optic link or other optical communication channel. In response, the photodetector 22 generates a detection signal 34 that is representative of the received optical data signal 32. The detection signal 34 can include a data component, corresponding to the data encoded in the optical data signal 32, as well as a distortion component introduced in the photodetector 22. The distortion component can be caused by one or more operating characteristics or conditions of the photodetector 22. If ac coupling circuit 26 is included in the optical receiver 20, the detection signal 34 is passed to the amplifier 24 by way of ac coupling 26; otherwise the detection signal 34 can be passed directly to the amplifier 24, which can be a transimpedance amplifier (TIA). The amplifier 24 amplifies the detection signal 34 into an amplified detection signal 36, which is passed to the signal compensation circuit 28, in some cases, intermediately through equalizer 30 for signal processing. Signal compensation circuit 28 generates a reconstructed data signal 38, corresponding to the optical data signal 32 received originally at the photodetector 22, from the amplified detection signal 36. When the signal compensation circuit 28 is properly matched to the photodetector 22, the reconstructed data signal 38 can be substantially free of distortion and correspond closely to the received optical data signal 32.

[0042] Photodetector 22 can be implemented in one of many different IC
processes as described herein, such as CMOS, SiGE and BiCMOS. Thus, photodetector 22 can comprise one or more photodiodes (i.e., reverse biased pn junctions) coupled together in a silicon or other semiconductor substrate to generate the detection signal 34.
In some cases, the photodetector 22 can be a spatially modulated light (SML) detector, in which case the photodetector 22 can create a pair of differential detection signals.
Moreover, the amplifier 24 and optional ac coupling 26 can be fully differential, and the optical receiver 20 can further include a subtractor (not shown) coupled on the output of the amplifier 24 to generate the detection signal 34 by subtracting the differential detection signals. In either case, the cross-sectional area of photodetector 22 can be sized for interfacing with the optical communication link. For example, the area of photodetector 22 can equal or approximately equal 75 pm x 75 pm to facilitate coupling with multimode fibers. Also, the reverse bias voltage supplied to the photodetector 22 can be relatively large, for example about 3.3V, so that the optical receiver 20 can simultaneously achieve wide bandwidth and good responsivity overall.

[0043] Referring now to FIG. 2, there is illustrated a graph 50 showing a typical response for the photodetector 20 when implemented using CMOS, SiGe, or BiCMOS. The graph 50 plots time on the x-axis against normalized pulse height on the y-axis. Curve 52 represents the amplitude of the detection current induced in the photodetector 22 by a narrow pulse of light received at time, to, and lasting until about t1. For illustrative purposes, the amplitude of curve 52 is represented in arbitrary units normalized to the height of the received pulse of light. Thus, it should be appreciated that curve 52, because it is normalized, can represent either the detection signal 34 generated by the photodetector 22 or the amplified detection signal 36 generated by the amplifier 24 as the case may be.

[0044] In can be seen that different portions of curve 52 are characterized by potentially significantly different time constants. Curve 52 rises quickly according to a relatively short time during constant interval 54, which is defined between to and tj when the narrow pulse of light is incident on the photodetector 22. After reaching a maximum pulse height at ti, corresponding roughly to the end of the received pulse of light, curve 52 begins to drop back down toward zero. The rate of decay is quick during interval 56, which is defined between about t1 and t2, according to the same relatively short time constant that characterizes interval 54. A normalized amplitude of approximately 0.2 at t2 can be typical for the curve 52, though it may vary depending on how the optical receiver 20 and the photodetector 22 are configured. Around t2, however, curve 52 begins to decay much slower and continues to decay during interval 58 according to a relatively long time constant as compared to intervals 54 and 56. Thus, curve 52 can be characterized by a relatively short time constant during intervals 54 and 56, but a long time constant during interval 58 by comparison. As a result, the tail component of curve 52 (i.e.
intervals 56 and 58) has both a fast and slow portion.

[0045] The different time constants characterizing curve 52 during the different time intervals can correspond to different operating characteristics of the optical receiver 20 that dominate at different times. During intervals 54 and 56 when curve 52 rises and falls rather quickly, the response of the photodetector 22 can reflect generation large drift currents, but also bandwidth limitations of the amplifier 24. More specifically, when the pulse of light is incident on the photodetector 22, the drift current generated within the depletion region is large by comparison with the diffusive current generated deep in the underlying silicon substrate. It should be noted that diffusive carriers can be present simultaneously in time intervals 54 and 56, but are not as dominant as the drift current. The shape of the curve 52 during intervals 54 and 56 therefore reflects the faster speed of drift current. At the same time, the curve 52 during intervals 54 and 56 can also be rate limited by the bandwidth limitations of the amplifier 24. (From the standpoint of the photodetector 22, the input impedance of the amplifier 24 represents an effective load on the photodetector 22.) Thus, the rate of change of curve 52 during intervals 54 and 56 can also be subject to the finite bandwidth of the amplifier 24, which generally has a low-pass characteristic.
If included in the optical receiver 20, the optional ac coupling 28 can also rate limit the curve 52.

[0046] By about t2, substantially all of the drift current generated in the photodetector 22 has been collected and cleared leaving the comparatively slow diffusion current as the dominant component of curve 52. Thus, curve 52 assumes a comparatively slow time constant beyond t2 as curve 52 tends toward zero. As suggested by FIG. 2, the time constant of the diffusion current can be quite a bit slower than the time constant of the drift current. For example, the time constant of the diffusion current can be as much as two orders of magnitude slower. Combined with the fact that the undetected diffusion current at t2 can be sizable (i.e., about 20% of the maximum induced current), the tail component of curve 52 can have an exceptionally slow decay during interval 58 following the relatively fast decay during interval 56. Measured in terms of pulse widths, a decay lasting for one hundred or more pulses would not be uncommon for a photodetector fabricated in present IC technologies. As will be explained further below, the composite nature of the tail component having both a fast and slow portion, which is typical of an IC
photodetector, can be taken into account in the signal compensation circuit 28.

[0047] The received optical data signal 32 can comprise data encoded in a sequence of light pulses. Unless the data rate of the optical data signal 32 is slow enough, the tail component of the diffusion current associated with one received pulse of light can interfere with subsequently received pulses of light. In other words, with a fast enough data rate, those subsequent pulses of light can be received at the photodetector 22 before the diffusion current associated with previous pulses has had sufficient time to decay. Thus, to transmit the optical data signal 32 at a reasonably high data rate, the detection signal 34 generated by the photodetector 22 will generally include a distortion component, in addition to a data component (corresponding to the encoded data), which is attributable at least partly due to the diffusion current induced in the photodetector 22. The signal compensation circuit 26 can be optimized to compensate for the undesirable diffusion current when reconstructing the optical data signal 32. Larger effective data rates of 5Gpbs or more therefore become realizable in IC photodetectors. These large data rates can be realized simultaneously with the other associated advantages of IC
photodetectors mentioned previously, such as size and noise performance.

[0048] Referring back to FIG. 1, ac coupling 26 can comprise a resistor-capacitor network arranged so as to couple a high-frequency component of the detection signal 34 to the input of the amplifier 24. For example, the ac coupling 26 can simply comprise a capacitor in series between the photodetector 22 and the amplifier 24. If the photodetector 22 is an SML detector, the ac coupling 26 can comprise a capacitor for coupling each differential detection signal generated by the photodetector 22 into a corresponding differential input of the amplifier 24. To adjust the overall frequency response of the optical receiver 20, the ac coupling 26 can further comprise one or more resistors connected between the input of the amplifier 24 and the power supply or supplies of the optical receiver 20. Again, if the photodetector 22 is an SML detector, one or more resistors can be connected to each differential input of the amplifier 24.

[0049] As should be appreciated, capacitance and resistance values can be selected so as to attenuate low-frequency components of the detection signal 34. For example, capacitance and resistance values can be selected so as to attenuate the slow diffusion current appearing in the detection signal 34, which occurs at low frequency compared to the data rate of the optical data signal 32. At the same time, however, some attenuation of the faster drift currents, which are mostly responsible for transmitting the encoded data component of the optical data signal 32, may also occur. Inclusion of ac coupling 26 can therefore attenuate both the distortion and data components of the detection signal 34.
Signal compensation circuit 28 can be used to restore some of the low-frequency content lost due to ac coupling 26, thereby reconstructing the original optical data signal 32. As will be seen, signal compensation circuit 28 can compensate for the effects of ac coupling either with or without the use of feedback filters.

[0050] Amplifier 24 can be a transimpedance amplifier (TIA) having a large feedback resistor selected to achieve a high transimpedance gain. As a result, the amplified detection signal 36 generated by the amplifier 24 can be large relative to the noise contributions from later components of the optical receiver 20, which results in good signal-to-noise ratio in the reconstructed data signal 38. While increasing transimpedance gain, the large feedback resistor can also decrease the effective bandwidth of the amplifier 24, which varies inversely proportional to the size of the feedback resistor and can be approximated by A, BW =
27rRFC,,, where: A, represents the open-loop gain of the amplifier 24, C;,, represents the equivalent capacitance at the input to the amplifier 24, and Rf represents the feedback resistance.
Increasing the open-loop gain Ac can counteract some of the bandwidth reduction due to selection of a large feedback resistor Rf, but can also lead to gain peaking in the frequency response of the amplifier 24 if an insufficient phase margin is set. A
feedback capacitor in parallel with the feedback resistor can eliminate or reduce the gain peaking, but can also result in further bandwidth reduction.

[0051] Alternatively, a negative Miller capacitance can be incorporated into the core of the amplifier 24 as a way of extending the dominant pole of the amplifier 24 and thereby increasing its bandwidth. Extension of the dominant pole can also tend to increase the phase margin of the amplifier 24, thereby allowing the open-loop gain Ac to be increased without negatively impacting on the overall stability of the amplifier 24.

[0052] Signal compensation circuit 38 can be configured, as shown in FIG. 1, comprising a decision feedback equalizer 39 implemented by a summer 40, a non-linear element 42 and a feedback filter 44 coupled together to form a feedback compensation loop 45. The non-linear element 42 can be included in the forward branch of the feedback compensation loop 45 and configured to generate the reconstructed data signal 38 by transforming a compensated detection signal 46 generated by the summer 40. The feedback filter 44 can be included in the reverse branch of the feedback compensation loop 45 and configured to generate a feedback compensation signal 48 from the reconstructed data signal 38. The summer 40 can then be configured to generate the compensated detection signal 46 by subtracting the feedback compensation signal 48 provided by the feedback filter 44 from the amplified detection signal 36 provided by the amplifier 24. The signal compensation circuit 38 can be configured differently according to whether or not ac coupling 26 has been included.

[0053] The amplified detection signal 36 includes both a data component and a distortion component, for example, due to the slow tail component of the diffusion current generated in the photodetector 22. To reconstruct the original optical data signal 32 from the detection signal 34, the signal compensation circuit 28 generates the feedback compensation signal 48 to model the distortion component of the amplified detection signal 36, which is then used to cancel the distortion component when the feedback compensation signal 48 is subtracted from the amplified detection signal 36 in the summer 40. To provide an accurate reproduction of the distortion component, the decision feedback equalizer 39 can implement a transfer function modeling the distortion response of the photodetector 22 to a short pulse of light. Accordingly, the decision feedback equalizer 39 can be matched to one or more operating characteristics of the photodetector 22 being modeled. Operation of the signal compensation circuit 28 can be understood intuitively.

[0054] It can be assumed that the amplified detection signal 36 is representative of a continuous bit pattern encoded into the optical data signal 32, and that any transients in the feedback loop have settled so that the signal compensation circuit 28 is operating in a steady state. If the feedback filter 44 has been properly matched to the photodetector 22, the reconstructed data signal 38 will comprise a bit pattern identical to the bit pattern encoded originally in the optical data signal 32, once the signal compensation circuit 28 settles and achieves steady state. As a result, the input to the feedback filter 44 (i.e., the reconstructed data signal 38) comprises a sequence of short pulses corresponding closely to the sequence of pulses received at the photodetector 22. As the transfer function implemented in the feedback filter 44 models the distortion component of the amplified detection signal 36 due to a single pulse of light, the output generated by the feedback filter 44 (i.e., the feedback compensation signal 48) will effectively reproduce the distortion component of the amplifier detection signal 36 for the entire particular bit pattern encoded in the optical data signal 32. By comparing the amplified detection 36 with the feedback compensation signal 48, the distortion component of the amplified detection signal 36 can be substantially eliminated in the compensated detection signal 46.

[0055] Non-linear element 42 can then be used for shaping of the compensated detection signal 46 into a square wave to provide the reconstructed data signal 38. In this way, the reconstructed data signal 38 can be effectively a continuous-time digital signal, which can then be provided to an analog to digital converter (not shown) for sampling and conversion into a pure digital signal if desired. The order of the feedback filter 44 can be selected depending on the required accuracy of the feedback compensation signal 48.
Theoretically, non-linear element 42 could be omitted altogether if a complex and accurate enough feedback filter 44 is designed so that complete distortion cancellation is achieved and the compensated detection signal 46 is already essentially an ideal pulse train without the benefit of further shaping in the non-linear element 42. In that case, the compensated detection signal 46 could be provided directly as the reconstructed data signal 38 (and thus also to the input of the feedback filter 44.) However, inclusion of the non-linear element 42 can ease requirements for the order of the feedback filter 44, which can result in generally simpler and more cost-effective implementations. The quantizing function of the non-linear element 42 can also contribute to a faster overall response for the signal compensation circuit 28. As will be explained more below, inclusion of the non-linear element 42 can also provide a basis for calibration and control of the decision feedback equalizer 39.

[0056] If the ac coupling 26 has been included in the optical receiver 20, the signal compensation circuit 28 can be modified by exclusion of the feedback filter 44. With its high-pass characteristic, the ac coupling 26 can be configured to suppress substantially the entire distortion component of the detection signal 34. However, because the ac coupling 26 does not necessarily distinguish between the fast drift currents and the slow diffusive currents, some attenuation of both can occur resulting in loss of data components as well as suppression of distortion components. The non-linear element 42 can be utilized effectively to restore some of the lost low-frequency content, for example through signal quantization, thereby producing the reconstructed optical data signal 38.
Although it is possible to omit the feedback filter 44 when the non-linear element 42 is used in this way, it is also possible to include the feedback filter 44 for substantially the same use.

[0057] Referring now to FIG. 3, the signal compensation circuit 28 is illustrated in which a signal quantizer 60 is used to realize the non-linear element 42.
Signal quantizer 60 can be a binary (i.e., two-level) quantizer implemented using a high-gain comparator or differential amplifier, such as an op-amp, configured to compare the compensated detection signal 46 against an appropriate threshold level specified somewhere between the two defined quantization levels. Thus, the output of the signal quantizer 60 can be pulled up to a high-voltage level (e.g., equal to the positive power supply) when the compensated detection signal 46 is greater than the threshold level, and pulled down to a low-voltage level (e.g., equal to the negative power supply) when the compensated detection signal 46 is less than the threshold level. The resulting quantization of the compensated detection signal 46 can generate the reconstructed data signal 38 as a pulse train wave. Additional circuit components can be included in the signal quantizer 60, for example, to improve its frequency response.

[0058] Referring now to FIG. 4, the signal compensation circuit 28 is illustrated in which a combination of filter 70 and hysteretic comparator 72 is used alternatively to realize the non-linear element 42. As illustrated, filter 70 is coupled to the output of the summer 40 to receive the compensated detection signal 46. Hysteretic comparator 72 can then be coupled to the output of the filter 70 to generate the reconstructed data signal 38 from the intermediate signal 74 generated by the filter 70. For example, filter 70 can be a high-pass filter with a passband defined so as to suppress the low-frequency distortion.
In doing so, intermediate signal 74 can be generated so as to comprise a positive-going pulse for each rising (low-to-high) transition in the optical data signal 32 and a negative-going pulse for each falling (high-to-low) transition in the optical data signal 32. The hysteretic comparator 72 then generates the reconstructed data output 38 as a square wave toggled from low to high whenever a positive-going pulse is observed in the intermediate signal 74, and toggled from high to low whenever a negative-going pulse is observed. In doing so, the low-frequency component of the optical data signal 32 is restored without significant distortion.

[0059] Hysteretic comparator 72 can offer similar yet improved performance relative to signal quantizer 60 on account of input-output hysteresis. Thus, the output of the hysteretic comparator 72 can be pulled up to a high-voltage level (e.g., equal to the positive power supply) when the intermediate signal 74 rises above a first threshold level, and pulled down to a low-voltage level (e.g., equal to the negative power supply) when the intermediate signal 74 drops down below a second threshold level, which is different from and generally less than the first threshold level. If a common threshold level is used in both the upward and downward directions, as would be the case in the signal quantizer 60, then small voltage oscillations on the comparator input (e.g., due to noise) could cause rapid transitions between the low and high voltage levels on the output. However, this occurrence can be prevented by specifying two different input threshold levels depending on the current state of the output, as is done in hysteretic comparator 72 but not signal quantizer 60.

[0060] Referring now to FIG. 5, the signal compensation circuit 28 is illustrated explicitly using a plurality of filters 80,...80N to realize the feedback filter 44. The plurality of filters 80,...80N can be included in the feedback compensation loop between the output of the non-linear element 42 and corresponding inputs to the summer 82 so that the individual filters in the plurality of filters 80,...80N are connected together in parallel configuration.
Each individual filter can also be configured to generate a respective feedback compensation signal 48,...48N that are synthesized together in the summer 82 to generate the overall feedback compensation signal 48. Though summer 82 is illustrated in FIG. 5 explicitly as a discrete component, it should be appreciated that the summer 82 could alternatively be rolled into summer 40, so that the respective outputs of the filters 80, ...80N
are coupled directly into the summer 40. Thus, feedback compensation signal 48 would, in this case, be implicitly generated within the summer 40.

[0061] The plurality of filters 80,...80N can be configured, as required, to match the one or more operating characteristics of the photodetector 22 being compensated with the signal compensation circuit 28 in the aggregate. In other words, the plurality of filter 80,...80N can be designed to collectively simulate a single filter (e.g., feedback filter 44 shown in FIG. 1) designed to reproduce the distortion component of the amplified detection signal 36. The distortion component can again be caused by one or more operating characteristics of the photodetector 22, such as slow diffusive current associated with CMOS photodetectors. For example, each filter 80,...80N individually can be a single pole (i.e., first-order) low-pass filter defined by a dc gain and time constant.
The dc gains and time constants of the plurality of filters 80,...80N can also be generally different from each other, so that each respective feedback compensation signal 48,...48N can make an aggregate contribution to the feedback compensation signal 48. Alternatively, one or more of the plurality of filters 80, ...80N can be higher-order filters having more than one pole.

[0062] The number of individual filters 80,...80N is also variable depending on the desired complexity and accuracy of the signal compensation circuit 28.
Increasing the number of filters in the plurality of filters 80,...80N can result in closer matching of the photodetector 22 and reproduction of the distortion component of the amplified detection signal 36. However, increased complexity and bulk can be the tradeoff. In some cases, between three to five filters 80,...80N can be utilized; however, clearly more or less than this number could also be utilized in the signal compensation circuit 28.
Also, the number of individual filters 80,...80N can vary depending on the degree of distortion compensation provided by other components of the optical receiver 20. For example, the number can be reduced if the photodetector 20 is an SML detector, as this detector configuration already suppresses diffusion current. The same result could follow if the ac coupling 26 is included and used to suppress the low-frequency diffusion current.

[0063] Referring back to FIG. 2, curve 52 illustrates a typical response of the photodetector 22 to a short pulse of light can be broken into different intervals characterized by generally different time constants. The plurality of filters 80,...80N included the decision feedback equalizer 39 can be configured so that individual filters are matched to different portions or characteristics of the curve 52. A first filter (e.g., 801) can be matched to the fast tail component occurring during interval 56 by extracting the dc gain and time constant characterizing that portion of the curve 52, and designing a suitable low-pass filter based on these parameters, though it is not necessary for the first filter 80, to have only a single pole. As will be explained more fully below, these parameters of the curve 52 can be extracted by offline testing of the optical receiver 20 using a very low data rate test signal so that the entire curve 52 can be captured and subjected to frequency analysis. Bandwidth limitations of the amplifier 24 can also be taken into consideration when the curve 52 during interval 56 is being characterized. The additional filters 802...80N
can then be designed using the same general approach to match the transition point at t2 and slow tail component of the curve 52 occurring in interval 58. Amplifier bandwidth limitations, which only dominate at the fast parts of curve 52, can be neglected here. As the output of each individual filter 80,...80N is summed together in the summer 82 (or alternatively 40), the distortion response of the photodetector 22 can be synthesized piece by piece by designing each filter individually to match a different portion of the overall photodetector response.

[0064] Typically, the dc gain of the first filter 80, can be larger than the dc gains of any additional filters 802...80N. The time constant of the first filter 80, can also typically be faster than the time constants of the additional filters 802...80N. As seen in FIG. 2, the curve 52 drops to about 20% of its normalized height between t, and t2, which is a relatively brief interval of time as compared to the length of the long tail appearing after t2. The rate of decay of curve 52 during interval 56 therefore is relatively fast by comparison. Intuitively, a fast pole to synthesize the part of curve 52 occurring in the interval 56 will have little contribution during interval 58, despite a large dc gain, because its fast decay would be essentially zero-valued throughout the whole of the interval 58. Moreover, one or more additional slower poles to synthesize curve 52 during interval 58 can have little contribution during interval 56, despite having a slow decay, by keeping the dc gain of these additional poles relatively small. Optionally, one or more filters of the filters 80,...80N can also be designed to have intermediate poles located between the fast time constant characterizing interval 56 and the slow time constant characterizing interval 58, so as to provide better modeling of the transitional period between the two intervals 56 and 58. To a reasonable degree of error, therefore, the individual filters 80,...80N can be designed independently.
However, as will be explained in more detail below, feedback control can also be incorporated into the signal compensation circuit 28 to adjust the characteristics (i.e., dc gains and time constants) of the filters 80,...80N for better overall performance taking different operating characteristics of the optical receiver 20 into account, such as temperature, component aging, and data rate.

[0065] The plurality of filters 80,...80N are generally not restricted to being only first-order filters and can comprise one or more higher-order filters in addition to, or in place of, the single-pole filters 80, ...80N illustrated explicitly in FIG. 5. For example, the first filter 80, designed to match the fast tail component of curve 52 can be a higher-order filter, while each of the one or more of the filters 802...80N designed to match the slow tail component of curve 52 can be single-pole-order filters. Other configurations are possible as well.
Moreover, as should be appreciated, a high-order filter can be implemented equivalently as one or more single-order filters depending on the number of poles in the higher-order filter.
As will be explained in more detail below, it may be convenient to implement the plurality of filters 80 using only, or mostly, single-pole filters to provide simpler control over the dc gains and time constants of the individual filters 80,...80N.

[0066] Referring now to FIGS. 6A and 6B, the signal compensation circuit 28 is illustrated in which different arrangements and types of filters are used to implement the plurality of filters 80,...80N. In FIG. 6A, each of the filters 80,...80N is illustrated as a single-pole, continuous time filter having a low-pass characteristic. In FIG. 6B, the first filter 80, is illustrated as a higher-order, finite impulse response digital filter, while the additional filters 802...80N are illustrated as single-pole continuous-time filters. Due to the slow diffusive current generated by the photodetector 22, which results in the characteristic long tail evidenced in curve 52 of FIG. 2, implementing each individual filter 80, ...80N digitally (as either a finite impulse response or infinite impulse response filter) could result in unduly complex filter design. In other words, the extreme length of the tail component of curve 52 could require design of very slow and very bulky digital filters. This could be the case because a number of very high-order filters are required, or equivalently because a very large number of lower-order filters are required. It may therefore be convenient instead to implement the plurality of filters 80 using continuous-time configurations as shown in FIG.
6A, for example based on controllable resistor-capacitor (RC) networks fabricated on a semiconductor substrate.

[0067] Alternatively, as illustrated by FIG. 6B, the first filter 80, can be implemented digitally, while the additional filters 802...80N can be implemented using continuous-time configurations. Because the first filter 80, can comprise a relatively fast pole matched to the fast tail component of curve 52, as compared to the relatively slow poles matched to the slow tail component, filter bulk and complexity may not be as significant a consideration for the first filter 801. Thus it may be convenient to implement the first filter 80, but not the additional filters 802...80N digitally in order to exploit some of the performance advantages of digital filters. For example, digital filters tend to be less subject to component tolerances and non-linearities, as well as operating or environmental conditions like temperature.
Because digital filters store filter coefficients in memory, as opposed to realizing the coefficients using filter components, digital filters tend also to be more stable than continuous-time filters. If the filter order can be kept moderately low, therefore, digital filters can be preferred to analog filters. Though as described herein, the relative disadvantages associated with analog filters may be preferable to the bulk and slow computational performance associated with very high-order digital filters. It should also be appreciated that the permutations shown explicitly in FIGS. 6A and 6B are exemplary only, and that other permutations, both in terms of filter type and order, may be apparent as well.

[0068] Referring now to FIG. 7, there is illustrated a possible implementation of a digital FIR filter 180 used to implement at least one of the plurality of filters 80 included in the feedback compensation loop 45. The digital FIR filter 180 comprises a plurality of clocked flip-flops 1821...182N, a plurality of mixers 1840...184N, and a summer 186. The plurality of flip-flops 182,...182N can be arranged as illustrated in a cascade formation and driven by a common clock signal clk. By receiving the reconstructed data signal 38 into a first flip-flop 1821, the plurality of flip-flops 182,...182N can function as a progressive delay stage. Thus, relative to an arbitrary reference time, the output of the first flip-flop 182, can be the reconstructed data signal 38 delayed by one clock cycle, the output of the second flip-flop 1822 can be the reconstructed data signal 38 delayed by two clock cycles, and so on, so that the output of the Nth flip-flop can be the reconstructed data signal 38 delayed by N clock cycles. It should be appreciated that, as the reconstructed data signal 38 is effectively a continuous time representation of a digital signal, the outputs of the plurality of flip-flops 182 can be essentially the same reconstructed data signal 38 delayed by a corresponding number of clock cycles. It should also be appreciated that the number of flip-flops in the plurality of flip-flops 182 can be related to the order of the digital FIR filter 180.
As described herein, for accurate matching to the slow tail component of curve 52, the order of the digital FIR filter 180 could be anywhere from one to in the hundreds.

[0069] The plurality of mixers 1840...184N can be coupled respectively to the outputs of the plurality of flip-flops 1821...182N, with the exception that mixer 1840 can be coupled to the input of flip-flop 182, in order to receive the reconstructed data signal 38 without delay. Coefficients h0... hN can be supplied respectively to the mixers 1840...184N to generate weighted outputs, which are then summed together in summer 186 and outputted as the feedback compensation signal 48N. (The configuration shown in FIG. 7 can be used for each individual filter 80,...80N in the plurality of filters 80.) The coefficients h0...hN can be computed based on the desired performance characteristics (e.g., order, gain, frequency response) for the digital FIR filter 180. Optionally, the feedback compensation signal 48N can also be smoothed before or after being outputted.

[0070] Referring now to FIG. 8, there is illustrated a possible implementation of a continuous-time FIR filter 280 used to implement at least one of the plurality of filters 80 included in the feedback compensation loop 45. The continuous-time FIR filter 280 is similar in configuration to the digital FIR filter 180 illustrated in FIG. 7 but implemented in continuous-time. Accordingly, the continuous-time FIR filter 280 comprises a plurality of delay elements 282, ... 282N, a plurality of mixers 2840 ... 284N, and a summer 286. The delay elements 282,...282N can again be cascaded to progressively delay the reconstructed data signal 38, received into the first delay element 2821, by a time interval i.
For example, the delay elements 282,...282N can be micro transmission lines with an associated end-to-end delay equal to the interval t, though other types and configurations of delay elements 282....282N may be apparent. As in FIG. 7, the plurality of mixers 2840...284N can be coupled respectively to the delay elements 282,...282N to scale the delayed versions of the reconstructed data signal 38 by the appropriately computed coefficients a0... aN for summation in summer 286. Optional smoothing can also be applied to the feedback compensation signal 48N at the output of the summer 286.

[0071] Referring now to FIG. 9, there is illustrated a possible implementation of a digital infinite impulse response (IIR) filter 380 used to implement at least one of the plurality of filters 80 included in the feedback compensation loop 45. The digital IIR filter 380 differs in configuration from the digital FIR filter 180 and continuous FIR filter 280 in so far as the filter output (i.e., feedback compensation signal 48N) is fed back to give the digital IIR filter 380 its infinite impulse response. Accordingly, the digital IIR
filter 380 comprises a plurality of flip-flops 382,...382N, a plurality of mixers 3840 ... 384N, and a plurality of summers 386, ... 386N, connected as shown. The plurality of summers 386,...386N are interleaved with the plurality of flip-flops 382,...382N in cascade formation and coupled to the respective outputs of the plurality of mixers 384, ... 384N. A common clock signal clk is used to trigger the plurality of flip-flops 382, ... 382N, and filter coefficients d1...dN are provided to the plurality of mixers 384, ... 384N. The reconstructed data signal 38 is provided to a final pair consisting of flip-flop 382N and summer 386N. In the arrangement shown, the present output of the digital IIR filter 380 can equal a weighted summation of past output values and the reconstructed data signal 38, as required for an IIR filter. As before, the filter coefficients dl...dN can be designed to provide the digital IIR filter 380 with desired performance characteristics. For example, the filter coefficients d1...dN can be designed so that the digital IIR filter 380 is matched to the photodetector 22 and the overall response of the plurality of filters 80 accurately estimates the distortion component of the amplified detection signal 36 introduced by the operating characteristics of the photodetector 22.

[0072] Referring now to FIG. 10, there is illustrated a possible implementation of a continuous-time infinite impulse response (IIR) filter 480 used to implement at least one of the plurality of filters 80 included in the feedback compensation loop 45. The continuous time IIR filter 480 can be implemented, for example, using LCR network 488.
Through appropriate configuration of the LCR network 488, as should be appreciated, continuous-time IIR filter 480 can implement some arbitrary response of the form, H(s) bMsM +b,,_,sM-'+...+b,s+b0 =
CNSN + CN_,SN-' + ... + C1S + Cp As before, the filter coefficients bo... bM and co... cm, as well as the lumped circuit elements (resistors, capacitors, inductors, etc.), can be designed to provide the continuous-time IIR

filter 480 with desired performance characteristics, for example, to match the photodetector 22 response to a pulse of light.

[0073] An IIR filter of either type illustrated in FIGS. 9 and 10 can be effective for compensating the slow tail part of the distortion component of the amplified detection signal 36. As mentioned previously, FIR filters could, for the same purpose, have undue complexity and bulk issues due to the extreme length of the tail part (reflecting diffusive current in the photodetector 22). However, any of the filter implementations illustrated in FIGS. 7-10 could be appropriate for compensating the fast part of the distortion component, which is affected by drift current and the frequency characteristics of the amplifier 24 predominately. Because this part of the distortion component is characterized by a fast time constant in comparison, filter complexity is less of an issue. Either IIR or FIR, as well as digital or continuous-time, types of filters could be appropriate.

[0074] Referring now to FIG. 11, there is illustrated a signal compensation circuit 128 that utilizes a control module 90 to configure the signal compensation circuit 128 for the photodetector 22. The signal compensation circuit 128 is like the signal compensation circuit 28 illustrated in FIG. 1, for example, but further including the control module 90.
Elements common to the two signal compensation circuits 28 and 128 will not be discussed in detail. In the absence of ac coupling 26, control module 90 can be used to adjust one or more parameters of the signal compensation circuit 128 so that the decision feedback equalizer 39 is matched to, and thereby effectively compensates, for the one or more operating characteristics of the photodetector 22 causing distortion to the detection signal 34. For example, the control module 90 can configure the feedback filter 44 to reproduce the distortion component of the amplified detection signal 36 due to slow diffusive current generated in the photodetector 22. Configuration of the decision feedback equalizer 39 can also account for the operating temperature and/or supply voltage of the optical receiver 20, the data rate or received signal amplitude of the optical data signal 32, operating, physical characteristics (e.g., geometry, semiconductor dopant levels) of the photodetector 22, as well as component aging. However, if ac coupling 26 has been included in the optical receiver 20, then an alternative to control module 90 may be utilized instead to configure the decision feedback equalizer 39, or no control module at all in some cases.

[0075] Control module 90 can be coupled to the output of the summer 40 to receive the compensated detection signal 46 as a control input, and can further be coupled to the feedback filter 44 to provide one or more control values to the feedback filter 44 as outputs.
As explained more fully below, a bit frequency signal p can also be provided to the control module 90. When the signal compensation circuit 128 is closely matched to the response of the photodetector 22, the feedback compensation signal 48 should accurately reproduce the component of the amplified detection signal 36 representing distortion. By subtracting the feedback compensation signal 48 from the amplified detection signal 36, the compensated detection signal 46 should also then be substantially a square wave. If the distortion component of the amplified detection signal 36 has been fully compensated (resulting ideally in a perfect square wave), the dc component of the compensated detection signal 46 (i.e., its average value) will generally depend on the amplitude of the square wave and the bit distribution of the optical data signal 32. A balanced bit distribution, for example, would result in a dc component equal to one-half the square wave amplitude.

[0076] On the other hand, if the distortion component of the amplified detection signal 36 has not been fully compensated, the compensated detection signal 46 will not be an ideal square wave. The dc component of the compensated detection signal 46 may not then depend just on the bit distribution p of the optical data signal 32.
Uncompensated distortion remaining in the compensated detection signal 46 can skew the dc component up or down from its expected, or reference, level. Comparison of the measured and reference dc components can therefore indicate whether or not the amount of compensation provided is adequate. Adjustment to the decision feedback equalizer 39 can then be made offline (e.g. manually) or online (e.g. using feedback control).

[0077] Accordingly, control module 90 can comprise dc extractor 92, dc reference generator 94, and summer 96 arranged as shown to generate a compensation error signal 98, which is representative of uncompensated distortion remaining in the compensated detection signal 46. Each of the dc extractor 92 and dc reference generator 94 can be coupled to the output of the summer 40 in order to receive the compensated detection signal 46. The dc extractor 92 is configured to measure the dc component of the compensated detection signal 46. For example, the dc extractor 92 can comprise a low-pass filter, an integrator, or some other component suitable for measurement of dc components as will be appreciated. The measured dc component can then be provided to the summer 96 for comparison with a corresponding reference dc component generated by the dc reference generator 94.

[0078] The dc reference generator 94 can comprise a peak detector 100 and a scaler 102 coupled to the output of the peak detector 100. The peak detector 100 can be configured to generate a signal representing an envelope of the compensated detection signal 46. For example, the peak detector 100 can comprise a fast track and hold circuit or some other component suitable for tracking envelopes as will be appreciated.
Assuming essentially complete compensation of the distortion component, the compensated detection signal 46 will be substantially a pulse train and the envelope signal generated by the peak detector 94 should be nearly constant at a level equal to the height or amplitude of the pulses in the pulse train. By multiplying the envelope signal with the bit distribution p, the scalar 102 generates a reference dc component for the ideal case of a fully compensated detection signal 46. For example, if the distribution of high voltages (digital "1") compared to low voltages (digital "0") is approximately 0.5, then the dc component of the compensated detection signal 46 will be approximately half the height the envelope of the pulse train. In general, if the distribution of high voltages to low voltages is equal to p (0 s p :r. 1), then scaling the envelope signal by the bit distribution p can be used to specify the reference dc component corresponding to complete distortion compensation.

[0079] The summer 96 is coupled to the dc extractor 92 and the dc reference generator 94 to compare the measured and reference dc components of the compensated detection signal 46. The compensation error signal 98 generated by the comparison indicates the effectiveness of the distortion compensation. Optimal compensation will have been achieved when the compensation error signal 98 equals to zero. The measured dc component equaling or approximately equaling the reference dc component indicates that substantially the entire distortion component of the amplified detection signal 36 has been canceled. However, where the compensation error signal is greater than zero, it indicates that some part of the distortion component has not been compensated because the measured dc component of the compensated detection signal 46 is higher than expected.
As the slow tail component of curve 52 is essentially low-voltage dc, uncompensated distortion introduces additional dc and skews the measured dc component upward above expected reference levels. Likewise where the compensation error signal is less than zero, it indicates that the distortion component has been over compensated. The fact that the measured dc component of the compensated detection signal 46 is lower than expected, it can indicate that some part of the data component of the amplified detection signal 36, in addition to the distortion component, has been canceled by the feedback compensation signal 48. The sign and magnitude of the compensation error signal 98 in this way can represent the type and degree of adjustment needed to the decision feedback equalizer 39.

[0080] The filter controller 104 can be included in the control module 90 and coupled to the output of the summer 96 to receive the compensation error signal 98 as an input.
The filter controller 104 can be configured to use the compensation error signal 98 as an error signal for controlling the feedback filter 44 until optimal compensation of the amplified distortion signal 36 is achieved. Accordingly, the compensation error signal 98 can be used to adjust one or more parameters of the feedback filter 44 until the response of the feedback filter 44 matches that of the photodetector 22 (which will be indicated by a zero valued compensation error signal 98). For example, if the feedback filter 44 comprises a plurality of discrete filters (e.g., the individual filters 80,...80N
illustrated in FIG. 5), a dc gain and/or time constant of one or more of the discrete filters can be controlled according to the compensation error signal 98.

[0081] The dc gains and time constants of the individual filters 80,...80N can be pre-characterized through offline testing of the optical receiver 20 so as to match the response of the photodetector 20. For example, a very low data rate test signal can be provided to the photodetector 20. If the individual pulses in the test signal are spaced far enough apart in time, then the response of the photodetector 20 to one pulse will not interfere with subsequent pulses. The entire photodetector response can then be sampled and analyzed for its frequency content, for example using a Fourier transform or curve-fitting algorithm.
Different parts of the photodetector response curve can also be windowed so that the different parts of the overall transient response can be isolated during the analysis for computation of dc gains and time constants. Once the response of the photodetector 22 has been characterized in this way, the feedback filter 44 (or equivalently the plurality of filters 80,...80N) can then be designed to match.

[0082] However, because the response of the photodetector 20 can exhibit some dependency on different operating or environmental conditions, listed above, the pre-characterized values may not be acceptable over the entire range of operating or environmental conditions of the optical receiver 20. Accordingly, the control module 90 can initialize the individual filters 80,...8ON to their pre-characterized parameters and, as required, adjust the filter parameters during operation of the optical receiver 20 using the compensation error signal 98 in order to maintain a good match between the feedback filter 44 and the response of the photodetector 22. For this purpose, suitable gain controllers can be implemented in the filter controller 104, in some cases one for each parameter of the feedback filter 44 being controlled.

[0083] If the feedback filter 44 (or if one of the plurality of filters 80, ... 80N) is implemented using a filter configuration in which filter coefficients are provided explicitly (e.g., filters 180, 280 and 380), the filter controller 104 can comprise a processor or microcontroller configured to calculate the respective filter coefficients based on the compensation error signal 98. For example, the processor can determine and provide these filter coefficients directly using feedback control of the compensation error signal 98, but alternatively could use the compensation error signal 98 to adjust pre-stored initial filter coefficients.

[0084] If the feedback filter 44 (or if one of the plurality of filters 80,...80N) is implemented by an RC-network fabricated on a semiconductor substrate (e.g., filter 480), the filter controller 104 can further comprise a suitable actuator for controlling the frequency characteristics of the RC-network. For example, the RC-network can comprise variable, voltage-controlled resistors and capacitors. The filter controller 104 can then include a switch converter or some other controllable voltage supply, for providing control voltages to the variable resistors, and capacitors. Alternatively, the RC-network can comprise a plurality of different pre-defined resistors and capacitors arranged in a switch network.
Depending on the control signals supplied to the switch network, a different resistor-capacitor combination can be selected so as to adjust the parameters of the RC-network.
Either way, dc gain and time constant can again be controlled in order to adjust the amount of distortion compensation provided based on the compensation error signal 98.

[0085] Referring now to FIG. 12, there is illustrated a control module 190 that can be used in the signal compensation circuit 128 as an alternative to the control module 90 illustrated in FIG. 11. In the control module 190, the compensation error signal 98 is formed instead using a summer 106 coupled to each of the input and output of the signal quantizer 60 to calculate the difference between the reconstructed data signal 38 and the compensated detection signal 36. As discussed above, when the decision feedback equalizer 39 is properly matched to the photodetector 22 and providing complete distortion compensation, ideally the compensated detection signal 36 should exactly equal the reconstructed data signal 38. The difference between these two signals as a result of quantization in the non-linear element 42, therefore, can also provide a measure of how effectively the distortion component of the amplified detection signal 36 has been cancelled. Otherwise the control module 90 can function as described herein and illustrated in FIG. 11.

[0086] Referring now to FIG. 13, there is illustrated a control module 290 that can be used in the signal compensation circuit 128 as an alternative to the control modules 90 and 190 illustrated in FIGS. 11 and 12. In the control module 290, the compensation error signal 98 can be formed using feedback from an auxiliary quantizer 108, which is identical to the main quantizer 60, but is operated to have an effectively adjustable quantization threshold.
The compensated detection signal 46 is provided to the auxiliary quantizer 108 after addition of a small offset in summer 110. Shifting the compensated detection signal 46 up or down by a small amount can simulate a corresponding shift in the quantization threshold of the auxiliary quantizer 108. The quantizer controller 112 can be configured to generate the offset level provided to the summer 110, as well as the compensation error signal 98 provided to the filter controller 104, based jointly on the output signal 114 from the auxiliary quantizer 108 and the reconstructed data signal 38 generated by the main quantizer 60.
The filter controller 104 can function as described herein above.

[0087] The output of the auxiliary quantizer 108 can be a pulse train, similar but not necessarily identical to the reconstructed data signal 38 generated by the main quantizer 60, due to the effectively variable quantization threshold of the auxiliary quantizer 108. If the distortion component of the amplified detection signal 36 has been fully compensated (resulting ideally in a perfect square wave), the threshold of the auxiliary quantizer 108 can be varied over a wide range but still produce the output signal 114 to be substantially the same as the reconstructed data signal 38. If the compensated detection signal 46 is nearly an ideal square wave, the very fast transitions between the high and low voltage levels will cross different threshold values at approximately the same time, whereas that would not necessarily be the case if the compensated detection signal had a sizable uncompensated distortion component. By observing the output signal 114 generated in response to swept controllable offset in the auxiliary quantizer 108, the quantizer controller 112 can determine the range of offset levels over which the auxiliary quantizer 108 maintains the output signal 108 substantially equal to the reconstructed data signal 38. Based on the range of offset levels for which that condition holds, the quantizer controller 112 can provide the compensation error signal 98 to the filter controller 104 to reflect the effectiveness of the distortion cancellation. When the control module 290 has settled and the distortion component of the amplified detection signal 36 is fully compensated, the error compensation signal 98 can reduce to near zero.

[0088] While the above description provides examples and specific details of various embodiments, it will be appreciated that some features and/or functions of the described embodiments admit to modification without departing from the scope of the described embodiments. For example, the modulated output coupler can be suitable for operation with many different configurations of lasers. The detailed description of embodiments presented herein is intended to be illustrative of the invention, the scope of which is limited only by the language of the claims appended hereto.

Claims

1. An optical receiver comprising:

a photodetector for generating a detection signal representative of an optical data signal received at the photodetector, the detection signal having a distortion component caused by an operating characteristic of the photodetector;

an amplifier for amplifying the detection signal to generate an amplified detection signal; and a signal compensation circuit for generating a reconstructed data signal from the amplified detection signal, the signal compensation circuit comprising a decision feedback equalizer matched to the operating characteristic of the photodetector to substantially suppress the distortion component of the detection signal in the reconstructed data signal.

2. The optical receiver of claim 1, wherein the operating characteristic of the photodetector comprises a diffusion current induced in the photodetector by the optical data signal.

3. The optical receiver of claim 1, wherein the decision feedback equalizer comprises:
a summer configured to generate a compensated detection signal by subtracting a feedback compensation signal from the amplified detection signal;

a non-linear element coupled to the summer to generate the reconstructed data signal from the compensated detection signal; and at least one filter coupled between the non-linear element and the summer in a feedback compensation loop to generate the feedback compensation signal based on the reconstructed data signal, the at least one filter configured to model the operating characteristic of the photodetector so that the feedback compensation signal substantially reproduces the distortion component of the detection signal.

4. The optical receiver of claim 3, wherein the non-linear element comprises a signal quantizer.

5. The optical receiver of claim 3, wherein the non-linear element comprises a high-pass filter and a hysteretic comparator coupled to the high pass filter.

6. The optical receiver of claim 3, wherein the decision feedback equalizer comprises a plurality of filters coupled between the non-linear element and the summer in parallel in the feedback compensation loop, each filter configured to provide a respective portion of the feedback compensation signal.

7. The optical receiver of claim 6, wherein each filter is a single-pole continuous-time filter.

8. The optical receiver of claim 6, wherein the plurality of filters comprises at least one digital filter and at least one continuous-time filter, the at least one digital filter configured to compensate fast distortion components and the at least one continuous-time filter configured to compensate slow distortion components.

9. The optical receiver of claim 8, wherein each at least one continuous-time filter is a single-pole filter and the at least one digital filter comprises a higher-order finite impulse response filter.

10. The optical receiver of claim 6, wherein the decision feedback equalizer comprises between three and five filters arranged in parallel in the feedback compensation loop.

11. The optical receiver of claim 3, wherein the signal compensation circuit further comprises a control module for configuring the decision feedback equalizer to match the operating characteristic of the photodetector by adjusting at least one parameter of the decision feedback equalizer.

12. The optical receiver of claim 11, wherein the at least one parameter of the decision feedback equalizer comprises a time constant or a gain value for the at least one feedback filter.

13. The optical receiver of claim 11, wherein the control module comprises:

a dc extractor for measuring a dc component of the compensated detection signal;

a dc reference generator for generating a reference dc component of the compensated detection signal;

a summer configured to generate a compensation error signal representative of uncompensated distortion in the compensated detection signal by comparing the measured and reference dc components of the compensated detection signal; and a filter controller configured to generate control values based on the compensation error signal used to adjust the at least one parameter of the decision feedback equalizer.

14. The optical receiver of claim 13, wherein the dc reference generator comprises a peak detector for generating an envelope signal representative of a pulse height of the optical data signal, and a scaler coupled to the peak detector for scaling the amplitude signal according to a bit distribution of the optical data signal to generate the reference dc component of the compensated detection signal.

15. The optical receiver of claim 13, wherein the decision feedback equalizer comprises at least one continuous-time filter implemented by a controllable RC-network, and the filter controller is configured to apply control signals to the RC-network based on the compensation error signal used to vary effective resistance and capacitance values of the RC-network.

16. The optical receiver of claim 1, further comprising an equalizer coupled between the amplifier and the signal compensation circuit for providing high-frequency signal boosting.

17. The optical receiver of claim 1, further comprising an ac coupling circuit coupled between the photodetector and the amplifier for suppressing low frequency components of the detection signal.

18. The optical receiver of claim 1, wherein the photodetector is a spatially modulated light detector, and the optical receiver further comprises a subtractor downstream of the photodetector configured to generate the detection signal by subtracting a pair of differential detection signals generated by the spatially modulated light detector.

19. The optical receiver of claim 1, wherein the photodetector is integrated monolithically within the optical receiver on a semiconductor substrate.

20. The optical receiver of claim 19, wherein the optical receiver is implemented in CMOS, SiGe or BiCMOS.

21. The optical receiver of claim 1, wherein the optical receiver has a bandwidth of at least 5 Gbps.