CA2295435C

CA2295435C - Linear low noise phase locked loop frequency synthesizer using controlled divider pulse widths

Info

Publication number: CA2295435C
Application number: CA002295435A
Authority: CA
Inventors: Thomas Riley
Original assignee: Skyworks Solutions Inc
Current assignee: Skyworks Solutions Inc
Priority date: 2000-01-06
Filing date: 2000-01-06
Publication date: 2004-03-30
Anticipated expiration: 2020-01-06
Also published as: EP1252713A1; CA2295435A1; TW494637B; WO2001050610A1; US6404291B1; JP2003519951A

Abstract

A method and an apparatus relating to a PLL
circuit for frequency synthesizer applications. By using a composite PFD large and small phase variations between a reference signal and the divider output are compensated for. The composite phase frequency detector (PFD) has both a digital phase frequency detector (digital PFD) and an analog phase detector (analog PD) with the digital FFD compensating for large phase differences and the analog PD compensating for smaller phase differences. The PLL automatically chooses between these two components in the composite PFD by controlling the pulse width of the divider output. This is accomplished by synchronizing the dead zone of the digital PFD with the active pulse width of the divider output and by similarly synchronizing the phase detector window of the analog PD to be within both the dead zone of the digital PFD and the active pulse width of the divider output. Thus, during the active pulse of the divider output, the analog PD is operative while during the inactive pulse the digital PFD is operative. By essentially having only one PD active at any time, problems with analog/digital mixed circuits are avoided.

Description

UHIY,-UU UU~IfIU/ lI~UJ fHJIrHL~ Ot HJJVV, 1G~~U1J OLU 1JJJ f,UU4 s~9Poz I,INE~ I,QUT NQISE PHASE LOCKED LOOP FREQUENCY SYNTHESIZER
USING CONTROLLED DIVXDER PULSE WIDTHS
~~1 r1 of the InVan~ton The present invention relates to frequency synthesizers and, more Specifically to phase locked loop (PLL) frequ~ncy synthesizers which are suitable for S wireless communieatioizs.
F3r~c_-kq~~~~~nd th Inveritidn PLL frequency synthesizers are well known in the art and are extensively usdd in wireless applications. Essentially, a PLL is a circuit that uses feedback to maintain an output signal in a specific phase relationship with a stable reference signal.
Normally, a L~LL circuit contains d minimum of 4 components:
a) a phase detector which produces output voltage proportional to the phase difference between two input signals, normally the reference signal and a frequency divider output;
b) a voltage controlled oscillator (VCO) that produces an AC output signal whose frequency is proportional to the input control voltage, normally received from the loop filter;
c) a divider that produces a signal whose fz~equency is a division of the input signal frequency, normally the output of the VCO; and d) a loop filter that controls the PLL
dynamics.
l It should be noted that the phase detector output is fed into the loop filter for filtering before being transmitted to the VCV. The vG4 output is ideally the desired output from the PLL_ For frequency synthesizer applications, the VGO produces a signal with a desired frequency_ This desired frequency is generally much higher than the frequency of the stable reference signal and is ideally a multiple of that reference signal. By dividing the frequency of the output of the VCO and comparing the phase of that divided signal with the phase of the reference signal, any differences in phase, and therefore in frequency, can be compensated for by the phase detector and the VCO.
It i~ well known that PLL circuits for frequency synthesizer applications need to be low noise and must have low sensitivity to power supply variations. A frequency synthesizer having a PLL
circuit with these characteristics would produce an ~0 output which is low in both spurious tones and noise.
Currant PLL designs for frequency synthesizers which produce frequencies near cr above the GHz range often require the output of the VCs to ba synchronized to the PLL reference signal with a repeatability error of less than 1 picosecond i10-1z see) . This repeatability can be impaired in the PLL or in any of its components by intergerence, commonly referred to as cross-talk, from either nearby circuits or from the components which actually make up the PLL. The amount of cross talk that occurs in the PLL depends on the sensitivity to power supply variations and other sources of coupling from one PLL component to another. This problem can be more Severe in a highly integrated

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application specific integrated circuit (ASIC) or when the fregueney planing of these interfering signals cannot be well controlled. Failure to meet these repeatability requirements can result in spuriQUS output tones at the syn'~hesizer output. Such an Qccurrence can possibly prevent qualification of the radio to public standards.
Given the above, cross talk is not the only source of spurious tones. Different combinations of phase detectors and dividers also produce different levels of spurious output tones at multiples of the PLL
reference frequency. One method of avoiding this pitfall is the use of filters- However, such filters can be expensive. To avoid expensive filte r-ng components, it is desirable to reduce the amplitude of these tones.
One approach to solve the problem is to use edge triggered phase detectors. This avoids the problem of obtaining a 54~ duty cycle, but it also consumes more power and introduces more sensitivity to power supply variations as compared to a Single gate phase detector such as an exclusive OR.
Another area of concern for PLL circuits is its power consumption. To be suitable for wireless applications, such as in battery powered cellular phones, the PhL circuit would have to consume little power. This would extend the battery life of the wireless equipment.
A further problem with current PLL circuits is encountered when $nalog and digital circuits are used 3b together. In mixed analog and digital integrated circuits, it is common practice to have sensitive analog operations occur at different times ~rom digital operations. However, different phase detectors have

3 different phase offsets at which they locx. This phase offset can reduce the amount of "quiet time" available for sensitive analog operations, such as sampling in switched capacitor filters. As an example, a phase S offset of 96 degrees reduces the amount of quiet time by 50$
On the other hand, a phase offset of exactly zero can also cause problems. with a phase offset of zero, the divider and the source of the reference signal can experience Cross-talk or dead zone problems.
sy way of explanation, it should be noted that the dead zone refers to the attribute of a phase frequency detector (PFD) to produce an output which does not steer the output over a range of phase difference between the divider output and the reference signal. It should also be noted that other phase detectors (PD) typically do not operate linearly over a range of all possible phase errors. The region of operation where the phase detection is linear i$ often called the phase 2d detection window of the phase detector.
Returning to the question of phasa offsets, one possible solution to the problem of phase offsets is to use xbR phase detectors. These produce a two state output with substantial spurs (tones or frequency components) at twice the reference frequency_ These spurs can be removed with substantial filtering but, as noted above, at a cost of increased component count and complexity.
Another suggested solution to the above problems deal with the divider. It has been suggested that a synchronous counter can be used to perform frequency division. However, this approach has a JA1V.-Ub UU~ftlUl 11.U4 t'AaGRL k RaaUU. T~L.bIS FLU 15JJ t'.UU~
typically been rejected for applications requiring low power.
Another approach has been to use a prescaler stage and a eynchronaus counter stage. Dividers made up of several divider stages are more vulnerable to power supply and bias line interference. Generally, divider stages that consume less power are also more vulnerable to interference. This has been addressed in the past by resynchronizin~ the divider output. However, such a measure is undesirable from a power consumption point of view because the resynchronizing circuitry can consume considerable power without adding any functionality to the divider chip.
Dividers, on their own, also pose problems.
Dividere employing a single synchronous divider can sometimes produce a very narrow output pulse. This narrow pulse width contains high frequency spectral content which can interfere with sensitive analog circuits on or near the same ehlp.
The problem has traditionally been solved by either avoiding the use of fractional--N synthesizers or by pravidin$ shielding means to isolate sources of interference from sensitive circuits. Avoiding fractional-N synthesizers avoid3 the need for sensitive 23 analog components on or near the synthesizer chip.
Without these sensitive components, the potential sources of ~.nterference are avoided.
From the above, it is clear that a need exists for a PLL cireuit for frequency synthesizer applications that avoid the problems disclosed.

dHIV, 'U0 UU ~ I tiUJ 1 ! ~ UJ rHaUHL dL HaJUU, 1G~ ~ O1J FLU 1JJJ C, UU7 The present invera.tion is a method and an apparatus relating to a PLL circuit far frequency synthesizer applications- By using a composite PFD
large and small phase variations between a reference signal and the divider output are compensated for. The composite phase frequency detector (PFD) has both a digital phase frequency detector (digital PFD) and an analog phase detector (analog PD) with the digital PFD
compensating for large phase differences and the analog PD compensating for smaller phase differences. The PLL
automatically chooses between these two components in the composite PFD by controlling the pulse width of the divider output. This is accomplished by synchronizing the dead zone of the digital PFD with the active pulse width of the divider output and by similarly synchronizing the phase detector window of the analog PD
to be within both the dead zone of the digital PFD and the active pulse width of the divider output, Thus, during the active pulse of the divider output, the ~0 analog PD is operat~.ve while during the inactive pulse the digital PFD is operative. 8y essentially having only one PD active at any time, problems with analog/digital mixed circuits are avoided.) In one embodiment, the invention provides a 2S phase locked loop (PLL) frequency synthesizer comprising, a composite phase frequency detection (CPFD) having a CPFD output; a loop filter having a voltage output; a voltage controlled oscillator (Vlro) having a frequency output; and a divider having a divider output, 30 the divider output having an active pulse with a first fixed width and an inactive pulse with a second width;
wherein the GPFD is coupled to receive the divider outputs the CFA is coupled to receive a reference signal from a reference signal generator; the loop filter is coupled to receive the CPFD output; the VCO is coupled to receive the voltage output, said voltage output controllably affecting the frequency output; the divider is coupled to receive the frequency output; the loop filter generates the voltage output based on the CPFD output; the CFPD output indicates a phase difference between the divider output and the reference signal; and the frequency has a frequency that is a multiple of the frequency of the divider output, wherein the CPFD comprises: a digital PFD having a dead zone, said dead zone being a time range when the digital PFD has no . output; an analog phase detector (PD) with an analog PD
window and an analog PD output, said PD analog window being a time range when the analog PD operates linearly; and a converter coupled to receive both a digital PFD output and the analog PD output; wherein an output of the converter is the CFPD output; the digital PFD is coupled to receive the reference signal and the divider output; the analog PD is coupled to receive the reference signal and the divider output; the digital PFD output indicates a presence and a magnitude of a phase difference between the reference signal and the divider output when the digital PFD is not in the dead zone; the analog PD output is proportional to the phase difference between the reference signal and the divider output when the analog PD is within the analog PD
window; and the dead zone is within the analog PD window.
In another embodiment, there is provided a composite phase frequency detector (CPFD) for determining a phase difference between a reference signal and an input signal, the composite phase frequency detector ZO
comprising: a digital PFD having a dead zone, said dead zone being a time range when the digital PFD has no output, an analog phase detector (PD) with a PD window and a PD output, said PD window being a time range when the analog PD operates linearly; and a converter coupled to receive both a digital PFD output and the analog PD
outputs wherein an output of the converter is the CFPD
output, the digital PFD is coupled to receive the reference signal and the input signal, the analog PD is coupled to receive the reference signal and the input signal; the digital PFD output indicates a presence and a magnitude of a phase difference between the reference signal and the input signal when the digital PFD is not in the dead zone, the analog PD output is proportional to the phase difference between the reference signal and the input signal when the analog PD is within the PD
window, the dead zone is within the PD window and said digital PFD output being larger in magnitude than the ~ (a) analog PD output when the digital PFD is not in the dead zone.
In another embodiment, the invention provides an analog phase detector (PD) for detecting differences in phase between a reference signal and an input signal, the analog PD comprising, a NOT gate coupled to receive the reference signal and having an output, a first AND
gate coupled to receive the output of the NOT gate and the input signal, said first AND gate having an output, a second AND gate coupled to receive the reference signal and the input signal, said second AND gate having an output and a subtractor circuit coupled to receive the output of the first AND gate and the output of the second AND gate, said subtractor circuit having an output which is the difference between the output of the first AND gate and the output of the second AND
gate, wherein the output of the subtractor is an analog PD output.
In yet another embodiment, the invention provides a digital phase frequency detector (PFD) for detecting differences in phase and frequency between a reference signal and an input signal, the digital PFD
comprising, a phase frequency detector coupled to receive the reference signal and the input signal, the PFD producing a pump up signal and a pump down signal, said pump up signal and pump down signal being produced depending on the presence and magnitude of a phase difference between the input signal and the reference signal, a PFD NOT gate coupled to receive the input signal, a first PFD AND gate coupled to receive an output of the PFD NOT gate and the pump up signal from the PFD, a second PFD AND gate coupled to receive the output of the PFD NOT gate and the pump down signal from the PFD and a combiner coupled to receive the output of the first PFD AND gate and the output of the second PFD
AND gate, said combiner producing a digital PFD output based on the output of the first PFD AND gate and the output of the second PFD AND gate.
In yet another embodiment, the invention provides a frequency divider for dividing the frequency of an input signal, the divider having a divider output and comprising, a 2x1 multiplexer coupled to receive~:a divide ratio signal; a pulse width signal; and the divider output, said multiplexer multiplexing between the divide ratio signal and the pulse width signal to produce a multiplexer output, said multiplexing being determined by the divider output; a synchronous counter coupled to receive the multiplexer output and the input signal, said counter being constructed and arranged to conduct a count down from a value determined by the multiplexer output with the input signal providing a clock signal, said counter producing a count pulse when the count down is terminated and a T flip-flop coupled to receive the count pulse from the synchronous counter and further coupled to receive the input signal as the clock signal, the T flip-flop having an output which is the divider output.
In another embodiment, the invention provides a method of compensating for and detecting phase differences between a reference signal and a subject signal with a controllable pulse width using . an analog phase detector having a phase detector window in which the analog phase detector is active during the phase detector window, the analog phase detector receiving the reference signal and the subject signal, a digital phase frequency detector having a dead zone in which the digital phase frequency detector is inactive during the dead zone, t:he di~~it~aL ~o~ase ~:rec:y:~encl detect=or receiving the reference signal aced tine subject signal and a divider producing the subject signal with a S controllable pulse width signal, the method comprising:
controlling the control_ab:le pulse width of t=he subject signal and synchronizinc:~ the analog phase detector window and the dead zone such th:~t, r;he phase detector window and the dead zone are botrv within an active pulse of the subject signal.
In yet another embodiment, there is provided a method of compensating for and detecting phase signal differences between a refereice ;ignt~l and a subject signal with a pulse widt h, tl-~e m~,th~~7~:..~ comprising activating a digit~il phacse frec~iac:,rac~ detecl=c>r t:o provide phase frequency detect:_ic;~n betwe:ea tr~;:e re:Eerer_ce signal and the subject si:,~na'~ cxuring yin ina~;tive pulse of the subject signal, activating an analog phase detector to provide phase detection between r~:he reference signal and the subject signal during an :~cti.ve ~:aulse of the subject signal.
Brief Description of the Drawings A better understanding of tuhe invention may be obtained by reac~incf the detai led ;:iesc;ript:ion of the invention below, iru conjunction with the following drawings, in which:
Figure 1 is a schemati::~ di~~gram of a PLL
according to the ii-.vend on;
Figure 2 is a schemat:ic_: d_~agram of an analog phase detector accordinG to the ~rnver~t::ic:m;

JAN, -Ub' UU~'I'HU) ll:Ub NASCAL & ASSUG, '1'~;L~bI,S blU 1555 N. U14 Figure 2A is a collection of timing diagrams illustrating the relationships betwten the signals in the analog phase detector.
Figure 3 is a sohcmatic diagram of a digital phase/frequency detector according to the inventi.an:
Figure 4 is a schematic diagram of a divider according to the invention;
Figure 5 is a sGhdmatic diagram of an alternative divider according to the invention; and Figure 6 i$ a schematic diagram of a synchronous counter according to the invention.
e:~ r~ ~;~ti on ef the Pre rred Embodiment Referring to Figure I, the components of the PLL circuit 10 are as follows:
Voltage Controlled Output (VCO) 20:
The VCO 20 provides a frequency output (foJ
~0; the frequency of the output signal changes in response to a tuning signal (VTune) 40. Typically the ZO tuning signal is a voltage from the loop gilter 50.
Divider 50:
The divider provides a divider output signal (Dout) 70 which has a period consisting of an active pulse and an inactive time. The period of the output signal is controlled by an external signal, the divide ratio, (Ndiv) 80, and the VCO frequency output signal (fo) 30. Typically, the Divider counts oycles of the output signal (fo) 30 and provides an active pulse in which the divider output signal (bout) 7o is high for a predetermined number of cycles and err inactive time in which Daut '~o is low for another predetermined number of cycles. The sum of the active pulse and the inactive time is the period of Dout 70. The number of cycles in m the period of the divider output signal is c:ont:rolled by the Programable Divide Ratio generator 190. The Programmable Divide Rat:i_o Generator 190 may comprise a DE (delta-sigma) modulai~or to prc:~vide fractional-N
S operations. Optionally, the di~,~i.dez also :responds to a pulse width control ~;igznal, PWidt:h 9s:) such that the active pulse width of true divider is controlled by this signal. The width of the pulse during the inactive time (the inactive pulse width) can vary t.o provide the desired division ratio. This wil..l allow the generation of the desired output fuequency.
The composite PFD li)~i .s ~~::~mprisE~d of an analog PD 110, a digital PFD 1'2'0, arW a converter 130.
Analog PYnase Detector 1'10:
The analog phase detector 1.10 provides a phase error signal Ipa l~=!0, proportional. t~::~ the difference in phase between a reference signal Ref 150 and Dout 70.
Typically, the phase difference between Ref 150 and Dout 70 is defined by the differernce i~m~ime of the rising edges as in, for ex: ample., t:hEe MC1,%O~~ii type of phase detector produced ~.~y Mlot:orola. Plmse diff=erence can also be defined in terms of both r.isi.rug ancd falling edges as in an exclusive or (XOR) phase dc=te<eTor. Typically, the phase error signal is a currerut ~>rodlaced by a charge pump controlled by a MC12040 r_ypc.e o.' phase detector. It could also be a voltage produced by ~a sample and hold phase detector.
Digital Phase/ Frequenc:'y L:~etector 120 with a Dead Zone:
Thc~ digital phase/freq~.a.enc:y detector 120 provides a phase error corr_ect:ia:u s_i.c~nal ( Ipd) 160. The polarity of the phase error cc.orrc~c::t~.c~n signal Ipd 160 changes in correspondence to whe2:her a predetermined edge divider leads or lags a predetermined edge of the reference signal.
Loop Filter 5t):
The Loop F'i.iter 50 proGides a tuning signal, Vtune 40, responsive to a phase error signal Ip 70. The Loop Filter helps ~o remove high frequency components from the tuning signal and provides an open loop PLL
gain which will result in stable negative feedback when the loop is closed.
Reference Generator 180:
The reference qeneratc:~r 18() is an oscillator which provides a suff.i.ci_ently acc,urar:e reference signal 150 to the composite PFL~ 1.00, 5,;rh .~ reference generator 180 is well krsown in tr-e a~~t.
Programmable Divide Rat=io ~:)enerator 190:
The programmax>le div_~de rai::io generator 190 provides the external divider ravio :signal Ndiv 80 to the divider 60. Ti'~is device is wel L ~:r~.own in the art .
Converter 130:
The converter 130 i.:~ paArt ~:~L the composite PFD
100. The converter laCi i s co~~plF:~d tc~ .receive the signals Ipa 140 anc:I Ipd 160 from T::he di_g~_tai PFD 120 and analog PD 110. These t:~~~o signal.=,~ a_,~c=~ converted to form the signal Ip 170 which is received by the loop filter 50.
As can be seen from F.ic:)ure :,., t:he composite PFD 100 has 2 PD's: the digital fFD 1?,~ and the analog PD 110. The analog PD 11. c0 has .l~~w m~i;~e and low sensitivity to power supply var-i.ut: i.ou;s while the digital PFD 120 has a high p)uase detec~tox gaa.ru and can provide frequency steering. To prevent: ::e:~rnte>.ntion issues between these two components, trm r~I,l~ cJircuit 100 effectively switches between i::he~r;. c~:iven that the univ, -uo uupnu~ m;ua rnaun~ ~ aaau4, m~~DIJ aLU 1J~J r, um analog PD 114 is linear in characteristic only when in its phase detector window, it is desirable to use the analog PD 110 only in this time slot.
Furthermore, given that the digital PFD 120 is essentially useless while in its dead zone, it is not desirable to utilize it in this time slot. Thus, by synchronizing the analog PD 110's PD window to be within the dead zone of the digital PFD 120, we can take advantage of the benefits of both PD's.
The above can be accomplished by having the analog PD 110 active (within its PD window) during the active pulse of the divider output and by having the digital PFD 120 inactive (within its dead zone) during the same time frame (within the active pulse of the divider output). This means that, while the divider output is in its active pulse, the analog PD 110 provides the phase detection and freguency steering functions for the PLL with the digital PFD 120 effectively inactive. However, when the divider output is within its inactive pulse, the digital PFD 120 provides the phase detection and frequency steering functions for the PLI. with the analog PD effectively inactive.
The scheme outlined above can be implemented by designing the digital PFD 120 and the analog PD 110 accordingly.
Analog QD
with respect to the analog PD 110, Figure 2 illustrates one design which allows the analog PD 110 to be active within the active pulse of the divider output and Otherwise effectively inactive.
The analog PD 110 in F~-gore 2 is essentially a Balanced Tri-State phase detector which can produce a unw,-uo uu~inu~ m ua rriaun~ ~ naauu, iG~~DIJ aLU IJdJ r,Ula signal with a fixed amplitude A_ The phase error signal Ipa 140 can have three possible values: +A, 0, -A.
The phase error correction signal (Ipa) is of a predetermined amplitude during the active pulse cf the divider and of zero amplitude during the inactive time of the divider. The polarity of the phase error correction signal changes in correspondence to the state of the reference signal. If the reference signal is pos~.tive, the correction signal is positive. If the reference signal is negative, the correction signal is negative.
These three values are obtained by having the analog PD 11o internally generate two signals APU 200 and AFD 210.
I5 The digital signal Analog Fump Up APU 200 would cause the vCO to run faster as this means the frequency output 30 lags the reference signal. The digital signal (Analog Pump Dawns APh 210 would cause the vC0 to run slower as this means the frequency output 30 is leading the reference siqnal.
The signals APU 200 and APD 210 are obtained by suitably gating the reference signal R~f 150 and the divider output Dout 70. As can be see from Figure 2, the reference signal Ref 150 is fed into a NOT gate 220 and a first AND gate 230. A second AND gate 240 receives the output of the NOT gate 220 and the divider output Dout 70. Thus, the gating circuitry in Figure 2 implements the following logical equations:
3U APU = Ref AND Daut APD = Ref AND Daut JAN. -Ub' UU~'I'HU) 11:UB NASGAL & AJSUU, '1'LL~bIS bZU 155,5 i', Uly Hdwevar, to generate the phase error signal Ipa 140, a subtractor 250 is required. The subtractor 250 subtracts APD 210 from APU 200 to directly generate the phase error signal Ipa 140. If the divider output Dout 70 lags the reference signal 150, signal APU 200 would be logical LO and APD 210 would be logical HI
during the lag time, generating a phase error gignal Ipa of amplitude -A during the lag time. This is because the relationship between Ipa and the two digital signals is reprGSented by Ipa = APU-APD. Conversely, if the divider output bout 70 leads the reference signal 150, the signal APU 200 is logical HI during the lead time and logical LO otherwise. Signal APD 210 is logical HI
when the reference signal 150 and divider output Dout 70 are of opposite polarities. Because of this, the signal Ipa is of amplitude +A during the period when the signal Dout leads the reference signal. If the reference signal 150 is in phase with the divider output Dout 70, both APU 200 and APD 210 are logical HI during altexnating periods, giving a balanced phase error signal Ipa whose net effect is of no change to the divider output. To best illustrate the above relationships between the signals, Figure 2R shows the different relevant signals when the dividEr output Dout 70 leads, lags, or is in phase with the reference signal 150_ The signal Ipa can be generated by a charge pump within the subtractor 250 which produces a voltage with an amplitude suitable far the combiner 130.
~,gjtal PFD
With respect tb the digital FfD 120, it is clear from the above that the dead zone of the digital PFD 120 must be aligned with bath the active pulse of the divider output 70 and the PD window of the analog PD
is unn, uu uup uv~ ir~m rnvunu a nvvuu. ruu~UlV VLU IUUV LULU
110, This can be accomplished by implementing the circuit illustrated in Figure 3.
Figure 3 illustrates haw a well known and conventional PFD 260 can be used to extend a digital PFD's dead zone with suitable logical gating. It should be noted that the conventional PFD 260 produoes digital signals (Pump Up) PU 270 and iPump Down) PD 280 depending on whether and by how much the divider output 70 is leading or lagging the reference signal 150. The signals PU 270 and FD 280 are responsive to phase errors between the reference signal 150 and the divider output 70.
As can be seen from Figure 3, the dead zone of the digital PFD 320 ~.s extended by feeding the divider output 70 into a NoT gate 290 and deeding the qutput of the NOT gate 294 to an AND gate 300. Into this AND gate 300 is also fed th~ digital signal PU 270. A second AND
gate 310 receives the PD 280 signal and also the output of the NOT gate 290. The output of the AND gate 300 produces signal DPU 320 that is internal to the digital PFD 120. Also internal to the digital PFD is the signal DPD 330, the output of the AND gate 310.
Thus, from the logical gating in Figure 3, as long as the divider output 70 is nvt in its active pulse, either the signal DPD 330 or the signal DPU 320 is active. If the divider output 70 is within its active pulse, neither DPU 320 nor DPD 330 can have a logical HI value.
The signals PU 270 and fD 280, while digital, also provide the magnitude of the lead or lag betwe~n the reference signal 150 and the divider output Dout 70 by the length of their digital pulses.
i~

It should be noted that': a combiner 340 is also required in the digital PFD .~2u t:o produce the phase error correction sigrual Ipd 160. 'The co:mbiner 340 has a charge pump which produces the pl-aase error correction signal 160 depending on the sigroals DfiJ 320 and DPD 330.
If DPU 320 i.s logical. HI then cornbiner 340 produces a pump up signal to cause the VCO '.~~J to run faster while if the DPD 330 is logical HI then combiner 340 produces a pump down signal to clause the VCO ?0 to run slower.
From the digit:al PF'D 1 ~'u and the analog PD
110, the converter_ 13c) (fig. 1.receeves both the error correction signals i~>d a 60 arcci T:,E:>a l.~lc). Ideally, only one of these error co:rrectiori ~~i~:lraal.:~ shou_Ld be active at any one time . tfowevE~r, by set: t iry~ the error correction signal =pd 1~~0 to have: a h~_gher magnitude than Ipa 140, the ~c~ig_ital PFD L2i~ output Ipd 160 should take precedence, thereby causing the PLL to acquire lock faster. The converter 130 produces the signal Ip 170 based on whether tue sicrnals Ipd lE~~~ or Ipa 140 indicate a need to cause the VSO 20 to opt~~rat_~ faster' or slower.
The signal Ip 170 is rfece:itred by the :Loop filter SO which is also a volt~~g~.-' gc~xlerat:or. Based on the signal Ip 170 from the cornpo,~site PFD 100, tl-~e loop filter generates the proper ~~e~lt,--~~re -:i.gnal Vtune 40 too cause the VC0 20 to eitrer run f~~atc~:~ , slower o.r neither.
It should be noted than. tlE, above descriptions assume that the divider outpm:: i; a ic>gical one (or HIj during the active pulse.
Divider 60:
Figure 4 illustrate:, ara eml.>odiment of the divider 60. The divider 60 .:~ aol;~pr:.sed of a i8 dRIV, -UD UU ~ I 11UJ 1 l ~ Uy YRJLAL ~ HJ~UL, 1>;L ~ b t J bLU 1 J~J Y, ULL
multiplexes 350, a synchronous counter 360 and a T flip-flop 370.
The multiplexes 350 is a 2x1 multiplexes receiving the divide ratzo Ndiv 80 and the pulse width control signal Pwidth 9o as its inputs. The Select signal for the rnultiplexer 350 is provided by the divider output Dout '0. Thus, if the divider output Dout 70 is HI, the divide ratio Ndiv 84 is chosen, while if the divider output Oout 7o is Lo, the pulse width control signal Pwidth 90 is chosen. Whichever input is chosen, the multiplexes output 355 is transmitted to the synchronous countez~ 360_ The synchronous counter 36o is a Ioadable down counter. The counter 36o hay an input M, a clock input CLK, and an output TC. The input M is the value from which the counter counts down, with a decrement for every pulse received from the clock input CLK. The normal value for TC is LO while the counter 350 is counting down_ However, once the counter 360 has reached d, the value of the signal TC changes to HI
until a new value M is loaded.
For this application, the counter 360 receives its clock input ChK from the frequency o~xtput 30 and its initial value M from the multiplexes 350. Thus. if the pulse width control signal Pwidth 90 is loaded into the initial value M, the counter produces a HI pulse for every x pulses of the frequency output fo 30 with X
being the value of the pulse width control signal FWidth 90. Similarly, if the value of the divide ratio Ndiv 80 3D is Y, and if the divide ratio Ndiv 80 is loaded into M, the couz'iter 360 produces a pulse HI for every Y pulses of the frequency output fo 30.

JAIV. -Ub UUpItiU~ ll:lU t'AJ(iAL i~ HJaUIi. ThL:bIJ bLU lbJJ 1', U1J
The T flip-flop 3'10 receives the counter output TC of the counter 360 and a clock input GK from the frequency output fo 30. Since the T flip-flop (also known as toggle flip-flop} compliments its output Q only when a HI is detected in its input T, the state of its output Q is constant until a HI xs detected at its input T.
The workings of the above can now be explained_ Hy loading eith$r the pulse width contrpl signal PWidth 9D or the divide ratio signal Ndiv 80 into the counter 360, the appearance of a HY pulse at the oounter output Tc is controlled. So. if Ndiv = 10 and PWidth ' 15, the counter 360 produces a HI pulse every 10 or 15 pulses of the frequency output to 30, depending on which value was received by the counter 350. This HI
pulse causes the Q output of the T flip-flop 370 to change every 10 or 15 pulses of the frequency output fo 30. The Q output of the T flip-flop 370 is therefor the divider output Dout 70 with an active pulse width controlled by the pulse width control signal PWidth 90 and an inactive pulse width controlled by the divide ratio Ndiv e0.
It should be noted that while the embodiment illustrated shows a pulse width control signal PWidth 90 that is controllable, the divider 60 will be equally workable if the pulse width control signal PWidth 90 is a constant signal.
Figure 5 illustrates another embodiment of the divider 60. It is known that some synchronous counters 34 require that their inputs M only ohange at cextain times. For these counter, if the inputs change at other times, the counters do not work properly.
Normally, such counters provide an ENABLE LOAD (ENLOAD}

unn, uu uupnu~ m uu muunu a nuuuv. m u~UIV ULU lUVV
input which allows a change in the input M. The circuit in FigurE 5 provides a D flip-flop 380 to account for such counters. The D flip-flop 380 is used to regulate when the T flip-flop output Q (the same as the divider output Dout 70) is applied to the multiplexer 350_ Thusr the D flip-flop 380 applies the select signal to the multiplexer 350 when the ENLOAD signal is present.
In same cases, thane implementing this will find that it is necessary to have the T flip-flop 370 respond to the falling edge of the frequency output fo 30 while the counter 360 responds to the rising edge of the frequency output fo 30. If this is the case, it will be necessary to include a connection type 390 which would invert the frequency output fo 30 or alternatively to have the T
flip-flop 370 respond directly to falling edges instead of rising edges. The delay through the counter 360 from the frequency output fo 30 to the TC output of the counter 360 determines this decision and should be understood by those versed in th~ art.
Aside from the above differences. the embodiments of the divider 50 pictured in Figures 4 and 5 work similarly.
It should be reiterated that this divider 60 has the unique characteristic that its pulse width can ZS be controlled. The controllabiJ.ity of the pulse width allows some optimization with respect to the noise generated. When using a balanced Tri-State analog phase detector as illustrated above, the noise in the components may increase as the pulse width increases.
However, the linearity of the respon$e of the analog PD
also increases with the increased pulse width. Thus, obtaining the lowest acceptable linearity with the least z~oise requires using a certain pulse width. With regard to the controlled dead gone of the digital PFD 120 above, a narrow active pulse width is desirable in many cases . This is so that the Pl~L can have a wide bandwidth until the PLL has sett:l.ed to a small phase error. However, too narrow ~~ pu.l se .~an caus;e the dead zone to not be sufficiently dead, allowing digital feed through from the digi.tal_ PFD 1?0 to -orrupt the phase error signal.
The controllability of the pulse width thus allows the user of the j:.nvention leeway to optimize the performance of the PLL t:o hi:a or her own needs with the competing trade offs in mind.
Further ~~o 'she above, a few points should be made when implemeniv.ru:~ t:he d~..vidc=r 6~) in different PLLs.
When a Delta-Sigma moclu:';_ator i3 used in the programmable divide ratio generator :'.90 ;Fig. l~ sor the divider 60, and a balanced tri-state phase detecr_or .is used, the pulse width of the divider should be wider than the Delta-Sigma produced jit:ter on t.ne divider output Dout.
This has the advant=age that the ~~?elta-Sigma produced f fitter falls entire~l y wi than t:tue l irmar phase detector window of the balarm:eca tri.-~t.atE~ ptua><~ detector. This prevents Delta-Sigma qu~~nt:i~:at_~ora n~::~_se <~nd its non-linear intermodulat=ion c:ompone,nt.:~ f morn corrupting the base band phase error signal.
When a Delta-Sigma modulator is used in the divide ratio generator for the divider, and a Controlled Dead-Zone Digital PhaseiFrequency Detector is used, the pulse width of the divTider~ should L:.:~ wider than the Delta-Sigma produced fitter or,~ the cl~_v:i.de:r output. This has the advantage that the I:~elta-.~>~_grtua produced jitt:er falls entirely within the dead zone c>f thc~ Cont=rol.led Dead-Zone Digital Phase%Frequency Det~~>ctc~x . 'T'his prevents unm uu uup uu~ m m muunu a nuuvv, m u'U1V VLU lUUV I.ucv Delta-sigma quantization noise and its non-linear intermodulation components from corrupting the base band phase error signal, 'The divider 60 has the interesting characteristic that the T flip-flops 370, 380 should be substantially insensitive to interference from other components in the same integrated Circuit, packaging, or printed circuit board_ The advantage of this divider 60 is that since the frequency output signal fo 30, IO directly clocks the output Q of T flip-flop 370 which in turn directly controls the composite phase detector 100, the rest of the divider 60 is more insensitive to outside interference. Als4. the wider pulse width at the divider output generates less high frequency content and thereby less high frequency interference for the other sensitive circuits in and around the chip.
Referring to Figure 6, illustrated is an embodiment of the counter 360. Since the input M to the counter 360 is a digital value and since the C~.K input is provided by the frequency output fo 30, the circuit shown in Figure 6 produces the required signals fC. It should be noted that the counter 360 shown in Figure 6 requ7.res do ENABLE LpAD or ENLOAD signal as shown in Figure S.
~S A divide by 2 or divide by 3 stage divides its input signal CK to produce output signal DD whexe DO is either the input frequency divided by 2 or the input frequency divided by 3. It divides by 2 normally and when both its D23 and its ENI inputs are logically true, 3D it divides by 3. There are four of these divide by 2/3 indicated in Figure 5, 400, 410, 42D an4i 430_ The other component is a synchronous counter 440. There are many types of synchronous counters that are well established muw vv vv puy a r as a r.vvr.u ... ravuvvr auu vav vrv .vvv ~ . ~m in the prior art. The selection blacks which are unlabelled provide a means of selecting particular input bits from a larger bus. Now describing the connections in Figure 6, we have an input M, the 5 mast significant bits MSH's 450 of which are distributed amongst the various divide by 2/3 stages 400, 410, 420 and 430. The least significant bits LSH of M are applied to D23 input on the first divide by 2-3 block X100_ The second L58 znput of M is applied to D23 input of the second divide by 2-3 block 410. The third LSH of M is applied to the third divide by Z or 3 l~lpGk 420. And the fourth LSB of M is applied to the fourth divide by 2-3 151ock 430. If there are different numbers of divide by 2-3 blocks or a different bus width M. the widths of these would all change. In this particular example there are four divide by 2-3 blocks.
Regarding the synchronous counter 440, its input M2 is connected td the 5 MSB~s 450 of the M input bus. Its clock input is provided by output po and the fpurth divide by 2-3 block 430_ The output DO is also the EN load output signal indicating that it is an acceptable time to change the input M. 'Ihe terminal count of the synchronous Counter TC-sync is used as the ENI input of the foatrth divide by 2-3 block 430. This and the D23 input of the fourth divide by 2-3 black 430 is used t4 C4ntrpl when the divide by 2 or 3 block divides by 3. The other output of divide by 2/3 block 43D out or ENO which is used as the enable input lENI) of the third divide by Z-3 block 420. The clock input to the third divide by 2 or 3 block 420 DO the output from the second divide by 2-3 block 410. The DO output of 420 provides a clock input to the fourth divide by 2-3 block 430. Another output ENO of 420 is used as an JAN. -Ub' UU ('I'HU) 1 I:11 NAaGAL & A~~UU, '1'h~L: b 13 ~ZU 1553 t', U1~
enable input 8NI to the second divide by 2-3 block 410.
The second divide by 2-3 block 410 D23 has an input from the second LSB of the M input bus. The clock input is CK for block 424 is provided by the divider output DO of the first D 2-3 block 44. The output DO of the block A10 to the clock input of the third divide by 2 or 3 block 420. The other inputs for the first divide by 2-9 block 960 are the LSB of the M bus and a clock input from CLK. The D0 output of block 400 is applied as the 1.0 clock input CK of the second divide by 2-3 block 410 and its enable output ENO is the overall. counter output TC.
It should be noted that the logic q$tes illustrated in the Figures and described above can easily be implemented using components that provide functions which are logical equivalents to the logical gates. such components can be analog, digital, or a combination of both.
A person understanding thg above-described invention may now conceive of alternative designs, using the principles describr~d herein. All such designs which fall within the scope of the claims appended hereto are considered to be part of the present invention.

Claims

The embodiments of the invention in which an exclusive property or privilege is claimed are defined as follows:

1. A phase locked loop (PLL) frequency synthesizer comprising:
a composite phase frequency detection (CPFD) having a CPFD output;
a loop filter having a voltage output;
a voltage controlled oscillator (VCO) having a frequency output; and a divider having a divider output, the divider output having an active pulse with a first fixed width and and inactive pulse with a second width;
wherein the CPFD is coupled to receive the divider output;
the CPFD is coupled to receive a reference signal from a reference signal generator;
the loop filter is coupled to receive the CPFD
output;
the VCO is coupled to receive the voltage output, said voltage output controllably affecting the frequency output;
the divider is coupled to receive the frequency output;
the loop filter generates the voltage output based on the CPFD output;
the CFPD output indicated a phase difference between the divider output and the reference signal; and the frequency output has a frequency that is a multiple of the frequency of the divider output, wherein the CPFD comprises:
a digital PFD having a dead zone, said dead zone being a time range when the digital PFD has no output;
an analog phase detector (PD) with an analog PD
window and an analog PD output, said PD analog window being a time range when the analog PD operates linearly; and a converter coupled to receive both a digital PFD output and the analog PD output;
wherein an output of the converter is the CFPD output;
the digital PFD is coupled to receive the reference signal and the divider output;
the analog PD is coupled to receive the reference signal and the divider output;
the digital PFD output indicates a presence and a magnitude of a phase difference between the reference signal and the divider output when the digital PFD is not in the dead zone;
the analog PD output is proportional to the phase difference between the reference signal and the divider output when the analog PD is within the analog PD window; and the dead zone is within the analog PD
window.

2. A PLL frequency synthesizer as in claim 1 wherein the divider is coupled to receive a divide ratio signal, said divide ratio signal determining the second fixed width of the inactive pulse.

3. A PLL frequency synthesizer as in claim 1 wherein the divider is coupled to receive a pulse width signal, said pulse width signal determining the first fixed width of the active pulse.

4. A PLL frequency synthesizer as in claim 3 wherein the divider is coupled to receive a divide ratio signal, said divide ratio signal determining the second fixed width of the inactive pulse.

5. A PLL frequency synthesizer as in claim 1 wherein the dead zone and the analog PD window are within the duration of the active pulse.

6. A PLL frequency synthesizer as in claim 1 wherein the analog PD comprises:
a NOT gate coupled to receive the reference signal and having an output;
a first AND gate coupled to receive the output of the NOT gate and the divider output, said first AND
gate having an output;
a second AND gate coupled to receive the reference signal and the divider output, said second AND
gate having an output; and a subtractor circuit coupled to receive the output of the first AND gate and the output of the second AND gate, said subtractor circuit having an output which is the difference between the output of the first AND gate and the output of the second AND gate;
wherein the output of the subtractor is the analog PD output.

7. A PLL frequency synthesizer as in claim 1 wherein the digital PFD comprises:

a phase frequency detector coupled to receive the reference signal and the divider output, the PFD
producing a pump up signal and a pump down signal, said pump up signal and pump down signal being produced depending on the presence and magnitude of a phase difference between the divider output and the reference signal;
a PFD NOT gate coupled to receive the divider output;
a first PFD AND gate coupled to receive an output of the PFD NOT gate and the pump up signal from the PFD;
a second PFD and gate coupled to receive the output of the PFD NOT gate and the pump down signal from the PFD; and a combiner coupled to receive the output of the first PFD AND gate and the output of the second PFD
AND gate, said combiner producing the digital PFD output based on the output of the first PFD AND gate and the output of the second PFD AND gate.

8. A PLL frequency synthesizer as claimed in claim 1 wherein said digital PFD output is larger in magnitude than the analog PD output when the digital PFD
is not in the dead zone.

9. A PLL frequency synthesizer as claimed in claim 1 wherein the divider comprises:
a 2×1 multiplexer coupled to receive:
a divide ratio signal;
a pulse width signal; and the divider output;

said multiplexer multiplexing between the divide ratio signal and the pulse width signal to produce a multiplexer output, said multiplexing being determined by the divider output;
a synchronous counter coupled to receive the multiplexer output and the frequency output signal, said counter being constructed and arranged to conduct a count down from a value determined by the multiplexer output with the frequency output signal providing a clock signal, said counter producing a count pulse when the count down is terminated; and a T flip-flop coupled to receive the count pulse from the synchronous counter and further coupled to receive the frequency output signal as the clock signal, the T flip-flop having are output which is the divider output.

10. A PLL frequency synthesizer as claimed in claim 9 wherein the divider further includes:
a D flip-flop between the divider output and the multiplexer, said D flip-flop receiving:
the divider output as an input;
a load output of the synchronous counter as a clock signal;
and wherein the multiplexer receives the divider output as an output of the D flip-flop.

11. A PLL frequency synthesizer as in claim 6 wherein the digital PFD comprises:
a phase frequency detector coupled to receive the reference signal and the divider output, the PFD
producing a pump up signal and a pump down signal, said pump up signal and pump down signal being produced depending on the presence and magnitude of a phase difference between the divider output and the reference signal;
a PFD NOT gate coupled to receive the divider output;
a first PFD AND gate coupled to receive an output of the PFD NOT gate and the pump up signal from the PFD;
a second PFD AND gate coupled to receive the output of the PFD NOT gate and the pump down signal from the PFD; and a combiner coupled to receive the output of the first PFD AND gate and the output of the second PFD
AND gate, said combiner producing the digital PFD output based on the output of the first PFD AND gate and the output of the second PFD AND gate.

12. A PLL frequency synthesizer as in claim 11 wherein the divider is coupled to receive a divide ratio signal, said divide ratio signal determining the second width of the inactive pulse and wherein the divider is coupled to receive a pulse width signal, said pulse width signal determining the first fixed width of the active pulse.

13. A PLL frequency synthesizer as in claim 12 wherein the dead zone and the analog PD window are within the duration of the active pulse.

14. A PLL frequency synthesizer as in claim 13 wherein the divider comprises:
a 2×1 multiplexer coupled to receive:
a divide ratio signal;

a pulse width signal; and the divider output;
said multiplexer multiplexing between the divide ratio signal and the pulse width signal to produce a multiplexer output, said multiplexing being determined by the divider output;
a synchronous counter coupled to receive the multiplexer output and the frequency output signal, said counter being constructed and arranged to conduct a count down from a value determined by the multiplexer output with the frequency output signal providing a clock signal, said counter producing a count pulse when the count down is terminated; and a T flip-flop coupled to receive the count pulse from the synchronous counter and further coupled to receive the frequency output signal as the clock signal, the T flip-flop having an output which is the divider output.

15. A PLL frequency synthesizer as in claim 14 wherein the divider further includes:
a D flip-flop between the divider output and the multiplexer, said D flip-flop receiving:
the divider output as an input;
a load output of the synchronous counter as a clock signal;
and wherein the multiplexer receives the divider output as an output of the D flip-flop.

16. A composite phase frequency detector (CFPD) for determining a phase difference between a reference signal and an input signal, the composite phase frequency detector comprising:

a digital PFD having a dead zone, said dead zone being a time range when the digital PFD has no output;
an analog phase detector (PD) with a PD window and a PD output, said PD window being a time range when the analog PD operates linearly; and a converter coupled to receive both a digital PFD output and a PD output;
wherein an output of the converter is the CFPD output;
the digital PFD is coupled to receive the reference signal and the input signal;
the analog PD is coupled to receive the reference signal and the input signal;
the digital PFD output indicates a presence and a magnitude of a phase difference between the reference signal and the input signal when the digital PFD is not in the dead zone;
the analog PD output is proportional to the phase difference between the reference signal and the input signal when the analog PD is within the PD window;
the dead zone is within the PD window; and said digital PFD output being larger in magnitude than the analog PD output when the digital PFD
is not in the dead zone.

17. An analog phase detector (analog PD) for detecting differences in phase between a reference signal and an input signal, the analog PD comprising:
a NOT gate coupled to receive the reference signal and having an output;

a first AND gate coupled to receive the output of the NOT gate and the input signal, said first AND
gate having an output;
a second AND gate coupled to receive the reference signal and the input signal, said second AND
gate having an output; and a subtractor circuit coupled to receive the output of the first AND gate and the output of the second AND gate, said subtractor circuit having an output which is the difference between the output of the first AND gate and the output of the second AND gate;
wherein the output of the subtractor is an analog PD output.

18. A digital phase frequency detector (digital PFD) for detecting differences in phase and frequency between a reference signal and an input signal, the digital PFD comprising:
a phase frequency detector coupled to receive the reference signal and the input signal, the PFD
producing a pump up signal and a pump down signal, said pump up signal and pump down signal being produced depending on the presence and magnitude of a phase difference between the input signal and the reference signal;
a PFD NOT gate coupled to receive the input signal;
a first PFD AND gate coupled to receive an output of the PFD NOT gate and the pump up signal from the PFD;
a second PFD AND gate coupled to receive the output of the PFD NOT gate and the pump down signal from the PFD; and a combines coupled to receive the output of the first PFD AND gate and the output of the second PFD
AND gate, said combines producing a digital PFD output based on the output of the first PFD AND gate and the output of the second PFD AND gate.

19. A frequency divider for dividing the frequency of an input signal, the divider having a divider output and comprising:
a 2×1 multiplexer coupled to receive:
a divide ratio signal;
a pulse width signal; and the divider output;
said multiplexer multiplexing between the divide ratio signal and the pulse width signal to produce a multiplexer output, said multiplexing being determined by the divider output;
a synchronous counter coupled to receive the multiplexer output and the input signal, said counter being constructed and arranged to conduct a count down from a value determined by the multiplexer output with the input signal providing a clock signal, said counter producing a count pulse when the count down is terminated; and a T flip-flop coupled to receive the count pulse from the synchronous counter and further coupled to receive the input signal as the clock signal, the T
flip-flop having an output which is the divider output.

20. A frequency divider as claimed in claim 19 wherein the divider further includes:
a D flip-flop between the divider output and the multiplexer, said D flip-flop receiving:

the divider output as an input;
a load output of the synchronous counter as a clock signal;
and wherein the multiplexer receives the divider output as an output of the D flip-flop.

21. A method of compensating for and detecting phase differences between a reference signal and a subject signal with a controllable pulse width using:
an analog phase detector having a phase detector window in which the analog phase detector is active during the phase detector window, the analog phase detector receiving the reference signal and the subject signal;
a digital phase frequency detector having a dead zone in which the digital phase frequency detector is inactive during the dead zone, the digital phase frequency detector receiving the reference signal and the subject signal; and a divider producing the subject signal with a controllable pulse width signal, the method comprising:
controlling the controllable pulse width of the subject signal and synchronizing the phase detector window and the dead zone such that:
the phase detector window and the dead zone are both within an active pulse of the subject signal.

22. A method of compensating for and detecting phase signal differences between a reference signal and a subject signal with a pulse width, the method comprising:

activating a digital phase frequency detector to provide phase frequency detection between the reference signal and the subject signal during an inactive pulse of the subject signal;
activating an analog phase detector to provide phase detection between the reference signal and the subject signal during an active pulse of the subject signal.

23. A method as in claim 22 further including controlling the controllable pulse width of the subject signal such that during an active pulse of the subject signal:
the digital phase frequency detector is within a dead zone; and the analog phase detector is within a phase detector window.

24. A method as in claim 23 further including controlling the controllable pulse width of the subject signal such that during an inactive pulse of the subject signal:
the digital phase frequency detector is not within the dead zone; and the analog phase detector is not within the phase detector window.

25. A method as in claim 22 further including synchronizing the dead none and the phase detector window such that both the dead zone and the phase detector window occur at the same time and are both within an active pulse of the subject signal.