CA2254375A1 - Intermodulation correction for combined analog- and digital-format rf transmitters - Google Patents
Intermodulation correction for combined analog- and digital-format rf transmitters Download PDFInfo
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Abstract
Out-of-channel intermodulation distortion in a combined analog and digital television (DTV) signal is corrected using a circuit preferably operating at intermediate frequency (IF).
The circuit provides out-of DTV-channel suppression of DTV intermodulation components in a way that better protects an analog (for example, NTSC) signal in an immediately-adjacent channel at radio frequency (RF). When adjacent-channel DTV and analog television signals are amplified by a single, high-power amplifier, the intermodulation corrector circuit suppresses the out-of-DTV-channel DTV intermodulation products to below the threshold of visibility (TOV) of the analog channel viewer. In particular, an exemplary circuit has a first mixer receiving the input DTV signal at both inputs. A
second mixer receives the first mixer's output and the input DTV signal, and provides a broadband intermodulation cancellation signal. Finally, the input DTV signal is delayed in time so that it is aligned with a phase-and amplitude-adjusted cancellation signal, and then is combined with it (such as by summing) so as to provide a corrected DTV signal that provides for substantial elimination of the out-of-band intermodulation distortion at the RF output of the transmitter.
The circuit provides out-of DTV-channel suppression of DTV intermodulation components in a way that better protects an analog (for example, NTSC) signal in an immediately-adjacent channel at radio frequency (RF). When adjacent-channel DTV and analog television signals are amplified by a single, high-power amplifier, the intermodulation corrector circuit suppresses the out-of-DTV-channel DTV intermodulation products to below the threshold of visibility (TOV) of the analog channel viewer. In particular, an exemplary circuit has a first mixer receiving the input DTV signal at both inputs. A
second mixer receives the first mixer's output and the input DTV signal, and provides a broadband intermodulation cancellation signal. Finally, the input DTV signal is delayed in time so that it is aligned with a phase-and amplitude-adjusted cancellation signal, and then is combined with it (such as by summing) so as to provide a corrected DTV signal that provides for substantial elimination of the out-of-band intermodulation distortion at the RF output of the transmitter.
Description
INTERMODULATION CORRECTION FOR
COMBINED ANALOG- AND DIGITAL-FORMAT
RF TRANSMITTERS
BACKGROUND OF THE INVENTION
1. Field of the Invention The present invention relates to intermodulation correction for digital-format radio frequency (RF) transmitters. More specifically, the invention relates to intermodulation correction especially suitable for digital television (DTV) transmitters.
COMBINED ANALOG- AND DIGITAL-FORMAT
RF TRANSMITTERS
BACKGROUND OF THE INVENTION
1. Field of the Invention The present invention relates to intermodulation correction for digital-format radio frequency (RF) transmitters. More specifically, the invention relates to intermodulation correction especially suitable for digital television (DTV) transmitters.
2. Related Art It is recognized that, when linearly-amplified signals have more than one carrier signal, intermodulationproducts are produced at frequencies that are the sum and difference of the frequencies of the original Garner signals. FIG. 1 illustrates this phenomenon in the context of an analog-formattelevision signal (for example, NTSC at intermediate frequency (IF)). A channel of analog-format TV at IF occupies a 41-47 MHz band. Included in the analog-format signal are three carriers:
the aural carrier (herein designated fA 41.25 MHz), i s the color subcarrier (herein designated f~=42.171 MHz), and the visual carrier (herein designated f~ 45.75 MHz).
Second-order intermodulation components arise from the mixing of any two of these three original or "parent" components. For example, when the aural and visual carriers are mixed, second-order intermodulation components arise at f~ ~ fA MHz. Specifically:
f~ - fA = 4.5 MHz (I) f~ + fA = 87.0 MHz (II) as shown in FIG. 1. Similarly, when the color subcarrier and aural carrier are mixed, second-order intermodulation components arise at f~ ~ fA MHz. Specifically:
f~ - fA = 0.921 MHz (III) to f~ + fA = 83.421 MHz (IV) For clarity of illustration, the remaining (f~ t f~) second-order intermodulation components are not specifically shown in FIG. 1.
The second-order intermodulation components are outside of the 41-47 MHz band occupied by the analog TV signal, and thus are easily filtered. However, the same cannot be said of third-order intermodulationproducts. Third-order intermodulation products arise when second-order intermodulation products are mixed with the other "parent"
signal. For example, when the second-order intermodulation product from the color and aural carriers (at frequency f~ - fA = 0.921 MHz) is mixed with the remaining "parent" signal (the visual carrier at frequency f~ = 45.75 MHz), the following third-order intermodulationproducts are 2o produced:
f~ - (f~ - fA) = 44.829 MHz (V) f~ + (f~ - fA) = 46.671 MHz (VI) Undesirably, these third-order intermodulation products lie within the 41-47 MHz band occupied by the analog TV signal, and cannot be filtered. These intermodulation components must be removed or compensated for.
FIG. 2 illustrates a known arrangement for cancellation of the 44.829 MHz third-order intermodulationcomponent. In FIG. 2, the visual carrier at f~ = 45.75 MHz at a high level appropriate for the mixer of choice (for example, +10 dBm) and the aural carrier at fA = 41.25 and low level are input to a mixer 21. Mixer 21 provides its output signal to a 1 o band pass filter 22, which passes only the frequency-sum signal at f~ + fA
= 87.0 MHz.
The frequency-sum signal output by the band pass filter is level-adjusted by level adjuster 23 and amplified by amplifier 24 to a proper low level (for example, -10 dBm) for input to a mixer 25. As recognized by those skilled in the art, the choice of high (or low) level inputs to the mixers determines whether the mixers are (or are not) forced into switching mode.
In addition to the 87 MHz signal, mixer 25 receives the color carrier at 42.17 MHz and at a high level appropriate for the mixer of choice (for example, +10 dBm). Mixer 25 provides an intermodulationcancellation signal having a component at (f~ + f,~
- fc = 44.829 MHz. The 44.829 MHz frequency of the intermodulation cancellation signal output by mixer 25 corresponds to the frequency of the third-order intermodulation component formed in the main IF signal according to Equation V, above.
The intermodulation cancellation signal from mixer 25 differs in timing (delay), in phase, and in amplitude from the intermodulation component present in the main IF
signal. To compensate for delay in the corrector, a delay element 26 is provided at the input for the main IF signal. The delay element drives the first input of a summing circuit 29. To compensate for the differences in phase and amplitude, a phase adjuster 27 and an amplitude adjuster 28 are provided. These adjustment elements adjust the corresponding attributes of the intermodulationcancellation signal from mixer 25 to correspond to those of the delayed to intermodulation signal in the uncorrected main IF signal. This adjustment involves adjustment to the same amplitude but opposite in phase (180°).
To accomplish the correction for the 44.829 MHz intermodulation component, summing circuit 29 adds the adjusted intermodulation cancellation signal (provided by adjustment elements 27 and 28) to the uncorrected main IF signal (properly delayed by delay element 26) . As a result, summing circuit 29 outputs a sum signal whose 44.829 MHz intermodulation component has been cancelled.
To cancel the 46.671 intermodulationcomponentproducedaccordingto Equation VI, above, the positions of the color and aural carrier are reversed in the FIG. 2 circuit.
Operation of the circuit is the same in principle as that described above.
the aural carrier (herein designated fA 41.25 MHz), i s the color subcarrier (herein designated f~=42.171 MHz), and the visual carrier (herein designated f~ 45.75 MHz).
Second-order intermodulation components arise from the mixing of any two of these three original or "parent" components. For example, when the aural and visual carriers are mixed, second-order intermodulation components arise at f~ ~ fA MHz. Specifically:
f~ - fA = 4.5 MHz (I) f~ + fA = 87.0 MHz (II) as shown in FIG. 1. Similarly, when the color subcarrier and aural carrier are mixed, second-order intermodulation components arise at f~ ~ fA MHz. Specifically:
f~ - fA = 0.921 MHz (III) to f~ + fA = 83.421 MHz (IV) For clarity of illustration, the remaining (f~ t f~) second-order intermodulation components are not specifically shown in FIG. 1.
The second-order intermodulation components are outside of the 41-47 MHz band occupied by the analog TV signal, and thus are easily filtered. However, the same cannot be said of third-order intermodulationproducts. Third-order intermodulation products arise when second-order intermodulation products are mixed with the other "parent"
signal. For example, when the second-order intermodulation product from the color and aural carriers (at frequency f~ - fA = 0.921 MHz) is mixed with the remaining "parent" signal (the visual carrier at frequency f~ = 45.75 MHz), the following third-order intermodulationproducts are 2o produced:
f~ - (f~ - fA) = 44.829 MHz (V) f~ + (f~ - fA) = 46.671 MHz (VI) Undesirably, these third-order intermodulation products lie within the 41-47 MHz band occupied by the analog TV signal, and cannot be filtered. These intermodulation components must be removed or compensated for.
FIG. 2 illustrates a known arrangement for cancellation of the 44.829 MHz third-order intermodulationcomponent. In FIG. 2, the visual carrier at f~ = 45.75 MHz at a high level appropriate for the mixer of choice (for example, +10 dBm) and the aural carrier at fA = 41.25 and low level are input to a mixer 21. Mixer 21 provides its output signal to a 1 o band pass filter 22, which passes only the frequency-sum signal at f~ + fA
= 87.0 MHz.
The frequency-sum signal output by the band pass filter is level-adjusted by level adjuster 23 and amplified by amplifier 24 to a proper low level (for example, -10 dBm) for input to a mixer 25. As recognized by those skilled in the art, the choice of high (or low) level inputs to the mixers determines whether the mixers are (or are not) forced into switching mode.
In addition to the 87 MHz signal, mixer 25 receives the color carrier at 42.17 MHz and at a high level appropriate for the mixer of choice (for example, +10 dBm). Mixer 25 provides an intermodulationcancellation signal having a component at (f~ + f,~
- fc = 44.829 MHz. The 44.829 MHz frequency of the intermodulation cancellation signal output by mixer 25 corresponds to the frequency of the third-order intermodulation component formed in the main IF signal according to Equation V, above.
The intermodulation cancellation signal from mixer 25 differs in timing (delay), in phase, and in amplitude from the intermodulation component present in the main IF
signal. To compensate for delay in the corrector, a delay element 26 is provided at the input for the main IF signal. The delay element drives the first input of a summing circuit 29. To compensate for the differences in phase and amplitude, a phase adjuster 27 and an amplitude adjuster 28 are provided. These adjustment elements adjust the corresponding attributes of the intermodulationcancellation signal from mixer 25 to correspond to those of the delayed to intermodulation signal in the uncorrected main IF signal. This adjustment involves adjustment to the same amplitude but opposite in phase (180°).
To accomplish the correction for the 44.829 MHz intermodulation component, summing circuit 29 adds the adjusted intermodulation cancellation signal (provided by adjustment elements 27 and 28) to the uncorrected main IF signal (properly delayed by delay element 26) . As a result, summing circuit 29 outputs a sum signal whose 44.829 MHz intermodulation component has been cancelled.
To cancel the 46.671 intermodulationcomponentproducedaccordingto Equation VI, above, the positions of the color and aural carrier are reversed in the FIG. 2 circuit.
Operation of the circuit is the same in principle as that described above.
The circuit of FIG. 2 has been useful in cancellation of intermodulation components in NTSC television signals because of the presence of discrete carrier frequencies in the original signal (the aural Garner at fA 41.25 MHz, the color subcarner at f~=42.171 MHz, and the visual carrier at f~=45.75 MHz).
Significantly,however, digital television (DTV) signals cannot be characterized by a small number of discrete-frequency component signals. Rather, DTV signals are essentially noise-like signals for which there is no identifiable set of "parent" signals from which to derive signals that can cancel intermodulationproduct signals that are present in the DTV signals. In DTV signals, there are no Garner signals or other definable signal that can l0 be fed into a mixer to generate a cancellation signal. Accordingly, when attempting to solve intermodulationproblems in DTV signals, one must adopt an approach that is distinct from that used in the NTSC intermodulation circuit of FIG. 2.
In addition to the technical desirability of removing or compensating for in-band intermodulationcomponents,the United States Federal CommunicationsCommission(FCC) has mandated out-of band signal suppression for digital television that is stricter than for analog (NTSC) television format signals. This mandate helps to ensure that television signals from adjacent channels do not interfere with each other.
The FCC Sixth Report and Order (April 1997), as modified by the Memorandum Opinion and Order on Reconsiderationof the Sixth Report and Order (M.O.&O., 47 C.F.R.
~ 73.622(h)) (February 23,1998), has mandated an emissionmask accordingto the following suppression requirements. For the 0.5 MHz immediately surrounding the channel, the suppression S (relative to the average in-band transmitted power) must be:
S z47 dB (of < 0.5 MHz) For frequencies distant from the channel by more than 0.5 MHz, the emission mask requires suppression according to the formula:
Sz 11.5(of+3.6)dB (0.5<of<6MHz) where:
S is the suppression requirement (in dB) relative to the average in-band power; and of is the distance in frequency (in MHz) from the channel edges.
1o Of course, just inside the channel band there should be zero suppression for optimum transmission of the channel of interest.
Out-of channel degradation in DTV signals occurs due to high-power amplification of the DTV signal in the DTV transmitter. A typical Class AB, high-power, linear amplifies in a DTV transmitter exhibits a regeneration of out-of channel signal components to about 37 dB below the in-channel signal power.
A band pass filter may be designed to provide about 10 dB of additional suppression immediately outside the band's edges without excessive (uncorrectable) group delay. Thus, if a single-channel (6 MHz) band pass filter were used around an isolated DTV
channel, the total expected out-of channel suppression to be expected from the transmitter with the band 2o pass filter would be 37+10=47 dB, meeting the FCC minimum requirement.
Significantly,however, digital television (DTV) signals cannot be characterized by a small number of discrete-frequency component signals. Rather, DTV signals are essentially noise-like signals for which there is no identifiable set of "parent" signals from which to derive signals that can cancel intermodulationproduct signals that are present in the DTV signals. In DTV signals, there are no Garner signals or other definable signal that can l0 be fed into a mixer to generate a cancellation signal. Accordingly, when attempting to solve intermodulationproblems in DTV signals, one must adopt an approach that is distinct from that used in the NTSC intermodulation circuit of FIG. 2.
In addition to the technical desirability of removing or compensating for in-band intermodulationcomponents,the United States Federal CommunicationsCommission(FCC) has mandated out-of band signal suppression for digital television that is stricter than for analog (NTSC) television format signals. This mandate helps to ensure that television signals from adjacent channels do not interfere with each other.
The FCC Sixth Report and Order (April 1997), as modified by the Memorandum Opinion and Order on Reconsiderationof the Sixth Report and Order (M.O.&O., 47 C.F.R.
~ 73.622(h)) (February 23,1998), has mandated an emissionmask accordingto the following suppression requirements. For the 0.5 MHz immediately surrounding the channel, the suppression S (relative to the average in-band transmitted power) must be:
S z47 dB (of < 0.5 MHz) For frequencies distant from the channel by more than 0.5 MHz, the emission mask requires suppression according to the formula:
Sz 11.5(of+3.6)dB (0.5<of<6MHz) where:
S is the suppression requirement (in dB) relative to the average in-band power; and of is the distance in frequency (in MHz) from the channel edges.
1o Of course, just inside the channel band there should be zero suppression for optimum transmission of the channel of interest.
Out-of channel degradation in DTV signals occurs due to high-power amplification of the DTV signal in the DTV transmitter. A typical Class AB, high-power, linear amplifies in a DTV transmitter exhibits a regeneration of out-of channel signal components to about 37 dB below the in-channel signal power.
A band pass filter may be designed to provide about 10 dB of additional suppression immediately outside the band's edges without excessive (uncorrectable) group delay. Thus, if a single-channel (6 MHz) band pass filter were used around an isolated DTV
channel, the total expected out-of channel suppression to be expected from the transmitter with the band 2o pass filter would be 37+10=47 dB, meeting the FCC minimum requirement.
However, the United States Federal Communications Commission (FCC) has allocated television channels so that many broadcasters have their DTV
channels assigned immediately adjacent their analog (NTSC) channels in the frequency spectrum.
In this adjacent-channel scenario, the out-of channel DTV intermodulationproducts that lie outside the DTV channel, but inside the adjacent NTSC channel, must somehow be corrected.
In a transmitter in which both DTV and analog (e.g., NTSC) television signals are amplified by a single wideband (12 MHz) amplifier, a single-channel (6 MHz) band pass filter cannot be used on the DTV channel alone, because it would adversely affect the signal in the immediately-adjacent NTSC channel. Further, a dual-channel (12 MHz) band pass 1 o filter cannot provide the desired suppression of DTV
intermodulationproducts that lie in the NTSC channel. Yet, it is desired to provide the additional 10 dB of suppression that a band pass filter would otherwise enable, to ensure that the shoulder of the DTV
signal is suppressed to below the threshold of visibility (TOV) of the analog signal viewer.
It is to fulfill this need for additional suppression of out-of channel DTV
intermodulation products that the present invention has been developed.
SUMMARY OF THE INVENTION
The invention provides an arrangement especially suitable for use in correcting out-2o of channel intermodulation distortion in a combined analog and digital television (DTV) transmitter. The invention envisions that such intermodulation correction of a DTV signal at intermediate frequency (IF) provides the desired additional out-of channel suppression in a way that better protects an adjacent analog (for example, NTSC) channel signal. The invention is particularly useful in applications in which adjacent-channel DTV
and analog television signals are amplified by a single, high-power amplifier, a scenario in which a single-channel band-pass filter cannot be used. The inventive intermodulation corrector suppresses the out-of DTV-channel DTV intermodulation products to below the threshold of visibility (TOV) of the analog channel viewer.
The arrangement has a first mixer for receiving the input DTV signal at both a first 1 o input and a second input, and for providing a first mixer output signal to a second mixer.
The second mixer also receives the input DTV signal, and provides a broadband intermodulation cancellation signal. Finally, the input DTV signal is delayed in time so that it is aligned with the cancellation signal, and then is combined with it (such as by summing) so as to provide a corrected DTV signal that provides for substantial elimination of the out-of band intermodulation distortion.
The inventionfurrherprovidesthatthe intermodulationcancellationsignal is adjusted in phase and amplitude before being combined (summed) with the input DTV
signal to more precisely eliminate the intermodulationdistortion. In a particularpreferred embodiment, the timing and amplitude of the two signals are equal, but 180° (opposite) in phase.
_g_ Other objects, features and advantages of the invention will be apparent to those skilled in the art upon reading the following detailed description with reference to the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention is better understood by reading the following Detailed Description of the Preferred Embodiments with reference to the accompanying drawing figures, in which like reference numerals refer to like elements throughout, and in which:
l0 FIG. 1 is a frequency-domaindiagram of an analog (for example, NTSC) television signal format, further illustrating the phenomena of second- and third-order intermodulation components that lie inside and outside the television channel.
FIG. 2 illustrates a known arrangement for cancellation of the 44.829 MHz third-order in-band intermodulation component of an NTSC-format television signal.
FIG. 3 illustrates a preferred embodiment of an out-of channel intermodulation correction circuit for digital-format RF transmitters according to a preferred embodiment of the present invention.
FIGS. 4A-4E show frequency spectra of signals at various points in the circuit shown in FIG. 3.
FIG. 5 illustrates a combined NTSC/DTV transmitter in which the intermodulation corrector according to FIG. 3 may be included in processor 330.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
In describing preferred embodiments of the present invention illustrated in the drawings, specific terminology is employed for the sake of clarity. However, the invention is not intended to be limited to the specific terminology so selected, and it is to be understood that each specific element includes all technical equivalents that operate in a similar manner to accomplish a similar purpose. Moreover, components and design procedures that are 1o readily known to those skilled in the art are omitted for the sake of clarity.
Briefly, FIG. 3 illustrates a preferred embodiment of an intermodulation correction circuit 100 for digital-format RF transmitters according to a preferred embodiment of the present invention. Referring to FIG. 3 , the circuit includes a first mixer 110, an amplifier 120 and a second mixer 140 that are connected in series. First mixer 110 receives a DTV IF
input signal on both its inputs, while second mixer 140 receives the DTV IF
signal at its remaining input. At the output of the second mixer is a cancellation signal adjustment circuit 150 that includes a phase adjuster 154 and an amplitude adjuster 156. A delay element 152 receives the DTV input signal and provides a delayed DTV input signal.
Finally, the circuit includes a summing circuit 190 that receives an output of the cancellation adjustment circuit 150 as well as the delayed DTV IF television signal from delay element 152.
The FIG. 3 embodiment is preferably applied to the DTV signal at intermediate frequency (IF) (41-47 MHz) before it is converted to an RF frequency (for example, in the VHF or UHF band). (See the discussion of FIG. S, element 330, that is provided below.) The present discussion refers to intermodulation correction, but it should be noted that the goal in the preferred application is to provide a corrected signal at the output of the RF
transmitter (element 1150 in FIG. 5) and not at the immediate output of the FIG. 3 circuit itself (element 330 in FIG. 5).
The correction provided at IF by the FIG. 3 circuit is embodied in the signal that is amplified and translated to RF for transmission. It is the high-power transmitted (RF) signal to that reflects the substantial elimination of the out-of channel intermodulation products.
Thus, the invention of FIG. 3 may be considered to be a pre-correction circuit in the illustrated application, which performs correction on IF DTV signals for the benefit of RF
transmitted DTV signals. The invention of FIG. 3 is especially suitable for correcting intermodulation products that lie outside the DTV channel but within an adjacent analog channel.
In operation, a digital television (DTV) signal at intermediate frequency (IF) is input to the correction circuit on a path 102. A first level adjuster 104 (such as an attenuator) provides the same DTV signal at a lower power level on path 102'. A second level adjuster 106 (such as an attenuator) provides the same DTV signal at a lower power level on 2o path 102".
An adjustable delay element 152 delays the input DTV signal and provides it on path 102 "' to a summing circuit 190. The delay element compensates for the delay that the correction process imposes on the cancellation signal 158 (described below).
For most purposes, the signals on paths 102, 102', 102" and 102"' can be considered to be substantially the same signal, because their respective frequency spectra are identical for practical purposes. Path 102' carries a low-level version of the DTV input signal to both inputs of first mixer 110. Path 102" carries a low-level version of the DTV
input signal to one input of second mixer 140. Path 102"' carries the selectively-delayed DTV
input signal to a first input of summing circuit 190, by-passing the correction circuit.
to Conventionally, mixers have had two inputs, including a LO (local oscillator) port and an RF (radio frequency) port. Conventionally, for a mixer, the LO port has accepted a high-power-level (+7 dBm, or switching-level, appropriate for the particular mixer chosen) sinusoidal signal, and the RF port that has accepted an information-bearing signal of a much lower power level.
However, in the illustrated embodiment, the inputs to the first and second mixers are all of a lower power level, typically -10 dBm, to essentially allow the mixers to perform time-domain multiplication; no sinusoidal local oscillator signal, as such, is input to them.
First mixer 110 essentially multiplies the input signal by itself in the time domain, and generates linear sum and difference frequency components in the frequency domain, 2o preserving the amplitude information of the input signals.
Because the broadband, noise-like DTV signal is input to both inputs of mixer 110, the sum and difference components at the output of mixer 110 are correlated with the original DTV signal on paths 102, 102' in both amplitude and phase. However, the components are far from each other in frequency because they are second-order intermodulation components (the result of only one mixing--see the discussion in the Background of the Invention).
These intermodulation components are similar to those that are generated by amplifiers, such as those used in DTV transmitters. Generally, when such amplifiers amplify a noise-like input signal, regenerated sideband components ("shoulders") arise at the amplifier output that surround the frequency spectrum of the signal that is input to the 1o amplifier. Such sideband components are also noise-like and are correlated with the input signal, a fact used to advantage in the present invention.
Referring again to FIG. 3, the IF output of first mixer 110 is fed through an amplifies 120, and the resulting amplified signal is fed on path 122 to the input port of second mixer 140. Second mixer 140 thus receives both the original DTV signal on path 102"
and the mixed signal on path 122. The sum and difference frequency components in the mixed signal on path 122 are correlated with the original DTV signal on paths 102, 102', 102", 102"'.
The components in the mixed signal on path 122 are similar to the sideband components that are generated by the amplifiers in DTV transmitters, and they are correlated with the original signal. The invention envisions that such components on path 122 are useful in cancelling the undesirable sideband components. Accordingly, second mixer 140 outputs an intermodulationcancellation signal that may be adjusted for use in cancelling the undesirable out-of DTV-channel intermodulation components that are generated by high-power amplifiers in DTV transmitters. The inventive technique may be thought of as achieving high-power-levelRF output correction by means of pre-correctingthe DTV signal at low power levels at IF.
The phase and amplitude of a cancellation signal on path 142 must be properly adjusted in order to achieve proper cancellation of the intermodulation components at the output of the high-power amplifier in the DTV transmitter. Adjustment circuit 150 includes 1o phase adjuster 154 and amplitude adjuster 156 to perform these adjustments.
Adjustment circuit 150 outputs a phase- and amplitude-adjusted cancellation signal on path 158 to summing circuit 190. Adjustable delay element 152 is adjusted to ensure that the delayed signal on path 102"' is adjusted to compensate for the time delay in delivering the cancellation signal on path 158 caused by the correction circuit. In the preferred embodiment, the amplitude of the two signals is adjusted to be the same, but a phase adjustment is performed at IF so that the phase difference between the signals is 180°
(opposite in phase) at the RF output of the high-power amplifier.
Summing circuit 190 receives the phase- and amplitude-adjusted cancellation signal on path 158 and adds it to the uncompensated time-adjusted DTV IF television signal on 2o path 102"'. The sum signal output on path 192 has the out-of band intermodulation pre-correctionprovided by the inventive intermodulationcancellation circuit.
In a preferred application of the invention, this intermodulation correction at IF causes intermodulation cancellation to occur at the output of the RF high-power amplifier (see FIG.
5).
To illustrate operation of this embodiment, FIGS. 4A-4E show frequency spectra of signals at various points in the circuit shown in FIG. 3.
FIG. 4A shows the frequency spectrum of an original, uncorrected intermediate-frequency (IF) DTV signal that is input to the circuit. The input DTV signal's energy is concentrated in the 41-47 MHz band, with the pilot being near the upper end of the band.
The input DTV signal with this frequency spectrum is input at low power level to both inputs 1 o of first mixer 110.
Essentially, first mixer 110 multiplies each infinitesimally narrow frequency component in the 41-47 MHz band with every other infinitesimally narrow frequency component in the same band. This time-domain multiplication causes a frequency addition that results in a spectrum that is twice as wide as the original, and whose boundaries are at twice the respective frequencies of the original boundaries. FIG. 4B shows the frequency spectrum of the resulting signal that is output by first mixer 110.
The energy of the FIG. 4B signal is concentratedbetween 82 MHz and 94 MHz, with the spectrum also reflecting the effect of the pilot carrier multiplied by itself near the upper end of the band. The frequency spectrum of the signal output by first mixer 110 is essentially retained through amplification by element 120. The FIG. 4B signal is input at low power level (such as -10 dBm) to second mixer 140.
Second mixer 140 multiplies time-domain components in the 41-47 MHz band from signal on path 102" with components in the 82-94 MHz band from the signal on path 122.
Thus time-domain multiplication causes a frequency subtraction that results in a signal with a spectrum that extends from 3 5-53 MHz, with more energy focused in the 41-47 MHz band.
This frequency spectrum is illustrated in FIG. 4C.
FIG. 4C illustrates how the portions 441, 442 (in the outlying frequency bands 35-41 MHz and 47-53 MHz, respectively) are lower than the portion in the center portion l0 443 (in the 41-47 MHz frequency band). This signal, provided on path 142 (FIG. 3), may be considered to be the "unadjusted" intermodulation cancellation signal that is input to adjustment circuit 150.
Delay circuit 152, which may simply be a suitable length of coaxial cable, delays the signal on path 102 timewise to produce signal 102"', so that portions 441, 442 (FIG. 4C) line up timewise with the corresponding "shoulders" of the uncorrected signal.
Phase adjuster 154 may simply be a variable-length cable much shorter than the length of cable forming element 152. Phase adj uster 154 shifts the phase of the cancellation signal without introducing adversely affecting time delay. This phase adjustment allows opposite phase matching of portions 441, 442 with the corresponding "shoulders" of the 2o uncorrected signal in FIG. 4A.
Finally, amplitude adjuster 156 may be a suitable variable attenuator that adjusts the amplitude of the cancellation signal that has just been correctly aligned in time and phase by elements 152 and 154. FIG. 4D illustrates the frequency spectrum of the adjusted cancellation signal output by adjustment circuit 150 on path 158 to match that of the uncorrected signal. The signal whose spectrum is shown in FIG. 4D is 180° out of phase with the signal whose spectrum is shown in FIG. 4A (the amplitude-versus-frequency diagrams cannot explicitly illustrate phase difference). In this manner, the adjusted cancellation signal that is output by amplitude adjuster 156 precisely cancels the "shoulders"
of the uncorrected signal. This cancellation is accomplished by the addition performed by to summing circuit 190.
FIG. 4E shows the frequency spectrum that results when signals on paths 102"' and 158, with respective frequency spectra in FIGS. 4A and 4D, are added. Portions 491, 492 of the spectrum of the corrected signal are shown to be at a lower level than the portions 401, 402 (shown in dotted lines in FIG. 4E) of the uncorrected signal on path 102.
Summing circuit 190 achieves intelmodulation cancellation by adding portions and 451 (representing signals that are aligned in time and amplitude but 180 ° out of phase), and likewise by adding portions 403 and 453 (likewise representing signals that are aligned in time and amplitude but 180 ° out of phase). The summation performed by summing circuit 190 does not cause a significant subtractive effect between portion 103 (FIG. 4A) and portion 453 (FIG. 4D). Thus, the desired effect of out-of channel intermodulationreduction is achieved without substantial alteration of the in-channel RF signal.
FIG. 4E shows how the FIG. 3 circuit produces an output on path 192 that cancels shoulders at the output of the high-power RF amplifier (see FIG. 5 amplifier 1145). The FIG. 3 circuit is preferably used at IF, before the corrected DTV signal is amplified by the main DTV amplifier 1145. The amplitude, phase and timing delay adjustments (in FIG. 3 elements 152, 154,156) are made with the goal of achieving cancellationat the output of the RF transmitter (FIG. 5). These adjustments may be different than the amplitude, phase and timing delay settings that would achieve cancellation at the output of the FIG. 3 circuit as such.
FIG. 5 illustrates a transmitter in which the intermodulation corrector according to the present invention may be employed. Briefly, the transmitter transmits analog (e.g., NTSC) and DTV signals using a common antenna 1150 after amplifying them with a single wideband (12 MHz) amplifier 1145. The top portion of FIG. 5 shows the NTSC
portion, and the bottom portion of FIG. 5 shows the DTV portion. Such a circuit was disclosed in co-pending U.S. Application No. 09/050,109, which is incorporated herein by reference.
Refernng to FIG. 5, a video processor 210 manipulates an input video signal to compensate, in advance, for distortion that is expected to occur at intermediate frequency (IF) and broadcast radio frequency (RF) to the modulated visual signal. Video processor 210 provides such compensation functions as differential gain compensation and differential phase compensation and luminance non-linearity compensation, to achieve linearity of response.
An NTSC modulator 220 receives the compensated video signal from the video processor 210, along with an associated audio signal. Essentially, the modulator modulates the video and audio signals onto an intermediate frequency carrier from a phase lock loop (PLL) 225 that is phase locked to common frequency reference 100. The modulator may comprise, for example, a vestigial side band (V SB) filter implemented with surface acoustic wave (SAVE technology. NTSC modulator 220 outputs an intermediate-frequency signal with NTSC-standard visual and aural carriers at 45.75 MHZ and 41.25 MHZ
carrier 1 o frequencies, respectively.
Frequency reference 100 may include, for example, an oscillator 101 that provides a stable-frequency reference signal (such as 10 MHZ) to various circuit components via a signal splitter 102. Oscillator 101 may be implemented as a temperature-controlled crystal oscillator, an oven-controlledcrystal oscillator, a GPS (global positioning system) reference signal, and the like. Signal splitter 102 may be any circuit that fans out the reference signal, ensuring a constant phase relationship throughout the circuits it drives.
The NTSC IF signal from modulator 220 is pre-distorted by an NTSC IF processor 230 to compensate for distortion that is expected to occur at radio frequency (RF) to the modulated NTSC signal. NTSC IF processor 230 may compensate for such undesirable 2o phenomena as intermodulationdistortion, cross-modulationdistortion, and incidental carrier phase modulation distortion, and the like, resulting in a compensated, purely amplitude-modulated signal that is desired. The pre-compensatedNTSC IF signal is provided to IF-to-broadcast-frequency converter 240.
IF-to-broadcast-frequency converter 240 also receives a sinusoidal carrier of a frequency determined by the desired broadcast frequency of the particular broadcast channel "N", such as in the UHF range, that is allocated to the broadcast site involved. The carrier is provided by a phase lock loop (PLL) 250, which is phase-lockedto an output from the frequency reference 100. IF-to-broadcast-frequencyconverter 240 includes a mixer that provides a low-power (for example, one watt) modulated NTSC signal at Channel N's 1 o broadcast frequency. Converter 240 reverses the frequency order of the aural and visual components of this low-power, broadcast-frequency, modulated NTSC signal, so that the visual component carrier is now below the aural component carrier, in accordance with broadcast standards.
A series of amplifiers, shown by exemplary intermediate power amplifier (IPA) and driver amplifier 270, amplify the low-power, broadcast-frequency, modulated NTSC
signal from converter 240 to a power level closer to broadcast power levels.
For example, driver amplifier 240 may output a signal of 2.5 kW peak average power at sync, with 125 W
aural power. This signal is provided to the first input of the combiner 1140.
Preferably, the signal output to the combiner is subject to automatic gain control 2o (AGC). For this purpose, one example (not shown) of an AGC feedback path is provided from the output of driver amplifier 270 to the IF-to-broadcast-frequency converter 240.
Gain control circuitry that may be of conventional design, and located within converter 240, ensures that an NTSC signal of substantially constant power level is provided to the combiner.
Like the description of the NTSC signal modulator, the following description omits conventional elements known to those skilled in the art, with the understanding that commercially-available products perform the same overall function. Further, the present description is abbreviated because the functionsperfolmed by elements 320, 325, 330, 340, 350, 360, 370, and 377 perform functions that are analogous to the functions performed by to elements 220, 225, 230, 240, 250, 260, 270, and 277, respectively.
Modulator 320 receives a Garner frequency signal that is phase locked by PLL
to a reference Garner from frequency reference element 100. Modulator 320 further encodes a 19.39 MHZ, SMPTE 310M-compliantMPEG bit stream, and modulates a pilot carrier at 46.69 MHZ in accordance with (for example) the 8-VSB standard accepted by the Federal Communications Commission for terrestrial broadcast. Modulator 320 outputs an intermediate-frequency analog signal with a pilot carrier at 46.69 MHZ, at the upper edge of the 41-47 MHZ band allocated for television signals at IF.
A DTV IF processor 330 processes the IF signal from the modulator, performing pre-compensationand pre-conditioningfunctions generally analogous to that performed by 2o processor 230 for NTSC signals. However, DTV IF processor 330 is preferably implemented as a digital signal processor (DSP) to perform the pre-compensation and pre-conditioning functions on a digital-content signal, using techniques (such as finite impulse response filters) that are better suited to processing of such signals. In any event, the DTV IF processor 330 provides a pre-compensated and pre-conditioned signal to an IF-to-broadcast-frequency converter 340.
As mentioned above, the inventive intermodulation correction device of FIG. 3 may be part of processor 330 (FIG. 5). The FIG. 3 circuit is used at IF, before the corrected DTV
signal is RF-converted, and amplified by the main DTV amplifier 1145. The amplitude, phase and timing delay adjustments (in FIG. 3 elements 152, 154, 156) are made with the 1o goal of achieving cancellationat the FIG. 5 output 1150 of the transmitter.
These amplitude phase and timing adj ustments may be different than the amplitude, phase and timing settings that would achieve cancellation at the output of processor 330 itself.
Referring again to FIG. 5, an IF-to-broadcast-frequency converter 340 converts the analog IF signal from the DTV IF processor 330 to a broadcast-frequency signal. In the 1 s preferred application of the invention, in which the DTV channel is immediately adjacent the corresponding NTSC channel in the frequency spectrum in accordance with FCC
channel assignments, two situations are encountered. The "N-1 " situation involves a DTV channel that is immediately below the NTSC channel, and the "N+1" situation involves a DTV
channel that is immediately above the NTSC channel. The IF-to-broadcast-frequency 2o converter 340 is therefore illustrated as providing a signal on Channel N-1 or Channel N+1, where "N" is the channel assigned the corresponding NTSC channel. Essentially including a mixer, converter 340 receives a sinusoidal carrier signal from a phase lock loop 350 that is driven by frequency reference element 100 to be modulated by the DTV IF
signal.
The converter's operation results in a reversal of the frequency order of the DTV
signal from the upper end of the channel (46.69 MHZ is near 47 MHZ) to the lower end of the channel at broadcast frequencies. It is noteworthy that, in the N+1 situation, this placement of the DTV signal at the lower end of the DTV channel places it only 510 kHz away from the deviated NTSC aural carrier.
Converter 340 provides a low-power, broadcast-frequency signal to a series of Io amplifiers, shown as including an intermediate power amplifier (IPA) 360 and a driver amplifier 370. Driver amplifier 370 provides to the combiner, an 8-VSB-compliant DTV
signal, in Channel N-1 for the "n-1" channel allocation situation or in Channel N+1 for the "N+1 " channel allocation situation. The signal from driver amplifier 370 is of a power level sufficient to drive the high power amplifier 1145 to provide the desired broadcast output power. An AGC feedback path (not shown) may be provided from the drivers's output back to the converter 340, which ensures that the signal provided to the combiner is of substantially constant power level.
Combiner 1140 is preferably implemented as a conventional quadrature hybrid combiner of the type discussed in detail in commonly-assigned U.S. Patent No.
4,804,931.
2o As is readily appreciated by those skilled in the art, hybrid combiners are four-port devices that have two outputs, each one of which receives half the signal power from each of the combiner's two inputs. Thus, an undesirable characteristic of hybrid combiners is that they halve the power of the sum signal. In the present use of hybrid combiners, half the power from each input signal is provided to the high-power amplifier 1145, while the other half of the power from each input signal is wasted through dissipation in a resistance to ground.
Despite the power loss to resistance, the desirable linearity of the hybrid combiner, and the isolation of the input signals from each other to thereby avoid undesirable mixing of the two inputs, make it a preferred implementation for combiner 1140.
High-power amplifier 1145 fulfills the demand of flatness of response (less than 1 to dB) across a two-channel-wide bandwidth (12 MHZ in the United States, 16 MHz in most countries outside the U.S.), and the requirement for meaningful power to the NTSC+DTV
signal s with minimal inter-channel interference. A tetrode-class device, and especially a diacrode amplifying device such as a Thomson TH-680, provide optimum performance for this application. Tetrode and diacrode implementationsare preferred because of their ability 15 to operate with cavity sections tunable to wide (two-channel wide) bandwidths, to exhibit sufficient linearity so that cross modulation and intermodulationdistortion may be corrected with established methods, and to provide meaningful broadcast power levels. Of course, the scope of the invention should not be limited to tetrode and diacrode solutions; alternative implementations, such as those involving solid state amplifiers, also lie within the 2o contemplation of the invention. The diacrode or tetrode power amplifier may be replaced by a suitable broadband solid state amplifier using an appropriate number of power RF
transistors to get to the required power, and advantageously can operate in both the UHF and VHF bands.
In an exemplary embodiment, the TH-680 can provide 104 kW of peak envelope s power, which may (as a non-limiting example) include the following allocation of power levels. To reduce interference with channels outside the adjacent-channel pair, a suitable two-channel-wide( 12 MHZ in the U. S.) band pass filter (BPF) 1146 is provided at the output to the amplifier 1145. To comply with broadcast power standards, the amplifier 1145 must amplify the combined NTSC+DTV signal so that the BPF provides a signal 25 kW
average 1 o peak-of syncpower (NTSC),1.25 kW NTSC average aural power, and 2.5 kW
average DTV
power. Of course, variation of the above particulars in accordance with commonly-known principles lies within the ability of those skilled in the art.
As is readily appreciated by those skilled in the art, such an amplifier involves a tube that performs the power amplification, as well as a resonator cavity that limits the frequency 15 range in which the tube amplifies signals. For any assigned adjacent-channel pair (either Channel N-1 through N, or Channel N through N+1 ), one skilled in the art, upon reading this specification,is readily capable, without undue experimentation,of implementing a properly-tuned amplifier using a suitable tetrode-class device and resonator cavity.
The implementationis different for each adjacent-channel pair, but the design principles remain the same regardless of the particular assignment, and further details need not be provided here to illustrate the implementation and operation of the invention.
For many applications, high-power amplifier 1145 comprises a tetrode-class device, especially a Thomson TH-680 diacrode, available from Thomson Tubes Electronique. It is to be understood that the scope of the invention should not be limited to a particular component or to a specific set of signal types.
FIG. 5 emphasizes an implementation of the power level AGC feedback arrangements that ensure that output power levels are maintained substantially constant. In FIG. 5, feedback paths 276 and 376 are shown leading from the broadcast signal output by to the band pass filter 1146, back to respective IF-to-broadcast-frequency converters 240 and 340. Paths 276 and 376 are provided in lieu of paths (now shown) from amplifiers 270, 370, respectively. NTSC channel bandpass filter 277, and DTV channel band pass filter 377, are provided in feedback paths 276, 376, respectively, so that only in-channel frequency components are returned to converters 240, 340.
The feedback arrangements operate on similar principles of feedback control, known to those skilled in the art. When average power varies from a desired steady-state power level, either at the outputs of driver amplifiers 270, 370 or at the output of BPF 1146, feedback arrangements within converters 240, 340 act to correct the variation to return the power level at the sensed point back to the desired steady-state power level.
The gain factor 2o that converters 240, 340 apply to the feedback signals on paths 275, 375 or 276, 376 are different, and are determined by the differences in magnitude of power between the outputs of driver amplifiers 270, 370 and of BPF 1146. However, the principles remain the same.
Gain correction achieved locally (within the respective NTSC and DTV paths) compensates only for variations that occur through amplification paths 260, 270 and 360, 370. However, the implementation shown in FIG. 5 achieves a more comprehensive gain correction over the entire path between the IF-to-broadcast-frequency converters 240, 340 and the ultimate output of BPF 1146.
Those skilled in the art will recognize that, although the disclosed embodiment is designed especially for eliminating intermodulation distortion from broadband, noise-like, to DTV signals, the invention is equally applicable to eliminating intermodulation distortion from analog (e.g., NTSC) signals that essentially constitute a set of carrier signals at discrete frequencies. Accordingly, an embodiment with the structure of FIG. 3 may be used to implement part of element 230 (FIG. 5) as well as element 330 (FIG. 5). Thus, it is to be understood that the invention has utility in eliminating intermodulation distortion in either or both analog (e.g., NTSC) format signals and digital format (DTV) signals.
Modifications and variations of the above-described embodiments of the present invention are possible, as appreciated by those skilled in the art in light of the above teachings. For example, the particular electrical and electronic components used, the signals on which the components operate, the power levels of the signals, the particular frequency 2o band of the signals, the ordering of elements such as the adjustment elements, and the like, may be varied in accordance with principles known to those skilled in the art without departing from the scope of the present invention. It is therefore to be understood that, within the scope of the appended claims and their equivalents, the invention may be practiced otherwise than as specifically described.
channels assigned immediately adjacent their analog (NTSC) channels in the frequency spectrum.
In this adjacent-channel scenario, the out-of channel DTV intermodulationproducts that lie outside the DTV channel, but inside the adjacent NTSC channel, must somehow be corrected.
In a transmitter in which both DTV and analog (e.g., NTSC) television signals are amplified by a single wideband (12 MHz) amplifier, a single-channel (6 MHz) band pass filter cannot be used on the DTV channel alone, because it would adversely affect the signal in the immediately-adjacent NTSC channel. Further, a dual-channel (12 MHz) band pass 1 o filter cannot provide the desired suppression of DTV
intermodulationproducts that lie in the NTSC channel. Yet, it is desired to provide the additional 10 dB of suppression that a band pass filter would otherwise enable, to ensure that the shoulder of the DTV
signal is suppressed to below the threshold of visibility (TOV) of the analog signal viewer.
It is to fulfill this need for additional suppression of out-of channel DTV
intermodulation products that the present invention has been developed.
SUMMARY OF THE INVENTION
The invention provides an arrangement especially suitable for use in correcting out-2o of channel intermodulation distortion in a combined analog and digital television (DTV) transmitter. The invention envisions that such intermodulation correction of a DTV signal at intermediate frequency (IF) provides the desired additional out-of channel suppression in a way that better protects an adjacent analog (for example, NTSC) channel signal. The invention is particularly useful in applications in which adjacent-channel DTV
and analog television signals are amplified by a single, high-power amplifier, a scenario in which a single-channel band-pass filter cannot be used. The inventive intermodulation corrector suppresses the out-of DTV-channel DTV intermodulation products to below the threshold of visibility (TOV) of the analog channel viewer.
The arrangement has a first mixer for receiving the input DTV signal at both a first 1 o input and a second input, and for providing a first mixer output signal to a second mixer.
The second mixer also receives the input DTV signal, and provides a broadband intermodulation cancellation signal. Finally, the input DTV signal is delayed in time so that it is aligned with the cancellation signal, and then is combined with it (such as by summing) so as to provide a corrected DTV signal that provides for substantial elimination of the out-of band intermodulation distortion.
The inventionfurrherprovidesthatthe intermodulationcancellationsignal is adjusted in phase and amplitude before being combined (summed) with the input DTV
signal to more precisely eliminate the intermodulationdistortion. In a particularpreferred embodiment, the timing and amplitude of the two signals are equal, but 180° (opposite) in phase.
_g_ Other objects, features and advantages of the invention will be apparent to those skilled in the art upon reading the following detailed description with reference to the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention is better understood by reading the following Detailed Description of the Preferred Embodiments with reference to the accompanying drawing figures, in which like reference numerals refer to like elements throughout, and in which:
l0 FIG. 1 is a frequency-domaindiagram of an analog (for example, NTSC) television signal format, further illustrating the phenomena of second- and third-order intermodulation components that lie inside and outside the television channel.
FIG. 2 illustrates a known arrangement for cancellation of the 44.829 MHz third-order in-band intermodulation component of an NTSC-format television signal.
FIG. 3 illustrates a preferred embodiment of an out-of channel intermodulation correction circuit for digital-format RF transmitters according to a preferred embodiment of the present invention.
FIGS. 4A-4E show frequency spectra of signals at various points in the circuit shown in FIG. 3.
FIG. 5 illustrates a combined NTSC/DTV transmitter in which the intermodulation corrector according to FIG. 3 may be included in processor 330.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
In describing preferred embodiments of the present invention illustrated in the drawings, specific terminology is employed for the sake of clarity. However, the invention is not intended to be limited to the specific terminology so selected, and it is to be understood that each specific element includes all technical equivalents that operate in a similar manner to accomplish a similar purpose. Moreover, components and design procedures that are 1o readily known to those skilled in the art are omitted for the sake of clarity.
Briefly, FIG. 3 illustrates a preferred embodiment of an intermodulation correction circuit 100 for digital-format RF transmitters according to a preferred embodiment of the present invention. Referring to FIG. 3 , the circuit includes a first mixer 110, an amplifier 120 and a second mixer 140 that are connected in series. First mixer 110 receives a DTV IF
input signal on both its inputs, while second mixer 140 receives the DTV IF
signal at its remaining input. At the output of the second mixer is a cancellation signal adjustment circuit 150 that includes a phase adjuster 154 and an amplitude adjuster 156. A delay element 152 receives the DTV input signal and provides a delayed DTV input signal.
Finally, the circuit includes a summing circuit 190 that receives an output of the cancellation adjustment circuit 150 as well as the delayed DTV IF television signal from delay element 152.
The FIG. 3 embodiment is preferably applied to the DTV signal at intermediate frequency (IF) (41-47 MHz) before it is converted to an RF frequency (for example, in the VHF or UHF band). (See the discussion of FIG. S, element 330, that is provided below.) The present discussion refers to intermodulation correction, but it should be noted that the goal in the preferred application is to provide a corrected signal at the output of the RF
transmitter (element 1150 in FIG. 5) and not at the immediate output of the FIG. 3 circuit itself (element 330 in FIG. 5).
The correction provided at IF by the FIG. 3 circuit is embodied in the signal that is amplified and translated to RF for transmission. It is the high-power transmitted (RF) signal to that reflects the substantial elimination of the out-of channel intermodulation products.
Thus, the invention of FIG. 3 may be considered to be a pre-correction circuit in the illustrated application, which performs correction on IF DTV signals for the benefit of RF
transmitted DTV signals. The invention of FIG. 3 is especially suitable for correcting intermodulation products that lie outside the DTV channel but within an adjacent analog channel.
In operation, a digital television (DTV) signal at intermediate frequency (IF) is input to the correction circuit on a path 102. A first level adjuster 104 (such as an attenuator) provides the same DTV signal at a lower power level on path 102'. A second level adjuster 106 (such as an attenuator) provides the same DTV signal at a lower power level on 2o path 102".
An adjustable delay element 152 delays the input DTV signal and provides it on path 102 "' to a summing circuit 190. The delay element compensates for the delay that the correction process imposes on the cancellation signal 158 (described below).
For most purposes, the signals on paths 102, 102', 102" and 102"' can be considered to be substantially the same signal, because their respective frequency spectra are identical for practical purposes. Path 102' carries a low-level version of the DTV input signal to both inputs of first mixer 110. Path 102" carries a low-level version of the DTV
input signal to one input of second mixer 140. Path 102"' carries the selectively-delayed DTV
input signal to a first input of summing circuit 190, by-passing the correction circuit.
to Conventionally, mixers have had two inputs, including a LO (local oscillator) port and an RF (radio frequency) port. Conventionally, for a mixer, the LO port has accepted a high-power-level (+7 dBm, or switching-level, appropriate for the particular mixer chosen) sinusoidal signal, and the RF port that has accepted an information-bearing signal of a much lower power level.
However, in the illustrated embodiment, the inputs to the first and second mixers are all of a lower power level, typically -10 dBm, to essentially allow the mixers to perform time-domain multiplication; no sinusoidal local oscillator signal, as such, is input to them.
First mixer 110 essentially multiplies the input signal by itself in the time domain, and generates linear sum and difference frequency components in the frequency domain, 2o preserving the amplitude information of the input signals.
Because the broadband, noise-like DTV signal is input to both inputs of mixer 110, the sum and difference components at the output of mixer 110 are correlated with the original DTV signal on paths 102, 102' in both amplitude and phase. However, the components are far from each other in frequency because they are second-order intermodulation components (the result of only one mixing--see the discussion in the Background of the Invention).
These intermodulation components are similar to those that are generated by amplifiers, such as those used in DTV transmitters. Generally, when such amplifiers amplify a noise-like input signal, regenerated sideband components ("shoulders") arise at the amplifier output that surround the frequency spectrum of the signal that is input to the 1o amplifier. Such sideband components are also noise-like and are correlated with the input signal, a fact used to advantage in the present invention.
Referring again to FIG. 3, the IF output of first mixer 110 is fed through an amplifies 120, and the resulting amplified signal is fed on path 122 to the input port of second mixer 140. Second mixer 140 thus receives both the original DTV signal on path 102"
and the mixed signal on path 122. The sum and difference frequency components in the mixed signal on path 122 are correlated with the original DTV signal on paths 102, 102', 102", 102"'.
The components in the mixed signal on path 122 are similar to the sideband components that are generated by the amplifiers in DTV transmitters, and they are correlated with the original signal. The invention envisions that such components on path 122 are useful in cancelling the undesirable sideband components. Accordingly, second mixer 140 outputs an intermodulationcancellation signal that may be adjusted for use in cancelling the undesirable out-of DTV-channel intermodulation components that are generated by high-power amplifiers in DTV transmitters. The inventive technique may be thought of as achieving high-power-levelRF output correction by means of pre-correctingthe DTV signal at low power levels at IF.
The phase and amplitude of a cancellation signal on path 142 must be properly adjusted in order to achieve proper cancellation of the intermodulation components at the output of the high-power amplifier in the DTV transmitter. Adjustment circuit 150 includes 1o phase adjuster 154 and amplitude adjuster 156 to perform these adjustments.
Adjustment circuit 150 outputs a phase- and amplitude-adjusted cancellation signal on path 158 to summing circuit 190. Adjustable delay element 152 is adjusted to ensure that the delayed signal on path 102"' is adjusted to compensate for the time delay in delivering the cancellation signal on path 158 caused by the correction circuit. In the preferred embodiment, the amplitude of the two signals is adjusted to be the same, but a phase adjustment is performed at IF so that the phase difference between the signals is 180°
(opposite in phase) at the RF output of the high-power amplifier.
Summing circuit 190 receives the phase- and amplitude-adjusted cancellation signal on path 158 and adds it to the uncompensated time-adjusted DTV IF television signal on 2o path 102"'. The sum signal output on path 192 has the out-of band intermodulation pre-correctionprovided by the inventive intermodulationcancellation circuit.
In a preferred application of the invention, this intermodulation correction at IF causes intermodulation cancellation to occur at the output of the RF high-power amplifier (see FIG.
5).
To illustrate operation of this embodiment, FIGS. 4A-4E show frequency spectra of signals at various points in the circuit shown in FIG. 3.
FIG. 4A shows the frequency spectrum of an original, uncorrected intermediate-frequency (IF) DTV signal that is input to the circuit. The input DTV signal's energy is concentrated in the 41-47 MHz band, with the pilot being near the upper end of the band.
The input DTV signal with this frequency spectrum is input at low power level to both inputs 1 o of first mixer 110.
Essentially, first mixer 110 multiplies each infinitesimally narrow frequency component in the 41-47 MHz band with every other infinitesimally narrow frequency component in the same band. This time-domain multiplication causes a frequency addition that results in a spectrum that is twice as wide as the original, and whose boundaries are at twice the respective frequencies of the original boundaries. FIG. 4B shows the frequency spectrum of the resulting signal that is output by first mixer 110.
The energy of the FIG. 4B signal is concentratedbetween 82 MHz and 94 MHz, with the spectrum also reflecting the effect of the pilot carrier multiplied by itself near the upper end of the band. The frequency spectrum of the signal output by first mixer 110 is essentially retained through amplification by element 120. The FIG. 4B signal is input at low power level (such as -10 dBm) to second mixer 140.
Second mixer 140 multiplies time-domain components in the 41-47 MHz band from signal on path 102" with components in the 82-94 MHz band from the signal on path 122.
Thus time-domain multiplication causes a frequency subtraction that results in a signal with a spectrum that extends from 3 5-53 MHz, with more energy focused in the 41-47 MHz band.
This frequency spectrum is illustrated in FIG. 4C.
FIG. 4C illustrates how the portions 441, 442 (in the outlying frequency bands 35-41 MHz and 47-53 MHz, respectively) are lower than the portion in the center portion l0 443 (in the 41-47 MHz frequency band). This signal, provided on path 142 (FIG. 3), may be considered to be the "unadjusted" intermodulation cancellation signal that is input to adjustment circuit 150.
Delay circuit 152, which may simply be a suitable length of coaxial cable, delays the signal on path 102 timewise to produce signal 102"', so that portions 441, 442 (FIG. 4C) line up timewise with the corresponding "shoulders" of the uncorrected signal.
Phase adjuster 154 may simply be a variable-length cable much shorter than the length of cable forming element 152. Phase adj uster 154 shifts the phase of the cancellation signal without introducing adversely affecting time delay. This phase adjustment allows opposite phase matching of portions 441, 442 with the corresponding "shoulders" of the 2o uncorrected signal in FIG. 4A.
Finally, amplitude adjuster 156 may be a suitable variable attenuator that adjusts the amplitude of the cancellation signal that has just been correctly aligned in time and phase by elements 152 and 154. FIG. 4D illustrates the frequency spectrum of the adjusted cancellation signal output by adjustment circuit 150 on path 158 to match that of the uncorrected signal. The signal whose spectrum is shown in FIG. 4D is 180° out of phase with the signal whose spectrum is shown in FIG. 4A (the amplitude-versus-frequency diagrams cannot explicitly illustrate phase difference). In this manner, the adjusted cancellation signal that is output by amplitude adjuster 156 precisely cancels the "shoulders"
of the uncorrected signal. This cancellation is accomplished by the addition performed by to summing circuit 190.
FIG. 4E shows the frequency spectrum that results when signals on paths 102"' and 158, with respective frequency spectra in FIGS. 4A and 4D, are added. Portions 491, 492 of the spectrum of the corrected signal are shown to be at a lower level than the portions 401, 402 (shown in dotted lines in FIG. 4E) of the uncorrected signal on path 102.
Summing circuit 190 achieves intelmodulation cancellation by adding portions and 451 (representing signals that are aligned in time and amplitude but 180 ° out of phase), and likewise by adding portions 403 and 453 (likewise representing signals that are aligned in time and amplitude but 180 ° out of phase). The summation performed by summing circuit 190 does not cause a significant subtractive effect between portion 103 (FIG. 4A) and portion 453 (FIG. 4D). Thus, the desired effect of out-of channel intermodulationreduction is achieved without substantial alteration of the in-channel RF signal.
FIG. 4E shows how the FIG. 3 circuit produces an output on path 192 that cancels shoulders at the output of the high-power RF amplifier (see FIG. 5 amplifier 1145). The FIG. 3 circuit is preferably used at IF, before the corrected DTV signal is amplified by the main DTV amplifier 1145. The amplitude, phase and timing delay adjustments (in FIG. 3 elements 152, 154,156) are made with the goal of achieving cancellationat the output of the RF transmitter (FIG. 5). These adjustments may be different than the amplitude, phase and timing delay settings that would achieve cancellation at the output of the FIG. 3 circuit as such.
FIG. 5 illustrates a transmitter in which the intermodulation corrector according to the present invention may be employed. Briefly, the transmitter transmits analog (e.g., NTSC) and DTV signals using a common antenna 1150 after amplifying them with a single wideband (12 MHz) amplifier 1145. The top portion of FIG. 5 shows the NTSC
portion, and the bottom portion of FIG. 5 shows the DTV portion. Such a circuit was disclosed in co-pending U.S. Application No. 09/050,109, which is incorporated herein by reference.
Refernng to FIG. 5, a video processor 210 manipulates an input video signal to compensate, in advance, for distortion that is expected to occur at intermediate frequency (IF) and broadcast radio frequency (RF) to the modulated visual signal. Video processor 210 provides such compensation functions as differential gain compensation and differential phase compensation and luminance non-linearity compensation, to achieve linearity of response.
An NTSC modulator 220 receives the compensated video signal from the video processor 210, along with an associated audio signal. Essentially, the modulator modulates the video and audio signals onto an intermediate frequency carrier from a phase lock loop (PLL) 225 that is phase locked to common frequency reference 100. The modulator may comprise, for example, a vestigial side band (V SB) filter implemented with surface acoustic wave (SAVE technology. NTSC modulator 220 outputs an intermediate-frequency signal with NTSC-standard visual and aural carriers at 45.75 MHZ and 41.25 MHZ
carrier 1 o frequencies, respectively.
Frequency reference 100 may include, for example, an oscillator 101 that provides a stable-frequency reference signal (such as 10 MHZ) to various circuit components via a signal splitter 102. Oscillator 101 may be implemented as a temperature-controlled crystal oscillator, an oven-controlledcrystal oscillator, a GPS (global positioning system) reference signal, and the like. Signal splitter 102 may be any circuit that fans out the reference signal, ensuring a constant phase relationship throughout the circuits it drives.
The NTSC IF signal from modulator 220 is pre-distorted by an NTSC IF processor 230 to compensate for distortion that is expected to occur at radio frequency (RF) to the modulated NTSC signal. NTSC IF processor 230 may compensate for such undesirable 2o phenomena as intermodulationdistortion, cross-modulationdistortion, and incidental carrier phase modulation distortion, and the like, resulting in a compensated, purely amplitude-modulated signal that is desired. The pre-compensatedNTSC IF signal is provided to IF-to-broadcast-frequency converter 240.
IF-to-broadcast-frequency converter 240 also receives a sinusoidal carrier of a frequency determined by the desired broadcast frequency of the particular broadcast channel "N", such as in the UHF range, that is allocated to the broadcast site involved. The carrier is provided by a phase lock loop (PLL) 250, which is phase-lockedto an output from the frequency reference 100. IF-to-broadcast-frequencyconverter 240 includes a mixer that provides a low-power (for example, one watt) modulated NTSC signal at Channel N's 1 o broadcast frequency. Converter 240 reverses the frequency order of the aural and visual components of this low-power, broadcast-frequency, modulated NTSC signal, so that the visual component carrier is now below the aural component carrier, in accordance with broadcast standards.
A series of amplifiers, shown by exemplary intermediate power amplifier (IPA) and driver amplifier 270, amplify the low-power, broadcast-frequency, modulated NTSC
signal from converter 240 to a power level closer to broadcast power levels.
For example, driver amplifier 240 may output a signal of 2.5 kW peak average power at sync, with 125 W
aural power. This signal is provided to the first input of the combiner 1140.
Preferably, the signal output to the combiner is subject to automatic gain control 2o (AGC). For this purpose, one example (not shown) of an AGC feedback path is provided from the output of driver amplifier 270 to the IF-to-broadcast-frequency converter 240.
Gain control circuitry that may be of conventional design, and located within converter 240, ensures that an NTSC signal of substantially constant power level is provided to the combiner.
Like the description of the NTSC signal modulator, the following description omits conventional elements known to those skilled in the art, with the understanding that commercially-available products perform the same overall function. Further, the present description is abbreviated because the functionsperfolmed by elements 320, 325, 330, 340, 350, 360, 370, and 377 perform functions that are analogous to the functions performed by to elements 220, 225, 230, 240, 250, 260, 270, and 277, respectively.
Modulator 320 receives a Garner frequency signal that is phase locked by PLL
to a reference Garner from frequency reference element 100. Modulator 320 further encodes a 19.39 MHZ, SMPTE 310M-compliantMPEG bit stream, and modulates a pilot carrier at 46.69 MHZ in accordance with (for example) the 8-VSB standard accepted by the Federal Communications Commission for terrestrial broadcast. Modulator 320 outputs an intermediate-frequency analog signal with a pilot carrier at 46.69 MHZ, at the upper edge of the 41-47 MHZ band allocated for television signals at IF.
A DTV IF processor 330 processes the IF signal from the modulator, performing pre-compensationand pre-conditioningfunctions generally analogous to that performed by 2o processor 230 for NTSC signals. However, DTV IF processor 330 is preferably implemented as a digital signal processor (DSP) to perform the pre-compensation and pre-conditioning functions on a digital-content signal, using techniques (such as finite impulse response filters) that are better suited to processing of such signals. In any event, the DTV IF processor 330 provides a pre-compensated and pre-conditioned signal to an IF-to-broadcast-frequency converter 340.
As mentioned above, the inventive intermodulation correction device of FIG. 3 may be part of processor 330 (FIG. 5). The FIG. 3 circuit is used at IF, before the corrected DTV
signal is RF-converted, and amplified by the main DTV amplifier 1145. The amplitude, phase and timing delay adjustments (in FIG. 3 elements 152, 154, 156) are made with the 1o goal of achieving cancellationat the FIG. 5 output 1150 of the transmitter.
These amplitude phase and timing adj ustments may be different than the amplitude, phase and timing settings that would achieve cancellation at the output of processor 330 itself.
Referring again to FIG. 5, an IF-to-broadcast-frequency converter 340 converts the analog IF signal from the DTV IF processor 330 to a broadcast-frequency signal. In the 1 s preferred application of the invention, in which the DTV channel is immediately adjacent the corresponding NTSC channel in the frequency spectrum in accordance with FCC
channel assignments, two situations are encountered. The "N-1 " situation involves a DTV channel that is immediately below the NTSC channel, and the "N+1" situation involves a DTV
channel that is immediately above the NTSC channel. The IF-to-broadcast-frequency 2o converter 340 is therefore illustrated as providing a signal on Channel N-1 or Channel N+1, where "N" is the channel assigned the corresponding NTSC channel. Essentially including a mixer, converter 340 receives a sinusoidal carrier signal from a phase lock loop 350 that is driven by frequency reference element 100 to be modulated by the DTV IF
signal.
The converter's operation results in a reversal of the frequency order of the DTV
signal from the upper end of the channel (46.69 MHZ is near 47 MHZ) to the lower end of the channel at broadcast frequencies. It is noteworthy that, in the N+1 situation, this placement of the DTV signal at the lower end of the DTV channel places it only 510 kHz away from the deviated NTSC aural carrier.
Converter 340 provides a low-power, broadcast-frequency signal to a series of Io amplifiers, shown as including an intermediate power amplifier (IPA) 360 and a driver amplifier 370. Driver amplifier 370 provides to the combiner, an 8-VSB-compliant DTV
signal, in Channel N-1 for the "n-1" channel allocation situation or in Channel N+1 for the "N+1 " channel allocation situation. The signal from driver amplifier 370 is of a power level sufficient to drive the high power amplifier 1145 to provide the desired broadcast output power. An AGC feedback path (not shown) may be provided from the drivers's output back to the converter 340, which ensures that the signal provided to the combiner is of substantially constant power level.
Combiner 1140 is preferably implemented as a conventional quadrature hybrid combiner of the type discussed in detail in commonly-assigned U.S. Patent No.
4,804,931.
2o As is readily appreciated by those skilled in the art, hybrid combiners are four-port devices that have two outputs, each one of which receives half the signal power from each of the combiner's two inputs. Thus, an undesirable characteristic of hybrid combiners is that they halve the power of the sum signal. In the present use of hybrid combiners, half the power from each input signal is provided to the high-power amplifier 1145, while the other half of the power from each input signal is wasted through dissipation in a resistance to ground.
Despite the power loss to resistance, the desirable linearity of the hybrid combiner, and the isolation of the input signals from each other to thereby avoid undesirable mixing of the two inputs, make it a preferred implementation for combiner 1140.
High-power amplifier 1145 fulfills the demand of flatness of response (less than 1 to dB) across a two-channel-wide bandwidth (12 MHZ in the United States, 16 MHz in most countries outside the U.S.), and the requirement for meaningful power to the NTSC+DTV
signal s with minimal inter-channel interference. A tetrode-class device, and especially a diacrode amplifying device such as a Thomson TH-680, provide optimum performance for this application. Tetrode and diacrode implementationsare preferred because of their ability 15 to operate with cavity sections tunable to wide (two-channel wide) bandwidths, to exhibit sufficient linearity so that cross modulation and intermodulationdistortion may be corrected with established methods, and to provide meaningful broadcast power levels. Of course, the scope of the invention should not be limited to tetrode and diacrode solutions; alternative implementations, such as those involving solid state amplifiers, also lie within the 2o contemplation of the invention. The diacrode or tetrode power amplifier may be replaced by a suitable broadband solid state amplifier using an appropriate number of power RF
transistors to get to the required power, and advantageously can operate in both the UHF and VHF bands.
In an exemplary embodiment, the TH-680 can provide 104 kW of peak envelope s power, which may (as a non-limiting example) include the following allocation of power levels. To reduce interference with channels outside the adjacent-channel pair, a suitable two-channel-wide( 12 MHZ in the U. S.) band pass filter (BPF) 1146 is provided at the output to the amplifier 1145. To comply with broadcast power standards, the amplifier 1145 must amplify the combined NTSC+DTV signal so that the BPF provides a signal 25 kW
average 1 o peak-of syncpower (NTSC),1.25 kW NTSC average aural power, and 2.5 kW
average DTV
power. Of course, variation of the above particulars in accordance with commonly-known principles lies within the ability of those skilled in the art.
As is readily appreciated by those skilled in the art, such an amplifier involves a tube that performs the power amplification, as well as a resonator cavity that limits the frequency 15 range in which the tube amplifies signals. For any assigned adjacent-channel pair (either Channel N-1 through N, or Channel N through N+1 ), one skilled in the art, upon reading this specification,is readily capable, without undue experimentation,of implementing a properly-tuned amplifier using a suitable tetrode-class device and resonator cavity.
The implementationis different for each adjacent-channel pair, but the design principles remain the same regardless of the particular assignment, and further details need not be provided here to illustrate the implementation and operation of the invention.
For many applications, high-power amplifier 1145 comprises a tetrode-class device, especially a Thomson TH-680 diacrode, available from Thomson Tubes Electronique. It is to be understood that the scope of the invention should not be limited to a particular component or to a specific set of signal types.
FIG. 5 emphasizes an implementation of the power level AGC feedback arrangements that ensure that output power levels are maintained substantially constant. In FIG. 5, feedback paths 276 and 376 are shown leading from the broadcast signal output by to the band pass filter 1146, back to respective IF-to-broadcast-frequency converters 240 and 340. Paths 276 and 376 are provided in lieu of paths (now shown) from amplifiers 270, 370, respectively. NTSC channel bandpass filter 277, and DTV channel band pass filter 377, are provided in feedback paths 276, 376, respectively, so that only in-channel frequency components are returned to converters 240, 340.
The feedback arrangements operate on similar principles of feedback control, known to those skilled in the art. When average power varies from a desired steady-state power level, either at the outputs of driver amplifiers 270, 370 or at the output of BPF 1146, feedback arrangements within converters 240, 340 act to correct the variation to return the power level at the sensed point back to the desired steady-state power level.
The gain factor 2o that converters 240, 340 apply to the feedback signals on paths 275, 375 or 276, 376 are different, and are determined by the differences in magnitude of power between the outputs of driver amplifiers 270, 370 and of BPF 1146. However, the principles remain the same.
Gain correction achieved locally (within the respective NTSC and DTV paths) compensates only for variations that occur through amplification paths 260, 270 and 360, 370. However, the implementation shown in FIG. 5 achieves a more comprehensive gain correction over the entire path between the IF-to-broadcast-frequency converters 240, 340 and the ultimate output of BPF 1146.
Those skilled in the art will recognize that, although the disclosed embodiment is designed especially for eliminating intermodulation distortion from broadband, noise-like, to DTV signals, the invention is equally applicable to eliminating intermodulation distortion from analog (e.g., NTSC) signals that essentially constitute a set of carrier signals at discrete frequencies. Accordingly, an embodiment with the structure of FIG. 3 may be used to implement part of element 230 (FIG. 5) as well as element 330 (FIG. 5). Thus, it is to be understood that the invention has utility in eliminating intermodulation distortion in either or both analog (e.g., NTSC) format signals and digital format (DTV) signals.
Modifications and variations of the above-described embodiments of the present invention are possible, as appreciated by those skilled in the art in light of the above teachings. For example, the particular electrical and electronic components used, the signals on which the components operate, the power levels of the signals, the particular frequency 2o band of the signals, the ordering of elements such as the adjustment elements, and the like, may be varied in accordance with principles known to those skilled in the art without departing from the scope of the present invention. It is therefore to be understood that, within the scope of the appended claims and their equivalents, the invention may be practiced otherwise than as specifically described.
Claims (30)
1. An arrangement for correcting out-of-channel intermodulation distortion in an input digital television (DTV) signal, the arrangement comprising:
a first mixer for receiving the input DTV signal at both a first mixer input and a second mixer input, and for providing a first mixer output signal;
a second mixer for receiving the input DTV signal and the first mixer output signal, and for providing a broadband intermodulation cancellation signal; and means for combining the input DTV signal and the broadband intermodulation cancellation signal to output a corrected DTV signal that provides for substantial elimination of the out-of-channel intermodulation distortion.
a first mixer for receiving the input DTV signal at both a first mixer input and a second mixer input, and for providing a first mixer output signal;
a second mixer for receiving the input DTV signal and the first mixer output signal, and for providing a broadband intermodulation cancellation signal; and means for combining the input DTV signal and the broadband intermodulation cancellation signal to output a corrected DTV signal that provides for substantial elimination of the out-of-channel intermodulation distortion.
2. The arrangement of claim 1, further comprising:
a delay circuit that adjusts relative timing of the input DTV signal and the broadband intermodulation cancellation signal.
a delay circuit that adjusts relative timing of the input DTV signal and the broadband intermodulation cancellation signal.
3. The arrangement of claim 1, further comprising:
an adjustment circuit disposed between the second mixer and the combining means.
an adjustment circuit disposed between the second mixer and the combining means.
4. The arrangement of claim 3, wherein the adjustment circuit includes:
a phase adjuster circuit that adjusts the intermodulation component phase of the signal output by the second mixer to correspond to the intermodulation component phase of the input DTV signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
a phase adjuster circuit that adjusts the intermodulation component phase of the signal output by the second mixer to correspond to the intermodulation component phase of the input DTV signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
5. The arrangement of claim 3, wherein the adjustment circuit includes:
an amplitude adjustment circuit that adjusts the intermodulation component amplitude of the signal output by the second mixer to correspond to the intermodulation component amplitude of input DTV signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
an amplitude adjustment circuit that adjusts the intermodulation component amplitude of the signal output by the second mixer to correspond to the intermodulation component amplitude of input DTV signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
6. The arrangement of claim 1, wherein:
the inputs received by the first and second mixers are at a power level low enough not to force the mixers into switching mode.
the inputs received by the first and second mixers are at a power level low enough not to force the mixers into switching mode.
7. The arrangement of claim 1, wherein the combining means includes:
a summer that adds the input DTV signal to the broadband intermodulation cancellation signal.
a summer that adds the input DTV signal to the broadband intermodulation cancellation signal.
8. The arrangement of claim 1, wherein:
the input DTV signal is at intermediate frequency (IF); and the combining means constitutes means for substantially eliminating the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
the input DTV signal is at intermediate frequency (IF); and the combining means constitutes means for substantially eliminating the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
9. An arrangement for correcting out-of-channel intermodulation distortion in an input signal, the arrangement comprising:
a first mixer for receiving the input signal at both a first mixer input and a second mixer input, and for providing a first mixer output signal;
a second mixer for receiving the input signal and the first mixer output signal, and for providing an intermodulation cancellation signal; and means for combining the input signal and the intermodulation cancellation signal to output a corrected signal that provides for substantial elimination of the out-of-channel intermodulation distortion.
a first mixer for receiving the input signal at both a first mixer input and a second mixer input, and for providing a first mixer output signal;
a second mixer for receiving the input signal and the first mixer output signal, and for providing an intermodulation cancellation signal; and means for combining the input signal and the intermodulation cancellation signal to output a corrected signal that provides for substantial elimination of the out-of-channel intermodulation distortion.
10. The arrangement of claim 9, further comprising:
a delay circuit that adjusts relative timing of the input signal and the intermodulation cancellation signal.
a delay circuit that adjusts relative timing of the input signal and the intermodulation cancellation signal.
11. The arrangement of claim 9, further comprising:
an adjustment circuit disposed between the second mixer and the combining means.
an adjustment circuit disposed between the second mixer and the combining means.
12. The arrangement of claim 11, wherein the adjustment circuit includes:
a phase adjuster circuit that adjusts the intermodulation component phase of the signal output by the second mixer to correspond to the intermodulation component phase of the input signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
a phase adjuster circuit that adjusts the intermodulation component phase of the signal output by the second mixer to correspond to the intermodulation component phase of the input signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
13. The arrangement of claim 11, wherein the adjustment circuit includes:
an amplitude adjustment circuit that adjusts the intermodulation component amplitude of the signal output by the second mixer to correspond to the intermodulation component amplitude of input DTV signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
an amplitude adjustment circuit that adjusts the intermodulation component amplitude of the signal output by the second mixer to correspond to the intermodulation component amplitude of input DTV signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
14. The arrangement of claim 9, wherein:
the inputs received by the first and second mixers are at a power level low enough not to force the mixers into switching mode.
the inputs received by the first and second mixers are at a power level low enough not to force the mixers into switching mode.
15. The arrangement of claim 9, wherein the combining means includes:
a summer that adds the input signal to the intermodulation cancellation signal.
a summer that adds the input signal to the intermodulation cancellation signal.
16. The arrangement of claim 9, wherein:
the input signal is at intermediate frequency (IF); and the combining means constitutes means for substantially eliminating the out-of-channel intermodulation distortion in a broadcast signal at radio frequency (RF).
the input signal is at intermediate frequency (IF); and the combining means constitutes means for substantially eliminating the out-of-channel intermodulation distortion in a broadcast signal at radio frequency (RF).
17. The arrangement of claim 9, wherein:
the input signal is a digital television (DTV) signal.
the input signal is a digital television (DTV) signal.
18. The arrangement of claim 9, wherein:
the input signal is an analog television signal.
the input signal is an analog television signal.
19. The arrangement of claim 18, wherein:
the analog television signal is an NTSC-format television signal.
the analog television signal is an NTSC-format television signal.
20. A method for correcting out-of-channel intermodulation distortion in an input signal, the method comprising:
receiving the input signal at both inputs of a first mixer, and providing a first mixer output signal;
receiving the input signal and the first mixer output signal at a second mixer, and providing an intermodulation cancellation signal; and combining the input signal and the intermodulation cancellation signal to output a corrected signal that provides for substantial elimination of the out-of-channel intermodulation distortion.
receiving the input signal at both inputs of a first mixer, and providing a first mixer output signal;
receiving the input signal and the first mixer output signal at a second mixer, and providing an intermodulation cancellation signal; and combining the input signal and the intermodulation cancellation signal to output a corrected signal that provides for substantial elimination of the out-of-channel intermodulation distortion.
21. The method of claim 20, further comprising:
adjusting relative timing of the input signal and the intermodulation cancellation signal.
adjusting relative timing of the input signal and the intermodulation cancellation signal.
22. The method of claim 20, further comprising:
adjusting attributes of the signal output by the second mixer.
adjusting attributes of the signal output by the second mixer.
23. The method of claim 22, wherein the attribute adjusting step includes:
adjusting the intermodulation component phase of the signal output by the second mixer to correspond to the intermodulation component phase of the input signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
adjusting the intermodulation component phase of the signal output by the second mixer to correspond to the intermodulation component phase of the input signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
24. The method of claim 22, wherein the attribute adjusting step includes:
adjusting the intermodulation component amplitude of the signal output by the second mixer to correspond to the intermodulation component amplitude of the input signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
adjusting the intermodulation component amplitude of the signal output by the second mixer to correspond to the intermodulation component amplitude of the input signal, so as to substantially eliminate the out-of-channel intermodulation distortion in a broadcast DTV signal at radio frequency (RF).
25. The method of claim 20, wherein:
receiving, at the first and second mixers, signals that are at a power level low enough not to force the mixers into switching mode.
receiving, at the first and second mixers, signals that are at a power level low enough not to force the mixers into switching mode.
26. The method of claim 20, wherein the combining step includes:
adding the input signal to the intermodulation cancellation signal.
adding the input signal to the intermodulation cancellation signal.
27. The method of claim 20, wherein:
the input signal is at intermediate frequency (IF); and the combining step constitutes substantially eliminating the out-of-channel intermodulation distortion in a broadcast signal at radio frequency (RF).
the input signal is at intermediate frequency (IF); and the combining step constitutes substantially eliminating the out-of-channel intermodulation distortion in a broadcast signal at radio frequency (RF).
28. The method of claim 20, wherein:
the input signal is a digital television (DTV) signal.
the input signal is a digital television (DTV) signal.
29. The method of claim 20, wherein:
the input signal is an analog television signal.
the input signal is an analog television signal.
30. The method of claim 29, wherein:
the analog television signal is an NTSC-format television signal.
the analog television signal is an NTSC-format television signal.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US13275898A | 1998-08-12 | 1998-08-12 | |
| US09/132,758 | 1998-08-12 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2254375A1 true CA2254375A1 (en) | 2000-02-12 |
Family
ID=29548215
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA 2254375 Abandoned CA2254375A1 (en) | 1998-08-12 | 1998-11-13 | Intermodulation correction for combined analog- and digital-format rf transmitters |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA2254375A1 (en) |
-
1998
- 1998-11-13 CA CA 2254375 patent/CA2254375A1/en not_active Abandoned
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