CA2249564C - Filter and method and apparatus for manufacturing filters - Google Patents
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
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- H01P1/20—Frequency-selective devices, e.g. filters
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Abstract
A method of producing filters using lower unloaded Q factor components than filters with the same performance characteristics but requiring higher unloaded Q factor components. The method comprises the steps of defining a desired filter characteristic and applying an algorithm which provides a filter having infinite Q factor elements and having a theoretical characteristic corresponding to the desired characteristic transformed to compensate for the difference between finite Q
factor and infinite Q factor elements.
factor and infinite Q factor elements.
Description
FILTER AND METHOD AND APPARATUS FOR MANUFACTURING FILTERS
The present invention relates to filters and to a method and apparatus for manufacturing filters, and relates particularly, but not exclusively, to microwave filters and a method and apparatus for manufacturing microwave filters.
Microwave filters are often constructed from networks of coupled passive resonators, each passive resonator having a finite unloaded Q factor. In narrow bandwidth applications, the resistive loss associated with this finite unloaded Q
factor can lead to significant reduction in achievable performance, and in bandpass applications, designs with a good input and output reflection coefficient will exhibit significant bandpass loss variation.
In the narrow band bandstop case the resistive loss manifests itself as a roll off of insertion loss into the pass band, and also limits the achievable notch depth. The combination of these two effects limits the achievable selectivity from a bandstop filter designed using previously available techniques.
In an existing bandstop filter, resonators are coupled off from a main through transmission line with an electrical separation of an odd number of quarter wavelengths, as shown in Figure 1. Each resonator couples loss into the system, giving rise to the problems outlined above.
In various applications of microwave filters, such as in base stations for cellular telecommunications, the above difficulties are addressed by using components having very high Q factors, typically up to 40, 000. However, this increases the physical size of the devices involved, whereas it is usually desirable in such applications to make the devices as compact as possible.
Preferred embodiments of the present invention seek to provide a filter which, although constructed using finite Q
-dements, does not suffer from a reduction in selectivity as a result of resistive losses caused by these finite Q factor elements.
R'O 97/36339 PCT/GB97/00786 Preferred embodiments of the present invention also seek to achieve a desired filter characteristic using components having lower unloaded Q factor than in the case of the prior art.
Preferred embodiments of the present invention also seek to provide a bandstop/pass filter having a steep transition between the stop and pass band and using lower value unloaded Q factor components than in the case of the prior art.
According to an aspect of the present invention, there is provided a method of designing a filter, the method comprising defining a desired filter characteristic, and applying an algorithm to the desired characteristic to provide a filter having infinite Q factor elements and having a theoretical characteristic corresponding to the desired characteristic transformed to compensate for the difference between finite Q
factor and infinite Q factor elements.
According to another aspect of the present invention, there is provided a method of manufacturing a filter, the method comprising the steps of designing a filter according to a method as defined above, and constructing using finite Q
factor elements a filter corresponding to the theoretical filter.
This provides the advantage of a filter design technique which takes resistive losses of the individual components, such as inductors and capacitors, of the filter into account, and therefore enables a filter having a desired characteristic to be designed using finite Q value components. This in turn enables a filter having a particular characteristic to be realised using lower unloaded Q factor components than in the case of the prior art, which in turn enables the filter to be constructed more compactly than in the case of the prior art.
According to another aspect of the present invention, there is provided an apparatus for use in manufacturing filters, the apparatus comprising an input means ..in which a desired filter characteristic is defined in use, and means for applying an algorithm to the desired characteristic to provide a filter having infinite Q factor elements and having a theoretical- characteristic corresponding to the desired ,characteristic transformed to compensate for the difference between infinite Q and finite Q factor elements.
According to a further aspect of the invention, there is provided a filter manufactured according to a method or using an apparatus as defined above.
This has the advantage of enabling the realisation of a filter having lower Q value components than in the case of the prior art, which in turn enables the construction of a more compact filter.
According to a further aspect of the invention, there is provided a filter comprising first and second resonators interconnected by a quadruplet of impedance inverters, a ladder network connected to the quadruplet of impedance inverters via a series resistor and comprising a plurality of further resonators, wherein adjacent further resonators of the ladder network are coupled to each other by respective impedance inverters.
In a preferred embodiment, the filter is a reflection mode filter.
The filter is preferably a microwave filter.
A filter may be a bandstop and/or a band pass filter.
Preferably, the step of applying said algorithm comprises shifting the pole/zero plot of the desired filter characteristic by a constant amount.
A preferred embodiment of the invention will now be described, by way of example only and not in any limitative sense, with reference to the accompanying drawings, in which:
Figure 1 shows a conventional bandstop filter;
Figure 2 shows a reflection mode filter comprising a low loss circulator connected to an input of a microwave band pass resonator;
Figure 3 shows a lossless low pass ladder network;
Figure 4 shows a network comprising a resistive -.attenuator followed by a lossless ladder network in which N=3 ;
Figure 5 shows a complete synthesis cycle for a degree 4 network:
The present invention relates to filters and to a method and apparatus for manufacturing filters, and relates particularly, but not exclusively, to microwave filters and a method and apparatus for manufacturing microwave filters.
Microwave filters are often constructed from networks of coupled passive resonators, each passive resonator having a finite unloaded Q factor. In narrow bandwidth applications, the resistive loss associated with this finite unloaded Q
factor can lead to significant reduction in achievable performance, and in bandpass applications, designs with a good input and output reflection coefficient will exhibit significant bandpass loss variation.
In the narrow band bandstop case the resistive loss manifests itself as a roll off of insertion loss into the pass band, and also limits the achievable notch depth. The combination of these two effects limits the achievable selectivity from a bandstop filter designed using previously available techniques.
In an existing bandstop filter, resonators are coupled off from a main through transmission line with an electrical separation of an odd number of quarter wavelengths, as shown in Figure 1. Each resonator couples loss into the system, giving rise to the problems outlined above.
In various applications of microwave filters, such as in base stations for cellular telecommunications, the above difficulties are addressed by using components having very high Q factors, typically up to 40, 000. However, this increases the physical size of the devices involved, whereas it is usually desirable in such applications to make the devices as compact as possible.
Preferred embodiments of the present invention seek to provide a filter which, although constructed using finite Q
-dements, does not suffer from a reduction in selectivity as a result of resistive losses caused by these finite Q factor elements.
R'O 97/36339 PCT/GB97/00786 Preferred embodiments of the present invention also seek to achieve a desired filter characteristic using components having lower unloaded Q factor than in the case of the prior art.
Preferred embodiments of the present invention also seek to provide a bandstop/pass filter having a steep transition between the stop and pass band and using lower value unloaded Q factor components than in the case of the prior art.
According to an aspect of the present invention, there is provided a method of designing a filter, the method comprising defining a desired filter characteristic, and applying an algorithm to the desired characteristic to provide a filter having infinite Q factor elements and having a theoretical characteristic corresponding to the desired characteristic transformed to compensate for the difference between finite Q
factor and infinite Q factor elements.
According to another aspect of the present invention, there is provided a method of manufacturing a filter, the method comprising the steps of designing a filter according to a method as defined above, and constructing using finite Q
factor elements a filter corresponding to the theoretical filter.
This provides the advantage of a filter design technique which takes resistive losses of the individual components, such as inductors and capacitors, of the filter into account, and therefore enables a filter having a desired characteristic to be designed using finite Q value components. This in turn enables a filter having a particular characteristic to be realised using lower unloaded Q factor components than in the case of the prior art, which in turn enables the filter to be constructed more compactly than in the case of the prior art.
According to another aspect of the present invention, there is provided an apparatus for use in manufacturing filters, the apparatus comprising an input means ..in which a desired filter characteristic is defined in use, and means for applying an algorithm to the desired characteristic to provide a filter having infinite Q factor elements and having a theoretical- characteristic corresponding to the desired ,characteristic transformed to compensate for the difference between infinite Q and finite Q factor elements.
According to a further aspect of the invention, there is provided a filter manufactured according to a method or using an apparatus as defined above.
This has the advantage of enabling the realisation of a filter having lower Q value components than in the case of the prior art, which in turn enables the construction of a more compact filter.
According to a further aspect of the invention, there is provided a filter comprising first and second resonators interconnected by a quadruplet of impedance inverters, a ladder network connected to the quadruplet of impedance inverters via a series resistor and comprising a plurality of further resonators, wherein adjacent further resonators of the ladder network are coupled to each other by respective impedance inverters.
In a preferred embodiment, the filter is a reflection mode filter.
The filter is preferably a microwave filter.
A filter may be a bandstop and/or a band pass filter.
Preferably, the step of applying said algorithm comprises shifting the pole/zero plot of the desired filter characteristic by a constant amount.
A preferred embodiment of the invention will now be described, by way of example only and not in any limitative sense, with reference to the accompanying drawings, in which:
Figure 1 shows a conventional bandstop filter;
Figure 2 shows a reflection mode filter comprising a low loss circulator connected to an input of a microwave band pass resonator;
Figure 3 shows a lossless low pass ladder network;
Figure 4 shows a network comprising a resistive -.attenuator followed by a lossless ladder network in which N=3 ;
Figure 5 shows a complete synthesis cycle for a degree 4 network:
Figure 6 shows a network corresponding to the network of Figure 5 modified by the replacement of the first four elements shown in Figure 5 by a quadruplet of impedance inverters and two capacitors;
Figure 7 shows a reflection mode band stop microwave filter;
Figure 8 shows the simulated frequency response of the filter of Figure 7;
Figure 9 shows a general Nth degree circuit for the band stop reflection mode filter of Figure 7; and Figure 10 shows the measured frequency response of an actual filter.
Referring to Figure 2, there is shown a resonant circuit with finite loss which is coupled to one of the ports of a circulator. The transmission characteristic from ports 1 to 3 is the reflection coefficient from the network connected at port 2. If the input coupling to the resonant circuit is adjusted so that the resistive part of its input impedance is matched to the circulator, then at resonance all power supplied at port 1 will emerge at port 2 and be absorbed in the resistive part of the resonator.
Hence there is no transmission to port 3 and the 1-3 transmission characteristic is that of a resonator with infinite unloaded Q. For a resonator of centre frequency fo and 3 dB bandwidth B the unloaded Q is given by Qu _ 2~ C1) B
For example, if B = 250 KHz and fo=1 GHz, then Qu = 8000.
It can therefore be seen that the previously considered specification can be met with much lower Q resonators, with a consequent reduction in physical size, provided that a design procedure for multi-element filters is available.
In order to provide such a design procedure, the magnitude squared of the input reflection coefficient of a lossless lowpass prototype filter may be expressed as Sm (Jw),I2- - FN2 (w) 1+FNZ ( w ) Where FN(w) is the characteristic function for a Butterworth, Chebychev, Elliptic Function or other prototype network. This reflection coefficient may readily be synthesised as a lossless lowpass ladder network which is terminated in a resistor as shown in Figure 3. In order to include eventual resonator losses we can multiply by an arbitrary constant K to yield;
4 Sm( jw) I 2 - KFN2(w) 1 + FN2(w) This may now be synthesised as a resistive attenuator followed by a lossless ladder network which in turn is terminated in a resistor, as shown in Figure 4.
The resultant network now contains dissipative elements.
However, these are not distributed throughout the Nth degree network but remain concentrated at the input. A network containing lossy elements is required so that the required response can be achieved using finite Q resonators.
In order to achieve this, compensation is made for eventual resonator loss by shifting the poles and zeros of S11(p) towards the jw axis by a constant amount a, i.e.
p -~ p - a Thus for Sll(p) - K Nfp) D (p) -" Then S11 ( p-a ) - K N (p-a 1 (1) D (p-a) The reflection coefficient given in (1) may now be synthesised as one port impedance function. First the maximum value of K must be uniquely determined for any specific value of a, so that the resultant network is passive and has minimum loss for a given value of a.
The specific frequencies ~o and values of K are then determined such that:
I Sll ( P-a ) I 2 - 1 and d I S11 ( P-a ) I
do are simultaneously satisfied with the minimum value of K.
Having found the values coo and a then formulate S11 ( p-a ) - K N p-a~ - N1 D (p-a) D1(p) The input impedance Zin (p) may now be found from Zin(P) - _D~lP) + N~-LPG
' D~(P) - N~ (P) Zin has a transmission zero at coo and thus cannot be synthesised as a ladder network.
However any positive real function may be synthesised using Brunes' Procedure as set out in O Brune. "Synthesis of a Finite Two-Terminal Network whose Driving Point Impedance is a Prescribed Function of Frequency". Journal of Maths and Physics, Vol X no 3, 1931, p 191.
Given Yin(p) - 1 and evaluating Yin at p = j~o it is Zin(p) WO 97/36339 PCTlGB97/00786 -found that this is a pure susceptance. This is a consequence of the network being purely reflective at that frequency. This susceptance will be negative i.e.
Yin (j~o) - -jB
Extracting a shunt negative capacitor of value -Cl from Yin provides Yl(p) - Yin(p) + Clp Observing that Yl is one degree higher in p than Yin then since Yin ( jc~o) was purely imaginary, Yl must be equal to zero at this frequency . Consequently Yl ( p ) must have a quadratic f actor at p = ~
Thus Y,(P) - (PZ + X02) N(p) P(P) Inverting Yl ( p ) to form Z1 ( p ) a series branch composed of a parallel tuned circuit can be extracted, ie Z~(P) - ~(P) - AP - ZZ(P) ( PZ+~o2 ) N ( P ) p2+~o2 A is the residue of Zl ( p ) at p =jwo . Inverting Z2 ( p ) to obtain Y2 ( p ) then a shunt capacitor may be extracted from Y~ ( p ) as follows:
C3 - Yz(P) P p=°°
~. and Y3 ( P ) - Yz ( P ) -C3P
i i i i _g_ Forming Z, ( p ) + _ 1 Y3(p) A series resistor equal in value to the minimum real part of Z,(p) must now be extracted. This may be evaluated from the minimum value of the even part of Z, ( p ) .
Thus Z5(p) - Z,(p) -R
where R - min Ev(Z,(p)) In most cases the minimum value of Z,(p) will occur at c~
- ~ and the remaining network may be synthesised as a lossy ladder network.
The complete synthesis cycle is shown for a degree 4 network in Figure 5.
It is important to note that the network shown in Figure is not immediately suitable for realisation using microwave resonators. However, it may readily be transformed into the network of Figure 6 which consists entirely of inverters, capacitors and resistors.
The capacitors shown in Figure 6 are initially lossless but are transformed into finite Q elements by the final simple modification.
p ~ p + a The resultant lowpass prototype network may then be converted into a bandpass network by applying the appropriate transformation for any particular type of resonator.
Example The procedure outlined has been applied successfully to the design of a bandstop filter with specification as outlined above.
_g_ A fourth degree Elliptic Function Filter was synthesised.
The choice of a was 0.093 corresponding to approximately 6 dB
out of band loss. The resultant network is shown in Figure 7.
The simulated response of this network is shown in Figure 8, from which it can be seen that the response achieves the desired specification. This actual filter has been constructed using coaxial resonators. The measured performance characteristics are shown in Figure 10 and are in excellent agreement with theory.
It will be appreciated by persons skilled, in the art that the above embodiment has been described by way of example only, and not in any limited sense, and that various alterations and modifications are possible without departure from the scope of the invention as defined by the appended claims.
Figure 7 shows a reflection mode band stop microwave filter;
Figure 8 shows the simulated frequency response of the filter of Figure 7;
Figure 9 shows a general Nth degree circuit for the band stop reflection mode filter of Figure 7; and Figure 10 shows the measured frequency response of an actual filter.
Referring to Figure 2, there is shown a resonant circuit with finite loss which is coupled to one of the ports of a circulator. The transmission characteristic from ports 1 to 3 is the reflection coefficient from the network connected at port 2. If the input coupling to the resonant circuit is adjusted so that the resistive part of its input impedance is matched to the circulator, then at resonance all power supplied at port 1 will emerge at port 2 and be absorbed in the resistive part of the resonator.
Hence there is no transmission to port 3 and the 1-3 transmission characteristic is that of a resonator with infinite unloaded Q. For a resonator of centre frequency fo and 3 dB bandwidth B the unloaded Q is given by Qu _ 2~ C1) B
For example, if B = 250 KHz and fo=1 GHz, then Qu = 8000.
It can therefore be seen that the previously considered specification can be met with much lower Q resonators, with a consequent reduction in physical size, provided that a design procedure for multi-element filters is available.
In order to provide such a design procedure, the magnitude squared of the input reflection coefficient of a lossless lowpass prototype filter may be expressed as Sm (Jw),I2- - FN2 (w) 1+FNZ ( w ) Where FN(w) is the characteristic function for a Butterworth, Chebychev, Elliptic Function or other prototype network. This reflection coefficient may readily be synthesised as a lossless lowpass ladder network which is terminated in a resistor as shown in Figure 3. In order to include eventual resonator losses we can multiply by an arbitrary constant K to yield;
4 Sm( jw) I 2 - KFN2(w) 1 + FN2(w) This may now be synthesised as a resistive attenuator followed by a lossless ladder network which in turn is terminated in a resistor, as shown in Figure 4.
The resultant network now contains dissipative elements.
However, these are not distributed throughout the Nth degree network but remain concentrated at the input. A network containing lossy elements is required so that the required response can be achieved using finite Q resonators.
In order to achieve this, compensation is made for eventual resonator loss by shifting the poles and zeros of S11(p) towards the jw axis by a constant amount a, i.e.
p -~ p - a Thus for Sll(p) - K Nfp) D (p) -" Then S11 ( p-a ) - K N (p-a 1 (1) D (p-a) The reflection coefficient given in (1) may now be synthesised as one port impedance function. First the maximum value of K must be uniquely determined for any specific value of a, so that the resultant network is passive and has minimum loss for a given value of a.
The specific frequencies ~o and values of K are then determined such that:
I Sll ( P-a ) I 2 - 1 and d I S11 ( P-a ) I
do are simultaneously satisfied with the minimum value of K.
Having found the values coo and a then formulate S11 ( p-a ) - K N p-a~ - N1 D (p-a) D1(p) The input impedance Zin (p) may now be found from Zin(P) - _D~lP) + N~-LPG
' D~(P) - N~ (P) Zin has a transmission zero at coo and thus cannot be synthesised as a ladder network.
However any positive real function may be synthesised using Brunes' Procedure as set out in O Brune. "Synthesis of a Finite Two-Terminal Network whose Driving Point Impedance is a Prescribed Function of Frequency". Journal of Maths and Physics, Vol X no 3, 1931, p 191.
Given Yin(p) - 1 and evaluating Yin at p = j~o it is Zin(p) WO 97/36339 PCTlGB97/00786 -found that this is a pure susceptance. This is a consequence of the network being purely reflective at that frequency. This susceptance will be negative i.e.
Yin (j~o) - -jB
Extracting a shunt negative capacitor of value -Cl from Yin provides Yl(p) - Yin(p) + Clp Observing that Yl is one degree higher in p than Yin then since Yin ( jc~o) was purely imaginary, Yl must be equal to zero at this frequency . Consequently Yl ( p ) must have a quadratic f actor at p = ~
Thus Y,(P) - (PZ + X02) N(p) P(P) Inverting Yl ( p ) to form Z1 ( p ) a series branch composed of a parallel tuned circuit can be extracted, ie Z~(P) - ~(P) - AP - ZZ(P) ( PZ+~o2 ) N ( P ) p2+~o2 A is the residue of Zl ( p ) at p =jwo . Inverting Z2 ( p ) to obtain Y2 ( p ) then a shunt capacitor may be extracted from Y~ ( p ) as follows:
C3 - Yz(P) P p=°°
~. and Y3 ( P ) - Yz ( P ) -C3P
i i i i _g_ Forming Z, ( p ) + _ 1 Y3(p) A series resistor equal in value to the minimum real part of Z,(p) must now be extracted. This may be evaluated from the minimum value of the even part of Z, ( p ) .
Thus Z5(p) - Z,(p) -R
where R - min Ev(Z,(p)) In most cases the minimum value of Z,(p) will occur at c~
- ~ and the remaining network may be synthesised as a lossy ladder network.
The complete synthesis cycle is shown for a degree 4 network in Figure 5.
It is important to note that the network shown in Figure is not immediately suitable for realisation using microwave resonators. However, it may readily be transformed into the network of Figure 6 which consists entirely of inverters, capacitors and resistors.
The capacitors shown in Figure 6 are initially lossless but are transformed into finite Q elements by the final simple modification.
p ~ p + a The resultant lowpass prototype network may then be converted into a bandpass network by applying the appropriate transformation for any particular type of resonator.
Example The procedure outlined has been applied successfully to the design of a bandstop filter with specification as outlined above.
_g_ A fourth degree Elliptic Function Filter was synthesised.
The choice of a was 0.093 corresponding to approximately 6 dB
out of band loss. The resultant network is shown in Figure 7.
The simulated response of this network is shown in Figure 8, from which it can be seen that the response achieves the desired specification. This actual filter has been constructed using coaxial resonators. The measured performance characteristics are shown in Figure 10 and are in excellent agreement with theory.
It will be appreciated by persons skilled, in the art that the above embodiment has been described by way of example only, and not in any limited sense, and that various alterations and modifications are possible without departure from the scope of the invention as defined by the appended claims.
Claims (10)
1. A method of manufacturing a reflection mode filter network comprising finite Q-factor resonators and having a desired filter characteristic of a reflection function with a constant loss off-set in which the reflection function is substantially identical to the reflection function of a theoretical filter comprising infinite Q factor resonators and comprising the steps of:
specifying the desired reflection function;
transforming the desired reflection function by multiplying by a constant corresponding to loss of the filter network and so as to shift the poles and zeros of a plot of the reflection function on the imaginary/real plane towards the imaginary axis by a real amount;
synthesising a theoretical filter network including lossless resonators and having the transformed reflection function;
converting the theoretical filter network into an equivalent network comprising inverters, capacitors and resistors;
re-transforming the reflection function of the converted theoretical filter network so as to shift the poles and zeros of a plot of the reflection function away from the imaginary axis of the imaginary/real plane by the real amount so as to provide a low pass prototype reflection mode filter network having the desired reflection function and a constant loss off-set;
applying a frequency transformation to the low pass prototype reflection mode filter network to provide a reflection mode filter network configuration of electrical components, and specific values of the electrical components, having the desired filter characteristic;
assembling electrical components selected from inverters, capacitors and resistors according to the reflection mode filter network configuration and having the specific values to provide a reflection mode filter network having finite Q-factor resonators and the desired filter characteristic.
specifying the desired reflection function;
transforming the desired reflection function by multiplying by a constant corresponding to loss of the filter network and so as to shift the poles and zeros of a plot of the reflection function on the imaginary/real plane towards the imaginary axis by a real amount;
synthesising a theoretical filter network including lossless resonators and having the transformed reflection function;
converting the theoretical filter network into an equivalent network comprising inverters, capacitors and resistors;
re-transforming the reflection function of the converted theoretical filter network so as to shift the poles and zeros of a plot of the reflection function away from the imaginary axis of the imaginary/real plane by the real amount so as to provide a low pass prototype reflection mode filter network having the desired reflection function and a constant loss off-set;
applying a frequency transformation to the low pass prototype reflection mode filter network to provide a reflection mode filter network configuration of electrical components, and specific values of the electrical components, having the desired filter characteristic;
assembling electrical components selected from inverters, capacitors and resistors according to the reflection mode filter network configuration and having the specific values to provide a reflection mode filter network having finite Q-factor resonators and the desired filter characteristic.
2. A method of manufacturing a transmission mode filter including the steps of manufacturing a reflection mode filter network as claimed in claim 1 and the step of attaching a three port circulator to an input port of the reflection mode filter network and providing an input port to the circulator and an output port from the circulator and matching the resistive part of the input impedance of the reflection mode filter network to the circulator.
3. A filter manufactured according to the method of claims 1 and 2, in which the filter is a microwave filter.
4. A filter having a network of finite Q factor resonators and comprising electrical components chosen from capacitors, inverters and resistors in which the capacitance or resistance values of the respective electrical components and their configuration are selected such that the filter has a filter characteristic comprising a selectivity and a constant loss off-set, and the selectivity is substantially identical to a predetermined selectivity of a theoretical filter including infinite Q factor resonators, in which the network is a reflection mode filter network having an input connected to an input node of a first inverter and the first inverter having an output node connected in series with a resistance, and having a finite Q-factor resonator connected in parallel between the input and output nodes by a second and a third inverter.
5. A filter as claimed in claim 4, and having a second finite Q-factor resonator connected in parallel across the first inverter via the third inverter and a fourth inverter which is connected to the output node of the first inverter.
6. A filter as claimed in claim 4 or claim 5, and having a ladder network including a finite Q-factor resonator attached in series with the resistance.
7. A filter as claimed in any of claims 4 to 6 and including a three port circulator attached to the input of the reflection mode filter network and having its impedance-matched to the resistive part of the input impedance of the reflection mode filter network so that the filter acts as a transmission mode filter.
8. A filter as claimed in any of claims 4 to 7, in which the network is a bandstop filter.
9. A filter as claimed in any of claims 4 to 7, in which the network is a bandpass filter.
10. A filter as claimed in any of claims 4 to 9, which is a microwave filter.
Applications Claiming Priority (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GBGB9606178.3A GB9606178D0 (en) | 1996-03-23 | 1996-03-23 | Filter and method and apparatus for manufacturing filters |
| GB9606178.3 | 1996-03-23 | ||
| PCT/GB1997/000786 WO1997036339A1 (en) | 1996-03-23 | 1997-03-19 | Filter and method and apparatus for manufacturing filters |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CA2249564A1 CA2249564A1 (en) | 1997-10-02 |
| CA2249564C true CA2249564C (en) | 2005-01-11 |
Family
ID=10790935
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002249564A Expired - Fee Related CA2249564C (en) | 1996-03-23 | 1997-03-19 | Filter and method and apparatus for manufacturing filters |
Country Status (6)
| Country | Link |
|---|---|
| EP (1) | EP0888645B1 (en) |
| AT (1) | ATE223111T1 (en) |
| CA (1) | CA2249564C (en) |
| DE (1) | DE69714978T2 (en) |
| GB (2) | GB9606178D0 (en) |
| WO (1) | WO1997036339A1 (en) |
-
1996
- 1996-03-23 GB GBGB9606178.3A patent/GB9606178D0/en active Pending
-
1997
- 1997-03-19 AT AT97908392T patent/ATE223111T1/en not_active IP Right Cessation
- 1997-03-19 WO PCT/GB1997/000786 patent/WO1997036339A1/en active IP Right Grant
- 1997-03-19 EP EP97908392A patent/EP0888645B1/en not_active Expired - Lifetime
- 1997-03-19 CA CA002249564A patent/CA2249564C/en not_active Expired - Fee Related
- 1997-03-19 DE DE69714978T patent/DE69714978T2/en not_active Expired - Lifetime
- 1997-03-19 GB GB9819742A patent/GB2326296B/en not_active Revoked
Also Published As
| Publication number | Publication date |
|---|---|
| GB9819742D0 (en) | 1998-11-04 |
| GB2326296B (en) | 2000-03-08 |
| WO1997036339A1 (en) | 1997-10-02 |
| GB9606178D0 (en) | 1996-05-29 |
| CA2249564A1 (en) | 1997-10-02 |
| ATE223111T1 (en) | 2002-09-15 |
| DE69714978T2 (en) | 2009-09-24 |
| DE69714978D1 (en) | 2002-10-02 |
| EP0888645A1 (en) | 1999-01-07 |
| GB2326296A (en) | 1998-12-16 |
| EP0888645B1 (en) | 2002-08-28 |
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| MKLA | Lapsed |
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