CA2244609A1 - Digital modulation process, and modulator implementing the process - Google Patents
Digital modulation process, and modulator implementing the process Download PDFInfo
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- CA2244609A1 CA2244609A1 CA002244609A CA2244609A CA2244609A1 CA 2244609 A1 CA2244609 A1 CA 2244609A1 CA 002244609 A CA002244609 A CA 002244609A CA 2244609 A CA2244609 A CA 2244609A CA 2244609 A1 CA2244609 A1 CA 2244609A1
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/20—Modulator circuits; Transmitter circuits
- H04L27/2032—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner
- H04L27/2053—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases
- H04L27/206—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers
- H04L27/2067—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states
- H04L27/2071—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states in which the data are represented by the carrier phase, e.g. systems with differential coding
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03828—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties
- H04L25/03834—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties using pulse shaping
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- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
- Signal Processing For Digital Recording And Reproducing (AREA)
- Radar Systems Or Details Thereof (AREA)
- Supplying Of Containers To The Packaging Station (AREA)
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Abstract
The successive symbols (a j) of a digital stream are converted into phase increments (.DELTA.~) which are accumulated. A modulating phase (~) is obtained by filtering the accumulated phase. A complex signal is produced whose argument represents the modulating phase. Two phase quadrature radio waveforms are respectively modulated on the basis of that complex signal, and a radio signal resulting from a combination of the two modulated waveforms is transmitted. The complex signal is, in turn, filtered digitally. Its real and imaginary components (I,Q) are converted into analog form, and are subjected to anti-aliasing analog filtering and then mixed with the two radio waveforms. Appropriate sizing of the digital filters provides efficient modulation with small envelope variations, causing little adjacent channel interference and a low error rate.
Description
DIGITAL MODULATION PROCESS, AND MODULATOR
IMPLEMENTING THE PROCESS
This invention concerns digital radio communication systems. It concerns, in particular, the methods of modulation implemented in such 5 systems.
Digital modulation is generally designed to combine the following three requirements: a high transmission rate, minimum spectrum occupancy and a low transmission error rate under various operating conditions.
Various methods were proposed in the past with a view to 10 achieving a high transmission rate on a channel with reduced spectral bandwidth (transmission rate exceeding 1 biVs/Hz).
The first group of methods uses multi-level frequency modulation as a basis, together with adequate filtering of the modulating signal (e.g.
Gaussian filtering used with GMSK modulation) in order to reduce adjacent channel interference. These methods have the advantage that they are easily applied, and result in modulated signals of constant envelope. They consequently permit transmitters to be fitted with power amplifiers which operate in the saturated state. These amplifiers are readily available, cheap and very efficient. However, in order to comply with constraints relating to 20 adjacent channel interference, the modulation index must be limited considerably, and the modulating signal thoroughly filtered. This causes the symbol spacing to be reduced, and this adversely affects the noise immunity of the modulation. In other words, the sensitivity of the radio receivers is limited .
IMPLEMENTING THE PROCESS
This invention concerns digital radio communication systems. It concerns, in particular, the methods of modulation implemented in such 5 systems.
Digital modulation is generally designed to combine the following three requirements: a high transmission rate, minimum spectrum occupancy and a low transmission error rate under various operating conditions.
Various methods were proposed in the past with a view to 10 achieving a high transmission rate on a channel with reduced spectral bandwidth (transmission rate exceeding 1 biVs/Hz).
The first group of methods uses multi-level frequency modulation as a basis, together with adequate filtering of the modulating signal (e.g.
Gaussian filtering used with GMSK modulation) in order to reduce adjacent channel interference. These methods have the advantage that they are easily applied, and result in modulated signals of constant envelope. They consequently permit transmitters to be fitted with power amplifiers which operate in the saturated state. These amplifiers are readily available, cheap and very efficient. However, in order to comply with constraints relating to 20 adjacent channel interference, the modulation index must be limited considerably, and the modulating signal thoroughly filtered. This causes the symbol spacing to be reduced, and this adversely affects the noise immunity of the modulation. In other words, the sensitivity of the radio receivers is limited .
Another group of methods uses phase-shift keying (PSK) and, if necessary, differential phase shift keying (DPSK) as a basis, and the resulting signal is filtered to ensure that standards relating to adjacent channel interference are complied with. In general, a filter satisfying the Nyquist 5 criterion is used in order to limit inter-symbol interference. These methods generally provide satisfactory sensitivity at the expense of a large variation in the amplitude of the radio signal. Very linear amplifiers are therefore necessary, and they are difficult to design and set up. In addition, they are generally inefficient, and this seriously affects the autonomy of mobile stations.
10 A non-linear amplifier can be used in conjunction with a linearising method, but such method complicates a transmitter very considerably if there are large envelope variations.
Other solutions have also been proposed, e.g. in U.S. Patents 5,642,384 and 5,311,552, where an appropriate choice of a constellation and 15 of a coded modulation process prevents transitions in the constellation for which the phase change is relatively large. This permits the variation in amplitude of a radio signal to be reduced to values compatible with the characteristics of amplifiers which are easier to design. However, the reduction in amplitude is achieved at the expense of a considerable reduction 20 in the symbol spacing, which is very difficult to compensate by coding gains, in particular in the error rate range of the greatest importance to speech communications, i.e. for bit error rates (BER) of the order of 10-2, especially when the channel is affected by fading (Rayleigh fading).
An object of the present invention is to propose a digital 25 modulation group permitting joint optimisation of noise immunity, even in a channel affected by fading, adjacent channel interference, and variation in amplitude of the radio signal.
The invention thus proposes a digital modulation process wherein the successive symbols of a digital stream are converted into phase increments, an accumulated phase is obtained by adding the successive phase increments, a modulating phase is obtained by filtering the accumulated phase, a complex signal is produced whose argument represents the modulating phase, two phase quadrature radio waveforms are respectively modulated on the basis of said complex signal, and a radio signal resulting 10 from a combination of the two modulated waveforms is transmitted. According to the invention, said complex signal is digitally filtered, and digital signalsobtained from the real and imaginary components of the digitally filtered complex signal are converted into analog form before being respectively subjected to anti-aliasing analog filtering and then mixed with the two radio waveforms.
Said digital signals obtained from the real and imaginary components of the digitally fiitered complex signal typically consist of the real and imaginary components themselves. However, if an amplifier linearising process is used, by pre-distortion for example (see European patent 20 application No. 0 797 293), the real and imaginary components may be subject to correction before being converted into analog form. The use of a linearising process is not included directly in this invention. In many cases, the invention will permit such a process to be dispensed with. In other cases, it will permit the use of such processes to be simplified considerably (for 25 example, by not taking account of phase changes), in view of the small variations in the signal envelope permitted by an appropriate choice of parameters for filtering the accumulated phase and said real and imaginary components. The criteria for this selection will be specified further on.
The invention permits digital radio communication systems, in 5 particular professional radio communication systems, to be implemented in accordance with applicable standards relating to adjacent channel interference, and provides unequalled sensitivity and thus radio range, using power amplifier components which are readily available on the market and have a high power efficiency.
Another aspect of the invention relates to a digital modulator, including means for converting successive symbols of a digital stream into phase increments, a summator which accumulates the successive phase increments to produce an accumulated phase, a phase filter receiving the accumulated phase and producing a modulating phase, means for producing a 15 complex signal whose argument represents the modulating phase, and a modulator for respectively modulating two phase quadrature radio waveforms on the basis of said complex signal, and for transmitting a radio signal resulting from a combination of the two modulated waveforms, the modulator comprising a digital filter to which the complex signal is applied, analog-digital 20 converters respectively processing the digital signals obtained from the real and imaginary components of the digitally filtered complex signal, anti-aliasing analog filters receiving the output signals from the analog-digital converters, and two mixers each receiving one of the two radio waveforms and the output signal from one of the two anti-aliasing filters.
Other features and advantages of the invention will become apparent from the following description of non-limiting embodiments, with reference to the attached drawings where:
- Figures 1 and 2 are block diagrams of a digital modulator in accordance with the invention and an associated receiver respectively; and - Figures 3 and 4 are graphs respectively showing the constellation and spectrum of a digital modulator according to the invention.
The modulator shown in Figure 1 comprises a unit 10 which converts the successive symbols aj of a digital stream into phase increments 10 ~. The successive phase increments ~ produced by the unit 10 are accumulated by a summator 11. The unit 10 may merely consist of a register containing the possible values of the phase increments ~ and addressed by the current value of the symbol aj In the embodiment of Figure 1, the symbol stream aj may correspond either to a bit stream bj or another bit stream c; having with a lower transmission rate, processed by a redundancy encoder 12. The bit streams b and c; derive from digital sources such as speech encoders, data sources, etc., generally with error correction coding applied. If the encoder 12 is used, the modulator of Figure 1 operates in accordance with a coded modulation 20 (see G. Ungerboeck "Channel coding with multi-level/phase signals", IEEE
Trans. on Information Theory, Vol. IT-28, No. 1, January 1982, pages 55-67).
The unit 10, summator 11 and encoder 12 are timed by a clock signal CKS at the frequency 1/Ts of the symbols aj.
The summator 11 stores a integer digital value k representing an accumulated phase. This accumulated phase is thus stored as whole multiples of a sub-multiple of 7~, that is to say in the general form (k/P)~. For each cycle of the clock CKS, the accumulated phase is incremented by a value ~ depending on the current symbol aj. If each symbol represents m bits, M = 2m different values of the increment may be added in each cycle.
These M values are chosen so that the set of the increments is symmetrical with respect to the value 0 so that the spectrum is symmetrical. Values of k of the type k = k' x K will typically be used, where K/P represents the modulation index, and k' = -M+1, -M+3, ..., -1, 1, ..., M-3 or M-1. This choice of equally 0 distributed increments is not the only one possible. For example, k' = -7, -3, 3 or 7 could also be used if m=2. ~maX = (kmaX/P)~ designates the maximum value of the phase increment ~.
The accumulated phase is fed to a digital filter 15, referred to as phase filter, whose sampling frequency 1/Te, set by a clock signal CKE, is higher than the frequency 1/TS of the symbols aj (generally a multiple of that frequency).
The output signal from the phase filter 15 is a modulating phase ~, which a unit 16 converts into a complex signal, i.e. into two real signals, one (I) representing the real component of the complex signal, and the other (Q) representing the imaginary component.
That complex signal has a constant modulus, and an argument equil to the modulating phase ~. In other words, I = cos~ and Q = sin~. The unit 16 may merely consist of two read-only memory arrays addressed by the output of the filter 15 at every cycle of clock CKE.
The complex signal is filtered by a digital filter which, in the embodiment shown, consists of two identical filters 17 which respectively filter the components I and Q.
Two digital-analog converters 18 convert the output signals of the 5 two filters 17 into analog form. The two resulting analog signals are fed to low-pass filters 19 in order to eliminate spectral aliasing components. Using respective mixers 21, two quadrature radio waveforms at the carrier frequency, deriving from a local oscillator 20, are mixed with the signals deriving from the anti-aliasing filters 19. The two waveforms thus modulated are combined by a 10 summator 22 whose output is fed to the power amplifier 23 of the transmitter.
If the amplifier 23 were linearised by pre-distortion, it would be necessary to correct the filtered components I and Q, between the filters 17 and converters 18, before converting them into analog form.
The receiver shown in Figure 2 includes a low-noise amplifier 30 which amplifies the signal picked up by the antenna. Its output is converted to an intermediate frequency using a mixer 31. A band-pass filter 32 processes the intermediate frequency signal which is then amplified further by an amplifier 33. Two other mixers 34 provide baseband conversion by mixing with two quadrature waveforms. The two quadrature analog components 20 deriving from the mixers 34 are fed to identical low-pass filters 35, then converted into digital form by analog-digital converters 36. The digital components 1' and Q' deriving from the converters 36 are fed to a channel demodulator 37.
The demodulator 37 carries out demodulating operations 25 corresponding to the incomplete modulator consisting of the components 10, CA 02244609 l998-08-04 11, 15, 16 and (if necessary) 12 of the transmitter shown in Figure 1. Since this incomplete modulator essentially performs continuous phase modulation (CPM), the demodulator 37 may take the form of a conventional CPM
demodulator. It may, for example, be based on a demodulation trellis in order 5 to apply the Viterbi algorithm. The demodulator 37 delivers estimates b j or c of the bits bj or cj fed to the modulator.
Advantageously, the demodulator 37 may include two trellis. It uses either one of the trellis, depending on whether the encoder 12 is used at the transmitter or not. The first trellis includes modulation states. In principle, the number of these states is ML-1 x P, where L is the memory of the phase filter 15 expressed in number of samples, M is the number of points on the constellation, and P is the denominator of the modulation index. However, it is generally possible to considerably reduce the number of states of the demodulation trellis without adversely affecting the quality of reception significantly. The second one of the trellis further includes the coding states of the redundancy encoder 12, in accordance with the principle of coded modulations. This second trellis is employed if the encoder 12 is used at the transmitter.
In the modulator design, the values of the phase increments ~
20 are first chosen, as indicated hereabove. The filter 17 which processes the components I and Q and determines the spectral characteristics of the resulting signal is then constructed. The characteristics of this filter 17 must be as close as possible to those of the receiving filter consisting of the combination of filters 32 and 35.
CA 02244609 l998-08-04 _ 9 _ A advantageous form of the digital filters 17, used to process the components I and Q, is a filter with a finite impulse response selected to most closely fit a time characteristic of the form:
f(t) = Sinc(at/Ts).Sinc(,~t/Ts).e~ rvTs) (1) 5 where Ts is the duration of a symbol aj, and Sinc() is the cardinal sine function.
The approximation can be made by choosing the real coefficients cc, ~ and ~.
This provides digital filters whose restriction to a finite length is as accurate as possible by virtue of the fast decay of the Gaussian function. The secondary lobes caused by the limitation of the digital filter length are thus minimised.
The following step consists in defining the phase filter 15. The characteristics of this filter 15 are closely related to those of the digital filter 17.
A heuristic method is given hereunder based on the following mathematical property: the energy of a complex function eim(t) with unitary modulus is maximum in a filter whose spectral power template is the Fourier transform of a function h(t) (in other words, that energy is minimum outside the filter) if it satisfies the following equatio,n:
J h(u-t).eim(t)dt = ~,(u) eim(u) where ~(u) is a real function.
The following algorithm is used to define the phase filter 15:
1) A power template filter is selected, i.e. a function h(t) whose Fourier transform represents the required spectral template. A filter identical to the one selected as the l-Q filter is typically chosen. Other choices are obviously possible. In general, it is preferable to use a filter whose digital implementation with a fairly short finite impulse response is possible.
2) A function q)0 equal to 0 where t<0, equal to ~maXt/Ts where 0<t<Ts and equal to ~maX where t>Ts is used as a first approximation of the phase change function, i.e. of the function available at the output of the phase filter when the maximum phase increment ~maX is fed to the accumulator 11.
5 Other approximations using continuous functions equal to 0 where t<0 and ~max where t>Ts could be used.
10 A non-linear amplifier can be used in conjunction with a linearising method, but such method complicates a transmitter very considerably if there are large envelope variations.
Other solutions have also been proposed, e.g. in U.S. Patents 5,642,384 and 5,311,552, where an appropriate choice of a constellation and 15 of a coded modulation process prevents transitions in the constellation for which the phase change is relatively large. This permits the variation in amplitude of a radio signal to be reduced to values compatible with the characteristics of amplifiers which are easier to design. However, the reduction in amplitude is achieved at the expense of a considerable reduction 20 in the symbol spacing, which is very difficult to compensate by coding gains, in particular in the error rate range of the greatest importance to speech communications, i.e. for bit error rates (BER) of the order of 10-2, especially when the channel is affected by fading (Rayleigh fading).
An object of the present invention is to propose a digital 25 modulation group permitting joint optimisation of noise immunity, even in a channel affected by fading, adjacent channel interference, and variation in amplitude of the radio signal.
The invention thus proposes a digital modulation process wherein the successive symbols of a digital stream are converted into phase increments, an accumulated phase is obtained by adding the successive phase increments, a modulating phase is obtained by filtering the accumulated phase, a complex signal is produced whose argument represents the modulating phase, two phase quadrature radio waveforms are respectively modulated on the basis of said complex signal, and a radio signal resulting 10 from a combination of the two modulated waveforms is transmitted. According to the invention, said complex signal is digitally filtered, and digital signalsobtained from the real and imaginary components of the digitally filtered complex signal are converted into analog form before being respectively subjected to anti-aliasing analog filtering and then mixed with the two radio waveforms.
Said digital signals obtained from the real and imaginary components of the digitally fiitered complex signal typically consist of the real and imaginary components themselves. However, if an amplifier linearising process is used, by pre-distortion for example (see European patent 20 application No. 0 797 293), the real and imaginary components may be subject to correction before being converted into analog form. The use of a linearising process is not included directly in this invention. In many cases, the invention will permit such a process to be dispensed with. In other cases, it will permit the use of such processes to be simplified considerably (for 25 example, by not taking account of phase changes), in view of the small variations in the signal envelope permitted by an appropriate choice of parameters for filtering the accumulated phase and said real and imaginary components. The criteria for this selection will be specified further on.
The invention permits digital radio communication systems, in 5 particular professional radio communication systems, to be implemented in accordance with applicable standards relating to adjacent channel interference, and provides unequalled sensitivity and thus radio range, using power amplifier components which are readily available on the market and have a high power efficiency.
Another aspect of the invention relates to a digital modulator, including means for converting successive symbols of a digital stream into phase increments, a summator which accumulates the successive phase increments to produce an accumulated phase, a phase filter receiving the accumulated phase and producing a modulating phase, means for producing a 15 complex signal whose argument represents the modulating phase, and a modulator for respectively modulating two phase quadrature radio waveforms on the basis of said complex signal, and for transmitting a radio signal resulting from a combination of the two modulated waveforms, the modulator comprising a digital filter to which the complex signal is applied, analog-digital 20 converters respectively processing the digital signals obtained from the real and imaginary components of the digitally filtered complex signal, anti-aliasing analog filters receiving the output signals from the analog-digital converters, and two mixers each receiving one of the two radio waveforms and the output signal from one of the two anti-aliasing filters.
Other features and advantages of the invention will become apparent from the following description of non-limiting embodiments, with reference to the attached drawings where:
- Figures 1 and 2 are block diagrams of a digital modulator in accordance with the invention and an associated receiver respectively; and - Figures 3 and 4 are graphs respectively showing the constellation and spectrum of a digital modulator according to the invention.
The modulator shown in Figure 1 comprises a unit 10 which converts the successive symbols aj of a digital stream into phase increments 10 ~. The successive phase increments ~ produced by the unit 10 are accumulated by a summator 11. The unit 10 may merely consist of a register containing the possible values of the phase increments ~ and addressed by the current value of the symbol aj In the embodiment of Figure 1, the symbol stream aj may correspond either to a bit stream bj or another bit stream c; having with a lower transmission rate, processed by a redundancy encoder 12. The bit streams b and c; derive from digital sources such as speech encoders, data sources, etc., generally with error correction coding applied. If the encoder 12 is used, the modulator of Figure 1 operates in accordance with a coded modulation 20 (see G. Ungerboeck "Channel coding with multi-level/phase signals", IEEE
Trans. on Information Theory, Vol. IT-28, No. 1, January 1982, pages 55-67).
The unit 10, summator 11 and encoder 12 are timed by a clock signal CKS at the frequency 1/Ts of the symbols aj.
The summator 11 stores a integer digital value k representing an accumulated phase. This accumulated phase is thus stored as whole multiples of a sub-multiple of 7~, that is to say in the general form (k/P)~. For each cycle of the clock CKS, the accumulated phase is incremented by a value ~ depending on the current symbol aj. If each symbol represents m bits, M = 2m different values of the increment may be added in each cycle.
These M values are chosen so that the set of the increments is symmetrical with respect to the value 0 so that the spectrum is symmetrical. Values of k of the type k = k' x K will typically be used, where K/P represents the modulation index, and k' = -M+1, -M+3, ..., -1, 1, ..., M-3 or M-1. This choice of equally 0 distributed increments is not the only one possible. For example, k' = -7, -3, 3 or 7 could also be used if m=2. ~maX = (kmaX/P)~ designates the maximum value of the phase increment ~.
The accumulated phase is fed to a digital filter 15, referred to as phase filter, whose sampling frequency 1/Te, set by a clock signal CKE, is higher than the frequency 1/TS of the symbols aj (generally a multiple of that frequency).
The output signal from the phase filter 15 is a modulating phase ~, which a unit 16 converts into a complex signal, i.e. into two real signals, one (I) representing the real component of the complex signal, and the other (Q) representing the imaginary component.
That complex signal has a constant modulus, and an argument equil to the modulating phase ~. In other words, I = cos~ and Q = sin~. The unit 16 may merely consist of two read-only memory arrays addressed by the output of the filter 15 at every cycle of clock CKE.
The complex signal is filtered by a digital filter which, in the embodiment shown, consists of two identical filters 17 which respectively filter the components I and Q.
Two digital-analog converters 18 convert the output signals of the 5 two filters 17 into analog form. The two resulting analog signals are fed to low-pass filters 19 in order to eliminate spectral aliasing components. Using respective mixers 21, two quadrature radio waveforms at the carrier frequency, deriving from a local oscillator 20, are mixed with the signals deriving from the anti-aliasing filters 19. The two waveforms thus modulated are combined by a 10 summator 22 whose output is fed to the power amplifier 23 of the transmitter.
If the amplifier 23 were linearised by pre-distortion, it would be necessary to correct the filtered components I and Q, between the filters 17 and converters 18, before converting them into analog form.
The receiver shown in Figure 2 includes a low-noise amplifier 30 which amplifies the signal picked up by the antenna. Its output is converted to an intermediate frequency using a mixer 31. A band-pass filter 32 processes the intermediate frequency signal which is then amplified further by an amplifier 33. Two other mixers 34 provide baseband conversion by mixing with two quadrature waveforms. The two quadrature analog components 20 deriving from the mixers 34 are fed to identical low-pass filters 35, then converted into digital form by analog-digital converters 36. The digital components 1' and Q' deriving from the converters 36 are fed to a channel demodulator 37.
The demodulator 37 carries out demodulating operations 25 corresponding to the incomplete modulator consisting of the components 10, CA 02244609 l998-08-04 11, 15, 16 and (if necessary) 12 of the transmitter shown in Figure 1. Since this incomplete modulator essentially performs continuous phase modulation (CPM), the demodulator 37 may take the form of a conventional CPM
demodulator. It may, for example, be based on a demodulation trellis in order 5 to apply the Viterbi algorithm. The demodulator 37 delivers estimates b j or c of the bits bj or cj fed to the modulator.
Advantageously, the demodulator 37 may include two trellis. It uses either one of the trellis, depending on whether the encoder 12 is used at the transmitter or not. The first trellis includes modulation states. In principle, the number of these states is ML-1 x P, where L is the memory of the phase filter 15 expressed in number of samples, M is the number of points on the constellation, and P is the denominator of the modulation index. However, it is generally possible to considerably reduce the number of states of the demodulation trellis without adversely affecting the quality of reception significantly. The second one of the trellis further includes the coding states of the redundancy encoder 12, in accordance with the principle of coded modulations. This second trellis is employed if the encoder 12 is used at the transmitter.
In the modulator design, the values of the phase increments ~
20 are first chosen, as indicated hereabove. The filter 17 which processes the components I and Q and determines the spectral characteristics of the resulting signal is then constructed. The characteristics of this filter 17 must be as close as possible to those of the receiving filter consisting of the combination of filters 32 and 35.
CA 02244609 l998-08-04 _ 9 _ A advantageous form of the digital filters 17, used to process the components I and Q, is a filter with a finite impulse response selected to most closely fit a time characteristic of the form:
f(t) = Sinc(at/Ts).Sinc(,~t/Ts).e~ rvTs) (1) 5 where Ts is the duration of a symbol aj, and Sinc() is the cardinal sine function.
The approximation can be made by choosing the real coefficients cc, ~ and ~.
This provides digital filters whose restriction to a finite length is as accurate as possible by virtue of the fast decay of the Gaussian function. The secondary lobes caused by the limitation of the digital filter length are thus minimised.
The following step consists in defining the phase filter 15. The characteristics of this filter 15 are closely related to those of the digital filter 17.
A heuristic method is given hereunder based on the following mathematical property: the energy of a complex function eim(t) with unitary modulus is maximum in a filter whose spectral power template is the Fourier transform of a function h(t) (in other words, that energy is minimum outside the filter) if it satisfies the following equatio,n:
J h(u-t).eim(t)dt = ~,(u) eim(u) where ~(u) is a real function.
The following algorithm is used to define the phase filter 15:
1) A power template filter is selected, i.e. a function h(t) whose Fourier transform represents the required spectral template. A filter identical to the one selected as the l-Q filter is typically chosen. Other choices are obviously possible. In general, it is preferable to use a filter whose digital implementation with a fairly short finite impulse response is possible.
2) A function q)0 equal to 0 where t<0, equal to ~maXt/Ts where 0<t<Ts and equal to ~maX where t>Ts is used as a first approximation of the phase change function, i.e. of the function available at the output of the phase filter when the maximum phase increment ~maX is fed to the accumulator 11.
5 Other approximations using continuous functions equal to 0 where t<0 and ~max where t>Ts could be used.
3) A function (~n is calculated iteratively using the following formula:
¦ h(u-t) q)n (t) dt ¦ ¦ h(u-t). ~n (t) dt¦
¦ h(u-t) q)n (t) dt ¦ ¦ h(u-t). ~n (t) dt¦
4) The nth approximation of the impulse response of the phase filter, which is equal to the derivative of the function ~n~ is calculated. An approximation of this derived function can also be made using an approximate analytical formula in order to facilitate subsequent calculations. The analytical formula may be as follows:
g(t) = Sinc(a't/Ts).Sinc(~'t/Ts).e~(~l~tlTs) (2) where a', ~' and y' are real coefficients.
g(t) = Sinc(a't/Ts).Sinc(~'t/Ts).e~(~l~tlTs) (2) where a', ~' and y' are real coefficients.
5) For one of these approximations (for example n = 2 or 3), the characteristics of the modulation are evaluated with respect to the criteria of interference power in adjacent frequency channel, variation in amplitude and 20 noise immunity. If the approximation is unsuitable, the calculations 1) to 4) are repeated by modifying the values of the phase increments, and/or by modifying the shape of the l-Q filter 17, and/or by modifying the shape of the filter referred to under 1), and/or by modifying the approximation of the phase filter obtained using the algorithm.
The retained phase filter is then implemented as a finite impulse response digital filter.
The phase filter 15 of the modulator of Figure 1 could be replaced by a bank of phase filters selected in accordance with the origin of the symbol stream aj. This could provide a phase filter 15 optimised for cases where the redundancy encoder 12 is not used, and another one optimised for cases where the encoder 12 is used.
In a particular embodiment of the invention, the duration Ts of a symbol aj is 125 ,us. The number of bits per symbol is 2, the phase increments being ~ 13, ~13 or ~. The bit rate is then 16 kbiVs. The spectral specifications are those of the ETSI standard 300-113. The finite impulse response of the phase filter 15 has a length of 4 symbols and the form (2) with a' = 0.77, ,B' = 0.5 and ~' = 0. The finite impulse response of the l-Q filter 17 has a length of 8 symbols and the form (1) with a = 1.6, ~ = 0.1 and ~ = 0.12.
The values specified for the filter parameters could be replaced by values of the same order.
The constellation corresponding to that modulation is shown in 20 Figure 3. A very small variation in amplitude will be noted, since the ratio between the maximum instantaneous power and the average power is only 1.2 dB, whereas the ratio between the maximum and minimum instantaneous powers is less than 2.4 dB. Because of these characteristics, the modulation can be used with weekly linearised power amplifiers, easy to adjust and having an efficiency very close to that of saturated amplifiers.
The spectrum is shown in Figure 4. It can be noted that the level of adjacent channel interference is very low and compatible with the most demanding standards.
The noise immunity performance is excellent since, with a channel affected by Gaussian white noise, an error rate of 1% is noted for a signal-to-noise ratio Eb/No of 5.5 dB in the case of stationary stations, whereas the same error rate of 1% is obtained for a signal-to-noise ratio Eb/No of 16 dB
in a dynamic case (speed of 70 km/h and carrier at 400 MHz). These error 0 rates are obtained using simple conventional demodulators (Figure 2), i.e.
trellis demodulators with a very small number of states. A trellis with only three states can be used in the example shown.
In the same embodiment of the invention, the optional redundancy encoder 12 permits the implementation of a coded modulation.
Here, the redundancy coding is a convolutional coding of rate 1/2, the bit rate then being 8 kbiVs. The filter values are identical, and a trellis with only four states may be used at the demodulator. The coding gain is of the order of 2.5 dB, and an error rate of 1% is noted with Eb/No = 3.4 dB on a channel affected by Gaussian white noise in a stationary case.
The retained phase filter is then implemented as a finite impulse response digital filter.
The phase filter 15 of the modulator of Figure 1 could be replaced by a bank of phase filters selected in accordance with the origin of the symbol stream aj. This could provide a phase filter 15 optimised for cases where the redundancy encoder 12 is not used, and another one optimised for cases where the encoder 12 is used.
In a particular embodiment of the invention, the duration Ts of a symbol aj is 125 ,us. The number of bits per symbol is 2, the phase increments being ~ 13, ~13 or ~. The bit rate is then 16 kbiVs. The spectral specifications are those of the ETSI standard 300-113. The finite impulse response of the phase filter 15 has a length of 4 symbols and the form (2) with a' = 0.77, ,B' = 0.5 and ~' = 0. The finite impulse response of the l-Q filter 17 has a length of 8 symbols and the form (1) with a = 1.6, ~ = 0.1 and ~ = 0.12.
The values specified for the filter parameters could be replaced by values of the same order.
The constellation corresponding to that modulation is shown in 20 Figure 3. A very small variation in amplitude will be noted, since the ratio between the maximum instantaneous power and the average power is only 1.2 dB, whereas the ratio between the maximum and minimum instantaneous powers is less than 2.4 dB. Because of these characteristics, the modulation can be used with weekly linearised power amplifiers, easy to adjust and having an efficiency very close to that of saturated amplifiers.
The spectrum is shown in Figure 4. It can be noted that the level of adjacent channel interference is very low and compatible with the most demanding standards.
The noise immunity performance is excellent since, with a channel affected by Gaussian white noise, an error rate of 1% is noted for a signal-to-noise ratio Eb/No of 5.5 dB in the case of stationary stations, whereas the same error rate of 1% is obtained for a signal-to-noise ratio Eb/No of 16 dB
in a dynamic case (speed of 70 km/h and carrier at 400 MHz). These error 0 rates are obtained using simple conventional demodulators (Figure 2), i.e.
trellis demodulators with a very small number of states. A trellis with only three states can be used in the example shown.
In the same embodiment of the invention, the optional redundancy encoder 12 permits the implementation of a coded modulation.
Here, the redundancy coding is a convolutional coding of rate 1/2, the bit rate then being 8 kbiVs. The filter values are identical, and a trellis with only four states may be used at the demodulator. The coding gain is of the order of 2.5 dB, and an error rate of 1% is noted with Eb/No = 3.4 dB on a channel affected by Gaussian white noise in a stationary case.
Claims (9)
1. A digital modulation process wherein the successive symbols (a j) of a digital stream are converted into phase increments (.DELTA.~), an accumulated phase is obtained by adding the successive phase increments, a modulating phase (~) is obtained by filtering the accumulated phase, a complex signal is produced whose argument represents the modulating phase, two phase quadrature radio waveforms are respectively modulated on the basis of said complex signal, and a radio signal resulting from a combination of the two modulated waveforms is transmitted, characterised in that said complex signal is digitally filtered and in that the digital signals obtained from the real and imaginary components (I,Q) of the digitally filtered complex signal are converted into analog form before being respectively subjected to anti-aliasing analog filtering and then mixed with the two radio waveforms.
2. A process in accordance with claim 1, wherein the digital filtering of said complex signal consists of two identical filtering operations on the real and imaginary components thereof (I,Q).
3. A process in accordance with claim 2, wherein the digital filtering of the real or imaginary component of the complex signal has a finite impulse response corresponding to a time characteristic having the form:
f(t) = Sinc(.alpha.t/T s).Sinc(.beta.t/T s).e-(.pi..gamma.t/Ts)2, where T s is the duration of a symbol of the bit stream, .alpha., .beta. and .gamma. are real coefficients, and Sinc() is the cardinal sine function.
f(t) = Sinc(.alpha.t/T s).Sinc(.beta.t/T s).e-(.pi..gamma.t/Ts)2, where T s is the duration of a symbol of the bit stream, .alpha., .beta. and .gamma. are real coefficients, and Sinc() is the cardinal sine function.
4. A process in accordance with claim 3, wherein the filtering of the accumulated phase has a finite impulse response corresponding to a time characteristic having the form:
g(t) = Sinc(.alpha.'t/T s).Sinc(.beta.'t/T s).e-(.pi..gamma.'t/Ts)2, where .alpha.', .beta.' and .gamma.' are real coefficients.
g(t) = Sinc(.alpha.'t/T s).Sinc(.beta.'t/T s).e-(.pi..gamma.'t/Ts)2, where .alpha.', .beta.' and .gamma.' are real coefficients.
5. A digital modulator comprising: means (10) for converting the successive symbols (a j) of a digital stream into phase increments (.DELTA.~), a summator (11) which accumulates the successive phase increments to produce an accumulated phase, a phase filter (15) receiving the accumulated phase and producing a modulating phase (~), means (16) for producing a complex signal whose argument represents the modulating phase, and a modulator for respectively modulating two phase quadrature radio waveforms on the basis of said complex signal, and for transmitting a radio signal resulting from a combination of the two modulated waveforms, characterised in that the modulator comprises a digital filter (17) to which said complex signal is applied, analog-digital converters (18) respectively processing the digital signals obtained from the real and imaginary components (I,Q) of the digitally filtered complex signal, anti-aliasing analog filters (19) receiving the output signals from the analog-digital converters, and two mixers (21), each receiving one of the two radio waveforms and the output signal from one of the two anti-aliasing filters.
6. A modulator in accordance with claim 5, wherein the digital filter, to which said complex signal is applied consists of two identical filters (17) receiving respectively the real and imaginary components (I,Q) of the complex signal.
7. A modulator in accordance with claim 6, wherein the digital filter (17) processing the complex signal has a finite impulse response corresponding to a time characteristic having the form:
f(t) = Sinc(.alpha.t/T s).Sinc(.beta.t/T s).e-(.pi..gamma.t/T s)2, where T s is the duration of a symbol a i of the bit stream, .alpha., .beta. and .gamma. are real coefficients, and Sinc() is the cardinal sine function.
f(t) = Sinc(.alpha.t/T s).Sinc(.beta.t/T s).e-(.pi..gamma.t/T s)2, where T s is the duration of a symbol a i of the bit stream, .alpha., .beta. and .gamma. are real coefficients, and Sinc() is the cardinal sine function.
8. A modulator in accordance with claim 7, wherein the phase filter (15) has a finite impulse response corresponding to a time characteristic having the form:
g(t) = Sinc(.alpha.'t/T s).Sinc(.beta.'t/T s).e-(.pi..gamma.'t/T s)2, where .alpha.'.beta.' and .gamma.' are real coefficients.
g(t) = Sinc(.alpha.'t/T s).Sinc(.beta.'t/T s).e-(.pi..gamma.'t/T s)2, where .alpha.'.beta.' and .gamma.' are real coefficients.
9. A modulator in accordance with claim 8, wherein T s = 125µs, each symbol (a i) of the bit stream consists of two bits, the phase increments (.DELTA..PHI.) are - .pi., - .pi./3, .pi./3 or .pi., and .alpha. ~ 1.6, .beta. ~ 0 1, .gamma. ~ 0.12, .alpha.' ~ 0.77, .beta.' ~ 0-5 and .gamma.'~0.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| FR9709962A FR2767002B1 (en) | 1997-08-04 | 1997-08-04 | DIGITAL MODULATION METHOD, AND MODULATOR IMPLEMENTING SUCH A METHOD |
| FRFR9709962 | 1997-08-04 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2244609A1 true CA2244609A1 (en) | 1999-02-04 |
Family
ID=9510003
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002244609A Abandoned CA2244609A1 (en) | 1997-08-04 | 1998-08-04 | Digital modulation process, and modulator implementing the process |
Country Status (8)
| Country | Link |
|---|---|
| EP (2) | EP0936784B1 (en) |
| AT (2) | ATE265778T1 (en) |
| CA (1) | CA2244609A1 (en) |
| DE (2) | DE69800349T2 (en) |
| ES (1) | ES2152749T3 (en) |
| FR (1) | FR2767002B1 (en) |
| GR (1) | GR3035210T3 (en) |
| SG (1) | SG71825A1 (en) |
Family Cites Families (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2968350B2 (en) * | 1991-01-11 | 1999-10-25 | 三菱電機株式会社 | Quadrature modulation circuit |
| FR2675001B1 (en) * | 1991-04-03 | 1993-07-30 | Matra Communication | METHOD AND DEVICE FOR DIGITAL MODULATION WITH PHASE AND QUADRATURE COMPONENTS AND TRANSMISSION INSTALLATION COMPRISING AN APPLICATION. |
| US5642384A (en) * | 1993-07-06 | 1997-06-24 | Ericsson Inc. | Trellis coded modulation scheme with low envelope variation for mobile radio by constraining a maximum modulus of a differential phase angle |
| US5600676A (en) * | 1993-07-06 | 1997-02-04 | Ericsson Ge Mobile Communications Inc. | Modulation scheme with low envelope variation for mobile radio by constraining a maximum modulus of a differential phase angle |
-
1997
- 1997-08-04 FR FR9709962A patent/FR2767002B1/en not_active Expired - Fee Related
-
1998
- 1998-08-03 AT AT98401972T patent/ATE265778T1/en not_active IP Right Cessation
- 1998-08-03 EP EP99102638A patent/EP0936784B1/en not_active Expired - Lifetime
- 1998-08-03 EP EP98401972A patent/EP0896456B1/en not_active Expired - Lifetime
- 1998-08-03 AT AT99102638T patent/ATE196961T1/en not_active IP Right Cessation
- 1998-08-03 DE DE69800349T patent/DE69800349T2/en not_active Expired - Fee Related
- 1998-08-03 DE DE69823429T patent/DE69823429T2/en not_active Expired - Fee Related
- 1998-08-03 ES ES99102638T patent/ES2152749T3/en not_active Expired - Lifetime
- 1998-08-04 CA CA002244609A patent/CA2244609A1/en not_active Abandoned
- 1998-08-04 SG SG1998002761A patent/SG71825A1/en unknown
-
2001
- 2001-01-11 GR GR20010400024T patent/GR3035210T3/en not_active IP Right Cessation
Also Published As
| Publication number | Publication date |
|---|---|
| EP0936784A2 (en) | 1999-08-18 |
| EP0896456A1 (en) | 1999-02-10 |
| EP0936784A3 (en) | 1999-10-06 |
| ATE196961T1 (en) | 2000-10-15 |
| ATE265778T1 (en) | 2004-05-15 |
| EP0936784B1 (en) | 2000-10-11 |
| DE69823429D1 (en) | 2004-06-03 |
| DE69800349T2 (en) | 2001-05-17 |
| SG71825A1 (en) | 2000-04-18 |
| FR2767002A1 (en) | 1999-02-05 |
| GR3035210T3 (en) | 2001-04-30 |
| DE69800349D1 (en) | 2000-11-16 |
| DE69823429T2 (en) | 2005-02-24 |
| ES2152749T3 (en) | 2001-02-01 |
| FR2767002B1 (en) | 1999-10-08 |
| EP0896456B1 (en) | 2004-04-28 |
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| Date | Code | Title | Description |
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| EEER | Examination request | ||
| FZDE | Discontinued |