CA2160840C - Surround sound processor with improved control voltage generator - Google Patents

Abstract

A surround sound processor for presenting stereophonic audio signals on a number of loudspeakers surrounding a listening area, comprising an input matrix stage, a detector filter and matrix circuit, a direction detector circuit incorporating improved filters, a novel detector splitter circuit providing three direction signals from the two direction detector circuit outputs, a novel three-channel servologic circuit with improved variable filters responding selectively to the rates of change of the direction signals and providing six control voltage signals through linearity correction networks to six voltage-controlled amplifiers, the outputs of which are combined in an output matrix to provide a number of loudspeaker feed signals through buffer amplifiers to the output terminals of the processor.
The detector splitter circuit is configurable to either a forward oriented or a backward oriented mode of operation, corresponding changes being made to the voltage-controlled amplifiers and output matrix circuitry. Additionally the detector splitter may be switched to eliminate the third direction signal with corresponding changes to the output matrix circuitry.

Description

21 6 0 8 4~ 0 SURROUND SOUND PROCESSOR WITH
IMPROVED CONTROL VOLTAGE GENERATOR
Cross Reference to Related Patent This application is related to U. S. Patent No. 5,172,415, issued on December 15, 1992.
Technical held The present invention relates in general to processors for the periphonic reproduction of sound. More specifically, the invention relates to improvements in the servologic control voltage generator of a surround sound processor for multichannel redistribution of audio signals.
Background Of The Invention This invention relates to improvements in a surround sound processor.
A surround sound processor operates to enhance a two-channel stereophonic source signal so as to drive a multiplicity of loudspeakers arranged to surround the listener, in a manner to provide a high-definition soundfield directly comparable to discrete multitrack sources in perceived performance. An B

216()840 Docket No. 13096.20 illusion of space may thus be created enalbling the listener to experience the fullness, directional quality and aural dimension or "spaciousness" of the original sound environment. The foregoing so-called periphonic reproduction of sound can be distinguished from the operation of conventional soundfield processors which rely on digitally generated time delay of audio signals to simulate reverberation or "ambience" associated with live sound events. These conventional systems do not directionally localize sounds based on information from the original performance space and the resulting reverberation characteristics are noticeably artificial.
To accomplish this end, a surround sound processor typically comprises an input matrix, a control voltage generator and a variable matrix circuit.
The input matrix usually provides for balancf~ and level control of the input signals, generates normal and inverted polarity versions of the input signals, plus sum and difference signals, and in some case.. generates phase-shifted versions, and/or filters the signals into multiple frequency ranges as needed by the remainder of the processing requirements.
The control voltage generator includes a directional detector and a servologic circuit. The directional detector measures the correlations between the signals which represent sounds encoded at different directions in the stereophonic sound stage, generating voltages corresponding to the predominant sound directional location. The servologic circuit uses these signals to develop control voltages for varying the gain of voltage controlled amplifiers in the variable matrix circuit in accordance with the sound direction and the direction in which it is intended to reproduce the sound in the surrounding loudspeakers.
The variable matrix circuit includes voltage-controlled amplifiers and a separation matrix. The voltage-controllE~d amplifiers amplify the input matrix audio signals with variable gain, for application to the separation matrix, where they are used to selectively cancel crosstalk into different loudspeaker feed signals. The separation matrix combines the outputs of the input matrix -.3-Docket No. 13096.20 and of the voltage-controlled amplifiers in several different ways, each resulting in a loudspeaker feed signal, for a loudspeaker to be positioned in one .
of several different locations surrounding the listener. In each of these signals, certain signal components may be dynamiically eliminated by the action of the detector, control voltage generator, voltage-controlled amplifiers (VCA's) and separation matrix.
In surround sound processors, much of the subtleties of the presentation are due to the characteristics of the direction detector and servologic circuit of the control voltage generator and of the 'CA's. As these are further refined, the apparent performance becomes more transparent and effortless-sounding to the listener.
Summary of the Invention The present invention provides an improved surround processor for the reproduction of sound from a stereophonic source in a manner comparable to a live presentation from multiple sources in perceived performances. The present invention relates in particular to improvE;ments in the implementation of the circuitry of a servologic control voltage g<~nerator, employing multiple-axis control voltage signals.
In a departure from the art, the processor includes a direction detector circuit incorporating improved filters, a detector splitter circuit providing three direction signals from the two direction detector circuit outputs, and a three-channel servologic circuit with improved variable filters responding selectively to the rates of change of the direction signals and providing six control voltage signals, through linearity correction netvvorks, to six voltage-controlled amplifiers, the outputs of which are combined in an output matrix to provide a number of loudspeaker feed signals through buffer amplifiers to the output terminals of the processor. The detector splitter circuit is configurable to either a forward oriented or a backward oriented mode of operation, corresponding changes being made to the voltage-controlled amplifiers and output matrix _<<_ Docket No. 13096.20 circuitry. Additionally the detector splitter may be switched to eliminate the third direction signal with corresponding changes to the output matrix circuitry.
In a preferred embodiment, a surround sound processor is provided for receiving left and right audio signals of a stereophonic source and for processing the left and right signals for presentation on a plurality of loudspeakers surrounding a listening area so as to produce an impression of discrete sound sources surrounding a listener therein. The processor includes a pair of input terminals for receiving left and right audio signals; an input matrix circuit which receives the left and right audio signals from the pair of input terminals and comprises at least a~n inverting amplifier and a non-inverting amplifier of equal gain for each. channel to provide the left and right audio signals in either polarity to succeeding circuits; a detector filter which receives the left and right audio signals i:rom the input matrix circuit and filters each of them with a suitable transfer characteristic to provide left and right filtered signals; a detector matrix circuit which receives and combines the left and right filtered signals to provide i;heir sum and difference as additional front and back filtered signals respectively; a direction detector circuit which receives the left and right filtered signals and processes them to provide a left-right direction signal proportional to the logarithm of the ratio of the amplitude of the right filtered signal to that of the left filtered signal, up to a limiting voltage, and which also receives the front and back filtered signals and processes them to provide a front-back direction signal proportional to the logarithm of the ratio of the amplitude of the back filtered signal to the front filtered signal, up to the same limiting voltage; a detector sputter circuit which receives the left-right and front-back direction signals and processes them to provide one or more additional direction signals while modifying the left-right and front-back direction signals to maintain constancy of the sum of all the said direction signals; a servologic circuit which receives the modified left-right and front-back direction signals and the one or more additional direction -y_ _ ~I ~Q84 ~
Docket No. 13096.20 signals from the detector splitter circuil;, filters each of them with a variable low-pass filter, inverts each resulting signal and half wave rectifies each signal and its inversion to provide a plurality of smoothly varying control voltage signals of the same polarity; a plurality of voltage-controlled amplifiers equal to the plurality of the control voltage sign~~ls, a different one of the control voltage signals being connected to control the gain of each of the voltage-controlled amplifier, each of the plurality of voltage-controlled amplifiers receiving different proportions of the left or right signals or their inversions from the input matrix circuit, and summing the received signals with a variable gain dependent upon the corresponding one of said plurality of control voltage signals applied thereto; an output matrix circuit comprising a plurality of matrix circuits equal to said plurality of loudspeakers, each matrix circuit receiving one or more of the left and right signals and their inversions from the input matrix circuit and one or more oui:put signals from said plurality of voltage-controlled amplifiers, and combining them in suitable proportions to obtain a loudspeaker feed signal wherein unwanted directional components are canceled; a plurality of output terminals equal to the plurality of loudspeakers;
and a plurality of output buffers equal to the plurality of loudspeakers, each output buffer receiving one of the plurality of loudspeaker feed signals from the output matrix circuit and amplifying it to an appropriate level for driving a power amplifier connected to one said output terminal of the processor, for driving the corresponding one of said plurality of loudspeakers surrounding said listening area.
An advantage achieved with the irwention is that the surround processor provides faster but smoother and more realistic multichannel sound redistribution from a stereophonic source..
_5_ Docket No. 13096.20 Brief Description of the Drawings FIG. 1 is a diagrammatic representation of an energy sphere;
FIG. 2 is a block diagram which illustrates a typical surround sound processor involving the present invention;
FIG. 3 is a detailed schematic of the input stage of a surround sound processor according to the prior art and suitable for use with the present invention;
FIG. 4 is a detailed schematic of a detector filter and matrix according to the present invention;
FIG. 5 is a detailed schematic of a~n improved log-ratio detector according to the present invention;
FIG. 6 is a detailed schematic of a. detector splitter circuit according to the invention;
FIG. 7 is a graphical representation of the outputs of the detector sputter circuit of FIG. 6;
FIG. 8 is a detailed schematic of a three-axis servologic circuit according to the invention;
FIG. 9 is a detailed schematic of an improved VCA according to the invention;
FIGs. l0a and lOb are a block schematic of a first output matrix configuration according to the invention; and FIGs. 11a and llb are a block schematic of a second output matrix configuration according to the invention.
Detailed Description of the Invention The principal new featured of the present invention are the inclusion of a detector splitter circuit, permitting six different control voltage signals to be developed, each of which corresponds to a. different sound direction in the surround sound field, and the provision oaf two different modes of operation in the detector splitter circuit aired the VCA',; of the variable matrix circuit.

.. 2f 6Q840 Docket No. 13096.20 To understand these more fully, t;he preferred embodiments of the invention will be described in detail below. However, the principles on which the present invention are based will fire,t be reviewed.
In FIG. 1 is shown a conventionall representation of the amplitude and phase relationships between the left anti right channels of a stereophonic signal pair, called the Scheiber sphere (;after Peter Scheiber) or energy sphere (this term is due to M. A. Gerzon), which relates the relative amplitudes, or amplitude ratio, of the left and right channels to an angle O and the phase angle between the signals to another angle ~. These correspond to spherical polar coordinates of a sphere in which the polar axis runs ftom left to right in the horizontal plane, and the "longitude"' represents the ratio betyveen right and left signals, while the "latitude" represents the phase angle.
Specifically, for sinusoidal signals l, r, of any frequency, and having amplitudes L, R
respectively, R/I. = tan (O/2) (1) and r lags L by the phase angle ~. Thus the pair of signals L and r of frequency f can be represented by sinusoidal components:
L = E cos (O/2) cos (wt) (2) r = E sin (O/2) cos (wt - ~) (3) where L = E cos (O/2), R = E sin (~9/2), and w = 2nf.
In FIG. 1, this signal is represented by the vector OE, which lies on the plane LPRQ and makes an angle of O with the axis OR. The plane LPRQ
intersects the equatorial plane FUBD in the line PQ passing through the center O of the sphere, which makes an ;angle ~ with the horizontal plane LFRB.
It can be shown that the locus of t:he unit vectors of all possible values of ~ and O is a sphere, having a polar coordinate system described by the coordinates (O, ~), with one pole at the left or L point and one at the right or R
point. A conventional view of this sphere is one having the left pole at the left, the right pole at the right, and all in-phase or antiphase vectors lying in a ~~160840 Docket No. 13096.20 horizontal plane through the L and R poles. The line joining these poles is the left-right (or L-R) axis.
Conventionally, as in FIG. 1, this sphere is viewed from a point above the horizontal plane, in the antiphase half, and to the right of center, so that all three primary axes of the sphere, L-lft, B-F and D-U are seen in correspondence with the conventional z, x, and y Cartesian axes respectively, since the polar axis, usually called z, is horizontal in this figure. The equatorial circle of this sphere, FUBD, which is the locus of all vectors which have equal left and right signal amplitudes as the phase difference ~ between them varies, is in a vertical plane and tlhe viewpoint is to the right of this plane. The equatorial plane intersects the horizontal plane LFRB in a line which is termed the front-back (or F-B) axis.
Consider a stereophonic center front signal in which the left and right amplitudes are equal and the signals are in phase. The location of this signal on the Scheiber sphere is F, on the equator midway between the L and R poles, and is also in the horizontal plane, conventionally drawn away from the viewing direction at the "front" or F location on the sphere. The "back" or B
location is diametrically opposite this "front" location, F, so that the ftont-back or F-B axis, the line j oining these two points, is orthogonal to the left-right or L-R axis. This back location represents .a signal pair in which L and R
signals are equal in amplitude but of opposite polarity, i.e. in antiphase, or 180° out of phase.
In some multichannel signal processors, the third orthogonal axis extends from a point on the sphere directly below its center, the "down"
point, D, to the corresponding point directly above the sphere, or "up" point, U, thus forming a third orthogonal axis called the up-down or U-D axis. The "up" point represents a signal pair of equal amplitudes but having a quadrature phase relationship, in which the left signal leads the right by 90° of phase;
similarly at the "down" point, the left signal lags t:he right by 90°.
_g_ 21fi084~
Docket No. 13096.20 In order to generate the signal components corresponding to these directions, it is necessary to use wide-band all-pass quadrature phaase difference networks, which tend to be e~;pensive as they require high precision components. Therefore, most surround sound processors today do not employ the up-down axis either for detection or matrixing purposes, although the majority of older quadraphonic encoding and decoding formats, such as Sf~,l, QS, BMX and BHJ (Ambisonics), used such :phase quadrature networks in varying ways.
One tenet of the Ambisonics theory of Gerzon et ail. which has been largely followed in modern surround sound processors is that consistent encoding may be obtained if the locus of sound directions encoded around the listener in the horizontal plane forms a great circle on the energy sphere. In the Ambisonics or BHJ matrix, this plane is tilted and somewhat distorted to better approximate the optimum performance related to human hearing;
however, good results can also be obtained by using the horizontal plane only of the energy sphere, which results in economy by removing the need for quadrature phase networks. The original Scheiber matrix used such a pan locus, and as Scheiber first used the energy sphere representation (originally due to Poincare) in the context of quadraphony, it became known as the Scheiber sphere.
Although prior art surround soundl processors and those of the present inventor's previous patents employ detectors capable of generating control voltage signals corresponding to the front-back and left-right axis components of the signal, the present invention generates control voltage signals corresponding to the left front-right front (LF-RF) and left back-right back (LB-RB) nixes, shown in FIG. 1 on the horizor,~tal plane forward and rearward of the L-R axis. In fact, the LF and RF points are not collinear with the center O of the sphere, so that generation of detector signals based directly on these is not possible. One method by which such signals can be generated is to "boost" the sphere by means of mixing suitable proportions of left and right signals ._. 24i~0840 Docket No. 13096.20 together (see "Transformations of the Energy Sphere" by Martin Willcocks, JAES vol. 31 no. 1/2, 1983 January/February). This has the effect of swinging the lines O-LF and O-RF backwards until they lie along the L-R axis, also moving the O-L and O-R lines further back and the O-LB and O-RB lines still further back. An opposite boost direction can bring the O-LB and O-RB axes forward to the L-R axis.
To derive control signals equivalent to these axes, a method employed by the present invention employs summation means after the detectors for the L-R and F-B axes to generate detector outputs roughly corresponding to the boosted LF-RF and LB-RB axes.
Another aspect of the present invention concerns the signals that are fed to the various VCA's in order to obtain a stereo to surround synthesis. Two particular modes of operation have been found useful; the first, a forward emphasis mode, derives signals with a primarily frontal location between left front and right front, with surround sound information located toward the rear.
The second mode, a panoramic mode, spreads the front signals more around the listener and more towards the rear. To achieve this, the VCA's are fed with different combinations of the left and right signals, as will be explained further below with reference to FIGS. 9-lll.
Turning to FIG. 2, which is a block schematic of a surround sound processor according to the invention, the processor 1 is equipped with input terminals 2, 4, for receiving left (L) and aright (R) audio input signals respectively. These signals are processed by an input stage, 6, typically containing auto-balancing circuitry and other signal conditioning circuits, such as level controls and possibly a panorama control as described in other Fosgate patents or patent applications. A detailed schematic of a simple input stage is shown in FIG. 2, and will be described with reference thereto. The output signals from this stage are labeled LT and RT, and are applied via lines 5 to a detector filter 8, and via lines 3 to VCA's 18 - 28 and an output matrix 30.
Although not shown, to simplify the draining for improved clarity, the 29:60840 Docket No. 13096.20 inversions of these signals, -LT and -RT may be generated here and also provided via lines 3 to the VCA's 18-28 and output matrix 30.
The detector filter 8 provides filtered signals LTF and RTF labeled 7 to the inverter 9, the detector matrix circuit 10 and to a detector circuit 12.
The signal RTF is inverted by the inverter :~ and also applied to the detector matrix circuit 10. The detector matrix 10 generates outputs 11 labeled FTF and BKF
corresponding to front (L+R) and back (L-R) signal directions. These signals are also applied to detector circuit 12, which comprises two identical circuits.
One accepts input signals FTF and BKI~ and produces an output signal FB at 13, while the other accepts the input signals LTF and RTF to produce an output signal L/R, at 13.
The detector output signals 13 labeled FB and L/R, are applied to the novel detector splitter circuit 14, wherein are produced the three signals 15 labeled LF/R,F, FTBK and LB/R,B. These in turn are applied to the servo logic circuit 16 to provide six control voltage signals 17 labeled LFC, RFC, FTC, BKC, LBC and RBC, for controlling the six VCA's 18 through 28, and labeled LF, RF, FT, BK, LB, and RB VCA respE~ctively.
These VCA's receive the LT and Ii,T signals 3 in different proportions, according to the directional matrix they are intended to provide, and apply their output signals 19 through 29 each in both polarities to the output matrix 30, which also receives the unmodified hT and RT signals 3. As mentioned above, though not shown in FIG. 2, inverters may also be provided for these signals LT and RT to generate -LT and -RT respectively. These inverters may be considered to be a part of the input stage, as their outputs may also be applied to some inputs of VCA's 18 through 28. These details are shown in the accompanying FIGs. 3-11, as necessary i:or the understanding of the invention, but are not included in FIG. 2 in order to simplify the diagram and improve clarity.
Outputs from the matrix 30 are buffered by amplifiers 32 through 40, providing output signals LFO, CFO. RFO, LBO and RBO at terminals 42, 44, - l:l -~a.sos~o Docket No. 13096.20 46, 48 and 50 respectively. These form the five standard outputs of the processor 1, but other outputs (not shown) may also be provided. Typically, the outputs shown may be provided to eleci;ronic crossover components in order to provide subwoofer outputs L-SUB, R-St:JB and M-SUB (not shown in FIG. 2) as well as the five outputs shown. Such techniques are well known in the art and need no further explanation here.
Referring to FIG. 3, which shows a typical input stage, 6, suitable for use with the detector and matrix circuiitry of the present invention, this input stage comprises a left preamplifier 60 having alternate input jacks J101 and J 102, corresponding to terminal 2 of FIG. 2, for receiving the alternate input signals L2 and L1 respectively, a similar right preamplifier 62 having alternate input jacks J 103 and J 104, 4 in FIG. 2, for receiving the alternate input signals R2 and R1 respectively; left and right gain stages 64 and 66 respectively, left and right inverters 68 and 70 respectively, and an auto-balance circuit 72.
Also provided is some switching circuitry for modifying the characteristics of the processor to provide antiphase blending, as disclosed in Fosgate's prior patents and patent applications.
In FIG. 3, the left input signal L1. is passed from the input jack J102 through resistor 8104 in the left preamplifier circuit 60 into the left gain stage 64. The alternate left input signal L2 f~~om the input jack J101 goes into a shelf filter circuit forming part of the left input stage 60, comprising operational amplifier OA101 and the sw~rounding components, resistors R101-R105 and capacitors C 101 and C 102. Tlus filter provides a specific transfer characteristic which is not the subject of the present invention and will not be discussed further here. The output of the filter stage is applied via resistor 8106 to the input of the left gain stage fi4.
An identical input stage 62 is pro~~ided for the right channel input signals R 1 and R2, the right signal R 1 fi~om the input j ack J 104 being applied via resistor 8110 to the right gain stage 66, and the alternate right signal from the input jack J103 being applied to the filter stage comprising op-amp - 1~, -- Docket No. 13096.20 OA102, resistors 8107-8111, and capaci tors C 103 and C 104, the output of this filter being applied via resistor 8112 to 'the input of the right gain stage 66.
An alternative mode of operation ,allows the left and right channel input signals to be applied differentially to both pairs of terminals 2 and 4, thereby achieving a different filter characteristic. Optionally, the input circuit 6 may include high pass and low pass filter conaponents (not shown) for use in split-band applications.
The left gain stage 64 has a.c. ga.i:n defined by the feedback resistor 8118 in conjunction with resistor 8117 and the variable attenuator formed by resistor 8113 in conjunction with a junction field-effect transistor Q101, resistors 8114 and 8115, and capacitor C 105. When the gate voltage applied to Q101 is zero, the FET is in a low resistance state and about half the feedback current from resistor 8118 is bypassed through 8113 and Q101, so that the gain stage has a voltage gain of about 10 to either input signal L1 or the filtered input signal L2. The op-amp. OA103 has d.c. feedback provided by resistor 8119, and its inverting input is a.c. coupled via capacitor C 107.
Capacitor C108 provides a roll-off at frequencies well above the audio range.
As more negative voltages are applied to the gate of FET Q101 through the resistor 8115, its resistance rises, reducing the current shunted away from the feedback current and therefore reducing the gain to a minimum of about 5.
Capacitor C 105 and resistor 8114 provide negative feedback to the gate of Q101 of just the right magnitude to minimize the even-order distortion that otherwise occurs with this type of attenu;ator, because of the square-law gate transfer characteristic of junction FET's.
An identical gain stage comprising resistors 8122-8127, capacitors C 106, C 19 and C 110, op-amp OA104 and FET Q 102 provides the same function for the right channel input signals Rl or R2.
CMOS switches 5101 and S102 corulect resistor 8120 from the output of the right gain stage into the input of the left gain stage and resistor 8128 from the output of the left gain stage into the iinput of the right gain stage.
Since (2I608~0 Docket No. 13096.20 these gain stages are inverting, when switches S 101 and S 102 are on, antiphase cross-blending of the signals will occur. When switches S 101 and S 102 are off, resistors 8121 and 8129 ensure that a relatively small input voltage is applied to the CMOS switch. The input must not exceed the supply voltages for this chip which are typically t7.5V. The switches are normally off, their control terminals being pulled negative by resistor 8116 which goes to the -7.5V supply. They are turned on by applying +7.5V to the terminal 74 labeled BLEND.
The outputs of amplifiers OA103 and OA104 are connected respectively to the output terminals 76 and 80, and the signals present at these terminals are labeled LT and RT respectively. Each of these signals also passes to a unity gain inverter, the left inverter 68 comprising op-amp OA105 with resistors 8130 and 8131, and the right inverter 70 comprising op-amp OA106 and resistors 8132 and 8133. The outputs of these inverters are connected respectively to the terminals 78 and 82, labeled -LT and -RT. The signals LT, RT are also identified in FIG. 2 with the numerals 3 and 6, the signals 3 also including the inverted signals -LT and -RT, as stated in the discussion of FIG.

2 above.
The auto-balance circuitry 72 below these inverters receives its input signals 13 from the detector circuitry to be described below with reference to FIG. 5. It should be noted that this auto-balance circuitry is prior art and is not considered part of the present invention, but is included here for completeness as an element of the typic~~l input stage of a decoder according to the present invention.
The F/B signal is applied to terminal 84 to the inverting input of op-amp OA107, which is used as a voltage comparator. The non-inverting input of this op-amp is biased to a negative voltage b;y the resistors 8134 to the -15V
supply and 8135, taken to ground. The voltage at their junction is about -3.9V with the BLEND signal at terminal 74 left open or connected to -7.5V. When the BLEND signal is taken to +7.5V, a small current is applied to this junction via _ ly _ Docket No. 1309 6.20 the resistor 8136, changing the voltage to about -2.9V. Thus, whenever the FB signal goes more negative than the bias voltage applied to the non-inverting input of op-amp OA107, its output swings to the +15V supply rail, and the voltage at the control port of CnZOS switch 5103, provided by the resistive voltage divider comprising resistors 8137 and 8138, goes to about half this voltage or just under +7.5V, turnin~; this switch on. At other times, the comparator OA107 has a negative output and the voltage applied to switch S 103 is about -7.5V, turning it off.
The signal L~R represents the log-ratio between left and right signal amplitudes. It is applied via terminal 8~ to the non-inverting input of op-amp OA108, which is connected as a non-invE~rting amplifier, with feedback resistor 8139 and resistor 8140 from the inverting input to ground. Because this amplifier has a very high voltage gain, of about 150 to the junction of resistors 8141 and 8142, which form a voltage di~~ider to prevent the level at the input of the CMOS switch S103 from exceeding its t7.5V supply rail voltages, it responds to very small imbalances between the left and right signal amplitudes.
Thus, when the signal is predominantly ffontal, which typically occurs when dialog is present in a movie or a soloist is performing in a music recording, the switch S 103 is turned on, .and the imbalance signal at the output of op-amp OA108 is passed via S 103 to the resistor 8143 and capacitor C 111, which form an integrator with about a ten second time constant. The voltage at C 111 is buffered by the op-amp OA 109, connected as a non-inverting source follower. This voltage represents the long-term average imbalance due to slightly different left and right signal levels during such events as dialog and solo performances, etc. It is applied directly to the left channel attenuator formed by FET fa 101, with resistors 8113-8115 and capacitor C 105, and therefore reduces the left channel gain appropriately if the left channel signal predominates during these events. It is ~~lso inverted by op-amp OA110 with resistors 8144 and 8145, and applied to t;he right channel attenuator formed Docket No. 13096.20 by FET Q 102 with the associated resistors 8122-8124 and capacitor C 106, so that if the right channel signal predominates during such events, the right channel gain is reduced by an appropriate amount without affecting the left channel gain. The degree of balance correction that can be achieved is up to about 6dB in either direction.
Auto-balance circuits of this type are useful in providing for correction of improper balance, often due to the large numbers of stages that the stereophonic signals pass through prior to recording or transmission, if insufficient care is taken in the recording process, but can affect the correct performance when a soloist is deliberately recorded slightly to the right or left of center, such as in an orchestral concerto. Therefore, there is usually a provision (not shown in FIG. 3) to turn off the auto-balance circuit. The auto-balance circuit characteristics are different for the case when the BLEND
switch is on, because in this mode it has been found preferable to allow the auto-balance circuit to respond to input signals with a slightly wider range of imbalance.
Referring to FIG. 4, which shows a detailed schematic of a detector filter 8, inverter 9, and detector matrix 10, also according to prior art circuitry disclosed in previous Fosgate patents, input terminals 90 and 92 are provided for receiving the signals 5 labeled LT and RT respectively. These signals 5 are filtered by the first stage comprising operational amplifier OA301 and its associated components for the signal LT, and op-amp OA302 with its associated components for the signal RT. The outputs of this filter stage 8 are passed to the inverter 9 and to the detector matrix 10. The right channel filter output is inverted by inverter 9 which comprises o;p-amp OA303 with input resistor 8309 and feedback resistor 8310, with typical values shown. The output of op-amp OA301 is fed via resistor 8311 and capacitor C309 in series to output terminal 108, providing a filtered current signal LTF. The output of op-amp OA302 is fed via resistor 8316 and capacitor C317 to output terminal 110 providing the filtered current signal RTF. The outputs of both op-amps OA301 and OA302 __ _ z:~ sos4o Docket No. 13096.20 are combined via resistors 8314 and 8315 and capacitor C311 to provide the filtered current signal FTF at output terminal 100, and the outputs of op-amps OA301 and OA303 are combined via resistors 8312 and 8313 and capacitor C310 to provide the filtered current signal BKF at output terminal 102. This circuit is essentially similar to detector filters disclosed in Fosgate's earlier patents and patent applications, cited above.
In FIG. 5 is shown a detector circuit 12 according to the present invention. Although it is generally similar to the detector circuits of Fosgate's earlier patents and patent applications, this circuit differs from them in providing an improved detector filter axed by providing symmetrical limiting using zener diodes. It comprises two identical circuits 98 and 106.
In the circuit 98, terminals 100 and 102 respectively accept the filtered current signals 11 labeled FTF and BKh from the outputs of the detector filter of FIG. 4. These signals BKF and FTF are respectively applied to the inverting virtual ground inputs of op-amps OA401 and OA402, which have matched monolithic diodes D401, D402 and D403~, D404, connected in antiparallel as their feedback elements, their non-inverting inputs being grounded. These diodes perform a logarithmic function on the inputs as they have a strictly exponential current to voltage relationship. These signals are applied to full-wave rectifiers comprising inverter OA403 with matched diodes D405 and D406 and matched resistors 8401 and 8402, and inverter OA404 with matched diodes D407 and D408 and matched resistors 8403 and 8404. Thus far, the log-ratio detector of the present invention follows the prior art topology.
An improved filter circuit for the 1BKF' rectifier comprises the resistors 8405 and 8409 and capacitors C401, C403 and C405. Resistor 8407 provides forward bias current for the diodes D405, D406. An exactly similar circuit is provided for the FTF rectifier, comprising resistors 8406, 8410, capacitors C402, C404 and C406, with resistor 8408 providing bias current for diodes D407 and D408. The two filter outputs .are combined by resistors 8411 and 8412 into the virtual ground inverting input of op-amp OA405, which has a _1;~_ _ ~t X084 0 Docket No. 13096.20 feedback resistor 8414 and capacitor C407, and employs two zener diodes D409 and D410 connected back to back and in series to provide symmetrical limiting of the control signal generated by the rectifiers. Resistor 8413 and trimpot 8415 provide adjustable offset compensation to op-amp OA405, whose output at terminal 104 is the FB detector output signal. This signal goes positive for signals having predominantly antiphase or back information, and negative for signals having predominantly in-phase or front information content.
An exactly similar circuit 106 provides the output signal L/R,, which goes negative for left and positive for right signals, at terminal 112, when provided with inputs RTF and LTF at terminals :108 and 110 respectively. The FB and L/R, output signals 13 are then applied t,o the detector splitter. They are also connected to the auto-balance circuit of FIG. 3, as shown therein, although this connection is not shown in FIG. 2.
Although the changes to the filter circuitry in FIG. 5, relative to previous log-ratio detector circuitry disclosed in Fosgate's previous patents and patent applications, appears simple, the modifications greatly improve detector performance, particularly in providing signals that are more accurately related to the dominant information in the sound field, with much lower ripple, and less sensitivity to low-level information and artifacts in the input signals.
In FIG. 6, the incoming detector signals 13, labeled FB and LlR,, and applied to terminals 104 and 112 respectively, are split in a novel detector splitter circuit 14 to provide the output .signals FTBK, LF/R,F and LB/R.B.
This circuit is central to the improvements made in the present invention. To facilitate its understanding, reference should also be made to the graphical representation of the logic voltages shown in FIG. 7, as the direction of the signal varies from back B through left back LB, left L, left front LF to front F, then through right front RF, right R, and right back RB to back B once again.
Though not essential to the operation of this circuit, the signals FB and L/R, first pass through CMOS switches S501 and S502, which have their control inputs connected to the terminal 114 labeled KILL LOGIC with a pull-up _ zi~os~o Docket No. 13096.20 resistor 8501 ensuring that the switches are normally turned on, unless KILL
LOGIC is pulled low to -7.5V. Evidently, when this is done, no signals reach the remaining circuitry of FIG. 6, and the outputs FTBK, LF/R,F and LB/R,B
remain zero under all signal conditions.
A multi-pole switch S505 with poles labeled S505A through S505F
enables the circuit to operate in one of t,wo different modes, a first mode suitable for reproduction of stereo recordings with a frontal emphasis and a second mode suitable for reproduction with a panoramic effect. The switch is shown in the first position selecting the first mode. This switch is not actually present in this form in the full circuit of the surround sound processor but the switching functions it represents are carried out by other control means.
The FB signal, shown as the solid curve on the top line of FIG. 7, is applied via switch 5501 to the non-inverting input of an op-amp OA501, connected as a positive half=wave rectifier, such that when its non-inverting input signal is more positive than its inverting input, its output goes positive until diode D501 conducts, and when its non-inverting input is negative with respect to its inverting input, its output goes negative, cutting off' diode D501.
The FB signal is also applied to the inverter comprising op-amp OA505 with input resistor 8502 and feedback resistor 8503 defining its gain of -1. The output of this inverter, -FB, is shown as the dashed curve on the top section of FIG. 7.
The L/R, signal, shown on the second line of FIG. 7, is applied via switch 5502 to the non-inverting input of op-am:p OA502, with diode D502, also connected as a positive half wave rectifier. Since 8504 biases the junctions of diodes D501 and D502 into conduction, the voltage appearing at this junction, point A, is the more positive (or less negative) one of the two signals FB and L/R,. This is the solid curve on the third :line of FIG. 7.
Op-amp OA503 also receives its input from the switched L/R, signal and with diode D504 is configured similarly to OA502 and D502, except that the diode polarity is reversed, thereby providing a negative half wave rectifier.
In 2:~so$~o Docket No. 13096.20 the first position of switch S505B, op-amp OA504 receives the -FB signal from inverter OA505. Op-amp OA504 and diode D504 act as a negative half wave rectifier. Since resistor 8505 biases thE~ diodes D503 and D504 positively, the voltage at their junction point B is therefore the more negative or less positive one of the signals LlR, and -FB. The resulting signal at point B is shown as the solid curve on the fourth line of FICx. 7.
The signal at point A, the junction of diodes D501 and D502, is applied via a resistor 8506 to the non-inverting' input of op-amp OA506, which with diode D505 is connected as a negative half wave rectifier. A CMOS switch 5503 is connected between this point and ground. Similarly, the junction of diodes D503 and D504, point B, is connected via a resistor 8507 to the non-inverting input of op-amp OA507, which forms a positive half wave rectifier with diode D506. This point is also connected via switch 5504 to ground.
Both switches 5503 and S504 have a common control input tom terminal 116 labeled CORNER LOGIC KILL, which is biased to -7.5V via resistor 8508. When the voltage on terminal 116 is switched positive, both switches 5503 and S504 ground the inputs of op-amps OA506 and OA507 respectively, disabling them.
The output signal of op-amp OA5~06 at point C goes negative whenever both FB and L/R, signals are negative, and follows the one of these that is less negative. This is represented in FIG. 7 by the solid curve labeled C on the fifth line, which is zero except for the portion between L and F, and reaches its maximum negative excursion at LF.
The output of op-amp OA507 at point D goes positive whenever both L/R
and -FB signals are positive, following the less positive one of these signals.
This is represented by the curve D on the sixth line of FIG. 7, which is zero except between F and R, and reaches it.. maximum positive excursion at RF.
Points C and D go to the remaining circuitry of FIG. 6 in one of two different ways, this being represented by a multi-pole double throw switch 5505 having two poles labeled S505C and S505D.

_~~60840 Docket No. 13096.20 In the first position of switch 5505, point C is connected via switch pole S505D to the input resistor 8509 of an inverting amplifier OA508, which has a feedback resistor 8510 defining its gain as -1. At its output at point E, therefore, the signal is the inverse of that at C, and is shown on the seventh line of FIG. 7 as the solid curve.
The FB signal is connected through resistor 8511 to a summing junction at the virtual ground inverting input of op-amp OA509. Point D is connected via an equal summing resistor 8512 and point E via another equal summing resistor 8513 to the summing junction at the inverting input of op-amp OA509, so that the signals D, E and FB are swarmed. Op-amp OA509 has an equal feedback resistor 8514, and its output i;s connected to the terminal 118 to provide the signal FTBK, which is the inverted version of the sum of signals D, E, and FB, shown as the solid line F'TBK at the bottom of FIG. 7. Between L and R, this signal remains at zero from L to LF, rises to maximum at F, falls back to zero at RF, remaining at zero to R. In the back half it is identical to the -FB signal.
The signals C and D are summed into the input of op-amp OA510 through summing resistors 8515 and 8,516 respectively, the gain of this summing amplifier being defined by the feedback resistor 8517 to be 2. The output of OA510 will therefore swing positive in the region between L and F, and negative in the region between F and R, but remain zero between L and B
or between R and B. This output signal at point F is shown as the solid curve on the eighth line of FIG. 7.
In the first position of switch S50fi, the signal at point F is connected through switch pole S505E to appear at the LF/R,F output terminal 120.
The signal at point F is also fed via summing resistor 8518 to the summing j unction at the inverting virtu~~l ground input of op-amp OA511, and the L/R signal is fed to this junction via the summing resistor 8519, of half the value. The sum of these two signals is produced at the output G of op-amp Docket No. 13096.20 OA511, which has a feedback resistor 8520 defining its gain to be unity with respect to the signal L/R,.
The contribution from the signal at point F causes the signal at point G
to follow the pattern of the solid curve on the ninth line in FIG. 7.
Beginning at zero at the back point B, this signal ;rises to the most positive value at the L
point, returning to zero at LF, remaining at zero from LF to RF, then falling to the negative maximum value at R and rising back to zero at B once again.
With switch pole S505F in the first position, this output appears at the terminal 122, labeled LB/RB.
In the first mode, then, the sum of the magnitudes of the control signals FTBK, LF/R,F and LB/R,B is maintained as more or less a constant value, and furthermore, only two of the three signals are non-zero at any point on the pan locus. This is important to ensure that undesirable effects do not occur, as happens typically when more than two control voltage signals are partially on at the same time, or if the sum of the control signals is allowed to exceed the value any one of them reaches at its maxima.
In the second mode of operation, t;he switch S505 with poles S505A
through S505F are in the alternate position to the position shown. The FB
signal is connected via switch S505B to the input of op-amp OA504, while the inverted signal -FB from the output of op-amp OA505 now goes via switch S505A to the input of op-amp OA501. Thus the polarity of the front-back signal at these inputs is reversed.
Since point A now follows the more positive of -FB and L/R,, as shown by the dashed curve on the third line of FIC~. 7, it is negative only in the region between L and B. Similarly, point B is now the more negative of L/R and FB, and follows the dashed curve on the fourth line of FIG. 7, and is positive only between R and B.
Point C therefore goes negative between R and B, remaining zero elsewhere, while point D goes positive between B and L, remaining zero 21.60840 Docket No. 13096.20 elsewhere. These signals are shown as dotted curves on the fifth and sixth lines of FIG. 7, respectively.
With switch S505D in the altern;~te position to that shown, the signal at point E is now the inversion of that at point D, and follows the dotted curve on the seventh line of FIG. 7. The signal at point F, which is the inverted sum of the signals at points C and D, now follows the dotted curve on the eighth line of FIG. 7, going positive between B and L, zero across the fTOnt, and negative between R and B. This output signal at point F is applied via switch pole S505F to the LB/R,B terminal.
In the second position of switch 5505, the signals at points C and E are summed through resistors 8512 and 8513 respectively into the virtual ground at the inverting input of op-amp OA509, with the FB signal from 5501 via resistor 8511, and the feedback resistor 8514 provides unity gain, so that the output of op-amp OA509, labeled FTBK and appearing at terminal 118, is the inversion of the sum of the three signals FB, C and E. This is shown at the bottom line of FIG. 7, following the dashed curve. It is at its maximum negative value at B, rising to zero at LB, to maximum positive at F, then falling to zero at R, remaining zero between R and RB, and falling again to its maximum negative value at B.
Once again, the sum of the magnitudes of the three signals FTBK, LF/R,F and LB/R,B is maintained at apps oximately the maximum value reached by any one of them, and only two of then are non-zero at any point on the pan locus. However, it should be noted that in the first position of switch S505, the LF/R,F signal peaks at the points LF and RF on the pan locus, and the LB/R,B
signal peaks at L and R, while in the second position of switch S505, the LF/RF
signal peaks at the points L and R on th.e pan locus, and the LB/R,B signal peaks at the points LB and RB.
In either the first or second mode of operation, as selected by switch S505, when CORNER LOGIC KILL is in effect, the signals at points C and D
become zero, and therefore so do the signals at points E and F, disabling the Docket No. 13096.20 entire central portion of FIG. 6. The output G of op-amp OA511 in this case is the inversion of the L/R, signal, while the output of op-amp OA510 remains identically zero, and the FTBK signal i:ollows the -FB signal. Note that point G is still switched between the LF/R,F terminal 120 and the LB/R,B terminal 122, the other of these terminals being switched to point F, which remains at zero potential.
FIG. 8 shows a three-channel servologic system, 16, in which the three detector splitter output signals 15, labeled LF/R,F, FTBK and LB/R,B, are each filtered through variable low-pass filter elements realized with pulse width modulation circuitry, and then are bufFered and split into pairs of control voltage signals 17 labeled RFC, LFC, BKC, FTC, LBC and RBC.
The LF/R,F signal applied to terminal 120 is passed through a resistor 8801 in parallel with a CMOS switch S801, and then through resistor 8802 and 8803 to the non-inverting input of op-amp U801, connected as a unity gain source follower buffer. Capacitors C801 and C802, connected to ground respectively from the junction of resistors 8802 and 8,803 and from the non-inverting input of op-amp U801, form a two-pole smoothing filter, which represents an improvement over the single-pole type of filter used in prior servologic circuits as shown in Fosgate's previous patents and patent applications.
The action of the CMOS switch 5801, when operated at a high frequency by a pulse width modulated (PWM) rectangular waveform of variable duty cycle derived as explained below, is to vary the effective resistance of the paxallel combination of resistor 8801 anti switch S801, thus varying the time constant of these resistors with resistor 8802 and capacitor C801 to provide more or less smoothing of the detector sputter output signal LF/R,F.
The output of the filter formed by switch S801, resistors 8801 through 8803 and capacitors C801 and C802 is buffered by the op-amp U801, and inverted by the unity gain inverter formed by op-amp U802 with input resistor 8804 and equal feedback resistor 8805. It is also passed to the input of a 24~ -~;~sas~o Docket No. 13096.20 negative half wave rectifier formed by op-amp U803 and diode D801, whose output is the control voltage signal labE~led LFC at terminal 124. Since the signal LF/R,F is positive for leftward signals and negative for rightward signals (see FIG. 7), the signal RFC goes negative only for right front or right signals.
The output of the inverter U802 is passed to the input of a negative half wave rectifier formed by op-amp U804 with diode D802, whose output is the control voltage signal LFC appearing apt terminal 126. This goes negative only for left or left front signals.
The signal FTBK applied to terminal 118 is processed in exactly the same manner by the identical variable filter, inverter and rectifier circuit comprising CMOS switch 5802, resistors 8812 through 8816, capacitors C803 and C804, diodes D805 and D806, and op-amps U806 through U809, to provide output control voltage signals BKC and FTC at terminals 128 and 130 respectively. Since FTBK is negative for signals in the back half of the pan locus, the signal BKC goes negative for back signals, and the signal FTC taken from the inverter U807 goes negative for front signals.
Similarly, the signal LB/R,B applied to terminal 122 is processed in the same manner by the identical variable filter, inverter and rectifier circuit comprising CMOS switch 5803, resistors 8823 through 8827, capacitors C805 and C806, diodes D809 and D810, and op-amps U811 through U814, to provide output control voltage signals RBC and LBC at terminals 132 and 134 respectively.
The switches S801 through S803 in these three identical circuits are operated by a common PWM rectangular wave driven from the output of Schmitt trigger inverter U818, which signal is generated by the remaining circuitry of FIG. 8 as explained below.
The signal LF/R,F is applied via a resistor 8807 to the inverting input of op-amp U805. The filtered and inverted output signal at the output of op-amp U802 is also applied through the equal resistor 8806 to this point, so that the input current is proportional to the difference between the incoming LF/R,F
- 2:5 -2~ 60840 Docket No. 13096.20 detector splitter output signal and its filtered version after the buffer U801.
The op-amp U803, with diodes D803 and D804 and feedback resistor 8808, forms a half wave summing rectifier circuit which provides an output voltage that is the inverse of the difference of LF/R,F and its filtered version at whenever this is negative, and is zero whenever the difference signal is positive.
Resistors 8810 and 8809 respectively from the FTBK signal and its inverted filtered version at the output of op-amp U802, and resistor 8811 from the output of the half wave rectifier U805, are connected to a common virtual ground summing junction at the inverting input of op-amp U816. The resultant current from these three resistors is the full-wave rectified or absolute value of the difference signal between FTBK and its filtered version at U801, applied via an effective resistance in this case of 20kS2, the value of resistor 8810.
A similar full-wave rectifier circuit is formed by op-amp U810 with diodes D807 and D808, and resistors 8817 through 8822, in the middle section of FIG. 8, providing a current to the common summing junction at the inverting input of op-amp U816 equal to the absolute value of the difference signal between the FTBK input at terndnal 118 and its filtered version at the output of op-amp U806 divided by the effective resistance of lOkS2, the value of resistor 8821.
Another similar full-wave rectifier circuit is formed by op-amp U815 with diodes D811 and D812, and resistors 8828 through 8833, in the lower middle section of FIG. 8. This provides a current to the common summing junction of at the inverting input of the op-amp U8:16 equal to the difference between the LBlRB detector splitter output signal applied to terminal 122 and its filtered version at the output of op-amp U811, divided by the effective resistance of lOkS2, the value of resistor 8832.
Thus the three input signals LF/R,F, LB/R,B and FTBK each contribute to the absolute value current driving the: input of op-amp U816 in the lower -2Ei-Docket No. 13096.20 portion of FIG. 8, but not equally, the LFlR,F signal contributing at only half the level of the other two. This applies whatever the position of the switch S505 of FIG 6 in the detector splitter circuit. This is in order to prevent the operation of the LF/R,F control voltage f~~om having too much effect on the responsiveness of the control voltages, particularly in the presence of center front signals such as dialog in movie soundtracks and soloists in many stereo recordings.
The output voltage of op-amp U816 is developed across the feedback resistor 8834, and therefore has a gain of 1.5 to the difference signals for FTBK and LB/R,B and 0.75 to the difference signal for LF/R,F. It is applied via a two-pole low-pass filter comprising op-.amp U817 with associated resistors 8835 and 8836 and capacitors C807 and C808, to the potentiometer 8837, the wiper of which is connected via resistor 8838 to the input of Schmitt trigger inverter U818. A bias voltage is applied by returning the potentiometer to +15V.
A high frequency oscillator is formed by the two Schmitt trigger inverters U819 and U820, with associated resistors 8839 through 8841, capacitor C809 and diode D813. This operates the CMOS switch S804 connected between the negative -7.5V supply voltage and the junction of equal resistors 8842 and 8843 connected betwE~en the positive +7.5V and negative -7.5V supply voltages. A capacitor C810 vvith resistor 8844 in series provides a time constant with these resistors 8843 and 8842.
When the switch S804 is operated at the frequency determined by the oscillator circuit, a negative-going pulse waveform appears at the junction of resistors 8842 and 8843, having a fast negative-going leading edge and a much slower exponential shaped trailing edge. This waveform is applied via capacitor C811 to the input of Schmitt trigger inverter U818, and is effectively biased by the d.c. component of the voltage at the wiper of potentiometer 8837, which is set such that if the absolute value of the combined difference signals appearing at the output of op-amp U816 :is zero, the pulse waveform just barely _. ~~6Q8~0 Docket No. 13096.20 fails to switch the inverter U818, and hence to operate the switches S801 through 5803.
Therefore, in this quiescent condition, the time constants of the servologic filters are all at their longest possible value, defined by 8801, and C801, of about 22.7ms with the resistor and capacitor values shown.
When the output of op-amp U816 goes negative, the pulse waveform is allowed to pass through inverter U818, and its duty ratio increases with the value of the negative-going absolute difference signal, causing the effective resistance of the combinations of switchE~s S801 through S803 with their parallel resistors 8801, 8812 and 8823 respectively, to fall to an effective value of 100kS2 x ( 1 - d) where d is the duty ratio of the PWM signal at the output of Schmitt trigger inverter U818.
This shortens the time constants of each of the three filter circuits, causing them to respond more quickly to changes in the values of the input signals LF/R,F, LB/RB and FT/BK. The minimum effective time constant is about 3.5ms, but the second stage of filtE~ring provided by 8803 and C802 in the upper filter and corresponding components in the other filters provides a much smoother output than was the case in prior single-pole filters used in the servologic circuits described in Fosgate's previous patents and patent applications. Both the maximum and minimum time constants are shorter than was previously possible, due to the :improved smoothing provided by the extra filter pole, making the circuit significantly more responsive to rapid directionality changes in the audio input signals.
In effect, a negative feedback servo loop also exists in this circuit, as a shorter response time reduces the difference between the input and output of each of the filter circuits, thereby tending to reduce the duty ratio of the PWM
signal, hence the term servologic used to describe this circuit.
Thus the servologic circuit provides at its six output terminals a set of control voltage signals which respond to t;he output voltages produced by the 2~.~60840 Docket No. 13096.20 direction detector circuit and the detect;or splitter circuit in a manner which reflects the speed with which any of these voltages are changing.
The maj ority of this circuitry is .,imilar to that described in Fosgate's previous patents and patent applications, the novel features of the present circuitry being the extension to provide for three input signals and to generate six output control voltage signals, the provision of a two-pole variable filter circuit which greatly improves the speed and smoothness of operation of the circuit, and the different relative contributions of the LF/R,F, LB/R,B and FTBK
signals to the absolute difference signal used to control the PWM duty ratio.
Each of these six control voltage signals LFC, RFC, FTC, BKC, LBC and RBC is connected to the control port of a voltage-controlled amplifier, of the type shown in FIG. 9. While this is basically similar to the circuit used in the previous Fosgate patent applications, a novel feature of this circuit is a linearity correction netwvork placed between the control voltage generator and the VCA input.
In FIG. 9, the LF VCA block 18 of FIG. 2 is shown in a detailed form.
The LT and RT signals 3 from the input matrix 6 in FIG 2 are applied via the correspondingly labeled input terminals to a direct path and a side path, the latter incorporating a gain control elemE~nt. In the direct path, the LT and RT
signals are respectively applied through resistors 8902 and 8901 to the inverting input of op-amp U902, which has a feedback resistor of 26.?kS2. The values of 8901 and 8902 are so chosen that an in-phase blend of LT and RT
occurs, the level of RT being about -9.6dB relative to that of LT, and the gain to a true left front signal is 0.977. For this purpose a true left front signal is defined to have components LT = 0.92388 V cos wt, and RT = 0.38268 V cos wt, the effective rms amplitude of the pair being 1.
The side path comprises most of the remaining circuit in the top section of FIG. 9. Resistors 8903 and 8904 apply the signals LT and RT respectively to a low impedance junction of the resistor 8906 and the series combination of potentiometer 8915 and junction FET Q901. When the gate of Q901 is near _2c,~_ Docket No. 13096.20 ground potential, the device has an effective series resistance of about 20052 or less, the difference being taken up by the potentiometer, which is adjusted for an effective total resistance of about 30052. Thus about 83% of the total current flows into the FET and only 17% into the l.5kS2 resistor 8906 and hence into the inverting input of op-amp U901. This has a feedback resistor 8909 of 110kS2, and the values of resistors 8903 and 8904 are approximately four times those of resistors 8902 and 8901 respeci;ively.
The value of 8910 is 24.9K, and the values of 8909 and 8910 are chosen so that when Q901 is cut off, the signal current through the side chain resistor 8910 almost exactly cancels the signal currents in the direct path resistors 8901 and 8902 flowing into the inverting input of op-amp U902. Thus the gain of the VCA is minimum under this condition. The output of the VCA, the signal -LF, is taken at the output of op-amp U902, and an inverter formed by op-amp U903 with equal resistors 8913 and 8914 provides the LF signal output, this pair of outputs LF and -LF being identified by the numeral 19 as in FIG. 2.
The potentiometer 8905 and resisi;ors 8907 and 8908 provide an offset compensation voltage to the non-inverting input of op-amp U901 to minimize the DC voltages at the outputs of op-amps U902 and U903 as the gain of the VCA changes.
When FET (1901 has its minimum resistance, the side chain current is proportionally reduced to about 17% of its maximum value, allowing the overall gain to be 83% of the value calculated ffom the direct path alone.
The resistance of FET Q901 is varied by adjusting its gate voltage. Op-amp U904 provides this gate voltage via .a gate current limiting resistor 8916.
The LFC control voltage signal 17 from the servologic circuit of FIG. 8 at terminal 126 is applied to the virtual ground inverting input of the op-amp U904 through a linearity correction network comprising zener diode D901, diode D902 and resistors 8922 through R~904. In addition, a bias current is applied to this input through resistor 891.9 from a common +13.5V voltage x!16 0840 Docket No. 13096.20 derived from the +15V supply via two sE:ries diodes D903 and D904, and decoupled with capacitor C901. The effE~ct of this bias current is to make the voltage at the output of U904 negative, the exact voltage being determined by the setting of potentiometer 8917 in series with resistor 8918, and is about -when LFC is at ground potential. It is set to just below the pinch-off voltage of FET Q901, to the point at which the maximum cancellation of the direct signal is achieved through the side chain signal.
As the LFC control voltage signal at terminal 126 goes negative, the output voltage of op-amp U904 applied to the gate of transistor Q901 rises above the pinch-off voltage and causes the transistor's effective resistance to decrease, thereby increasing the attenuation through the side chain and increasing the gain of the VCA. Above a certain negative voltage value, zener diode D901 begins to conduct, and pulls the voltage on resistor 8922 negative.
Some current begins to flow through resistor 8923 and diode D902 when the voltage at LFC goes to about -4.5V. Beyond this, the gain of the control path increases, compensating to some extent for a flattening out of the control voltage signal as it nears -6V, and for the changing characteristics of the VCA
as the FET Ca901 nears its minimum resistance value.
The resistors 8921 and 8920 apply a proportion of the a.c. voltage at the drain of FET (1901 to its gate to compen.5ate for even-order distortion introduced by the square-law nonlinearity of the FET. These values have been selected to minimize VCA distortion at all gain settings of the VCA.
In discussing the properties of this VCA below with reference to matrixing in FIGS. 10 and 11, we shall rE~gard the VCA as having a gain coefficient kLF with reference to an effective input resistance of 24.9k, this being the value shown for 8929 in the LB VCA 26 to be further discussed below. In the case of the left front (LF) VCA, the output signal is kLF times the summed signals (0.8676 LT + 0.2875 RT), i.e.
LF = kLF (0.8676 LT + 0.2875 RT) Docket No. 13096.20 where LT and RT are the respective amplitudes of the signals in the left and right channels following the input matrix circuit 6.
An exactly similar circuit is provided in the FT VCA block 22 of FIG. 9, also shown in FIG. 2. in this case, however, the resistor values from LT and RT signal inputs are different. Resistors 8926 and 8925 respectively correspond to resistors 8901 and 8902 in VCA 18, but have equal values of 49.9 kS2. Similarly, resistors 8927 and 8928 correspond to resistors 8904 and 8904 respectively, but are also equal at; 200kS2. These values provide maximum gain for a front signal, definE~d by the equal LT and RT values of 0.70711 V cos wt. The control signal voltage FTC is applied via terminal 130 to a similar nonlinear correction network as shown in the circuitry of VCA 18, and acts to increase the gain of the fror.~t VCA as the FTC control voltage signal goes negative. The front outputs are provided at the terminals 23 labeled -FT and FT. In this case the coefficients are each exactly 0.5, yielding the equation FT = kF.z. (0.5 LT + 0.5 RT) Another similar circuit is provided in circuit block 26, the LB VCA of FIG. 2. In this case, there is no RT input, the LT input being applied to resistors 8929 and 8930 to the direct and side chain paths respectively. The outputs 27 appear at terminals LB and -LB. The LBC control voltage signal is applied to terminal 134. The equation describing this output is stall simpler, namely:
LB = kLB LT
The circuitry shown in FIG. 9 represents the left and front VCA's of FIG.
2, but in addition, there are similar VCA's provided for the RFC, RBC and BKC
control voltages, in an almost identical circuit (not shown). The RF VCA 20 in FIG. 2 for the RFC control voltage signal is exactly like the LF VCA 18, except that the signals LT and RT are applied to the opposite terminals, i.e. RT is applied to resistors 8902 and 8903, while LT is applied to resistors 8901 and 8904. Similarly, the BK VCA 24 is like the FT VCA 22 except that instead of ._. . -32-2~~0840 Docket No. 13096.20 the RT signal, the -RT signal is applied to resistors 8926 and 8927. In the RB
VCA 28, which is exactly like the LB VCA 26, the RT signal is applied to both resistors 8929 and 8930 instead of the LT signal. As a result, the equations for RF, BK and RB signals are, respectively:
RF = kRF (0.2875 LT + 0.8676 RT) BK = kBK (0.5 LT - 0.5 RT) RB = kRB RT
The configuration of the VCA's in FIG. 9 is shown for the case of switch 5505 of FIG. 6 in the first position. There is also an alternative configuration of the VCA's (not shown) for the case when switch S505 is in the alternate position. In this alternative configuration, the top VCA of FIG. 9 becomes the LB VCA 26, and receives the signals LT and -RT, the RB VCA 28 receiving signals RT and -LT. The lower VCA 26 becomes the LF VCA 18. The signals applied to the two lower VCA's of FIG. 9 are unchanged in this alternative configuration, which can be achieved in practice by switching the control voltage inputs and signal inputs and outputs as indicated in parallel with the action of switch 5505 of FIG. 6.
The set of signal equations for this alternative configuration is:
LF = kLF LT
FT = k~ (0.5 LT + 0.5 RT) LB = kLB (0.8676 LT - 0.2875 RT) RF = kRF RT
BK = kBK (0.5 LT - 0.5 RT) RB = kRB (-0.2875 LT + 0.8676 RT;) In FIG 10 is shown a first matrixi:ng network suitable for use with the first mode of the detector splitter circuit of FIG. 6, with switch 5505 in the first position as shown. FIG. l0a shows the circuitry for the portions of the OUTPUT MATRIX block 30 of FIG. 2 dedicated to the LF and RF outputs, with output buffers 32 and 34, while FICA. lOb shows the remaining matrix circuitry 30 for the CF, LB and RB outputs and the output buffers 36, 38 and _ 33 _ ~~ sog~o Docket No. 13096.20 40. The circuitry is conventional but the resistors are chosen for the specific matrixing coefficients that are desired for the first mode of the detector sputter circuit of FIG. 6 and the first configuration of VCA's shown in FIG. 9.
In FIG. 10a, the LF matrix, part of the matrix block 30 of FIG. 2, comprises the resistors 81001 through 81010 and CMOS switch S 1001. These resistors are connected into a virtual ground summing junction at the input of the LF buffer 32, which comprises the op-amp U1001 and associated capacitor C 1001 and resistors 81011 and 81012.
The feedback resistor 81011 is adjustable to yield a maximum gain for any input of -0.912. Typically the potentiometer will, however, be set back to give a gain of 0.707 or -3dB for the largest input, to avoid any possibility of overloading. The resistor 81012 and capacitor C 1001 are provided to a.c.
couple the op-amp and provide 100% d.c., negative feedback around it to minimize its output offset voltage.
The summing resistors 81001 through 81009 are chosen to provide specific matrixing coefficients, relative to the value of 27.4kS2 which results in a coefficient of 1.000. These coefficient values are shown to the right of the respective resistors. The input terminals of the circuit are labeled with the signals they receive, either the LT, RT, -:LT or -RT signals 3 from the input matrix 6 of FIG. 2, or the positive or negative polarity signals 19 - 29 from the six VCA circuits 18 - 28, labeled +LF, +RB etc. The signal labeled BF is obtained from the sum of signals LT and RT, passed through a three-pole low-pass filter (not shown) which is an optional feature of surround processors of this type discussed in an earlier Fosgate patent. The purpose of this filter is to cancel out the bass component of the signals presented to the surround processor so that the majority of the processing is performed essentially in the mid-frequency band.
The specific coefficients shown are .699 for LT, .294 for -RT, .113 for +LF, .294 for +RB, 1.000 for BF, .393 for -FT, .699 for -LB, 1.000 for -BK and .374 for +LB when switch 51001 is enabled by the signal CORNER LOGIC

Docket No. 13096.20 KILL at terminal 116, which was discussed with reference to FIG. 6. Thus the signal equation for the output LFO at terminal 42, using the first set of VCA
equations above for FIG. 9, is:
LFO = 0.699 LT - 0.294 RT
+ 0.113 kLF (0.8676 LT + 0.2875 RT) +0.294kRBRT+BF
- 0.393 k~, (0.5 LT + 0.5 RT) - 0.699 kLB LT
- kBK (0.5 LT - 0.5 RT) + 0.374 CLK kLB LT
Although this equation may appear complicated, it illustrates how the cancellation is achieved, if one remembers that the k values vary between 0 and 1 with the corresponding control voltage signals, that no more than two at a time are non-zero, and that the total of all the k values does not exceed 1.
Thus, for a pure LB signal, kLB = 1 and all the other k values are 0, so the LFO
signal's 0.699 LT component is canceled out by the -0.699 kLB LT term. The -0.294 RT term is similarly canceled out by the 0.294 kRB RT term for a pure RB
signal. When CORNER LOGIC KILL (CLK) is on, the LB cancellation term is reduced, since there is now a +.374 kLB LT term introduced via the switch S 1001. Thus the LT term in the LF output is not totally canceled but reduced from 0.699 to 0.325 or about 7dB lower.
The RF matrix portion of matrix 30 is symmetrically identical to the LF
matrix, i.e. with left and right signals interchanged. The BK term is in the opposite polarity. Resistors 81013 through 81022 again define the various coefficients, which are identical to those in the LF matrix, with S 1002 operated by the CORNER LOGIC KILL signal (CLK) at terminal 116, which is duplicated here only to improve clarity and aid understanding. The RF buffer 34 is identical to the LF buffer 32 and comprises the op-amp U1002 with resistors 81023 and 81024, and capacitor C 1002. Thus the RFO output signal at terminal 44 is represented by the equation:
RFO = 0.699 RT - 0.294 LT
+ 0.113 kRF (0.8676 RT + 0.2875 LT) ~ a~ s o s ~ 0 Docket No. 13096.20 + 0.294 kLB LT + BF
- 0.393 k~ (0.5 LT + 0.5 R'r) - 0.699 kRB RT
+ kBK (0.5 LT - 0.5 RT) + 0.374 CLK kRB RT
Referring to FIG. lOb, the CF matrix portion of matrix 30 is similar, but simpler, comprising resistors 81025 through 81031 with their corresponding coefficients as shown to the right of each resistor. The CF buffer comprising op-amp U1003, capacitor C1003, and re~~istors 81032 and 81033 is again identical to those for LF and RF. The corresponding output equation for the signal CFO at terminal 46 is also simplified, since there is no switch in this circuit:
CFO = 0.591 LT + 0.591 RT - 0.591 kLB LT
-0.591kRBRT+0.183k~.(:0.5LT+0.5RT) - 0.715 kLF (0.8676 LT + 0.2875 RT) 0.715 kRF (0.8676 RT + 0.2875 LT) The LB matrix portion of matrix 30 has nine resistors 81034 through 81042 defining the coefficients shown to their right, and the combined signal may also be switched via a delay circuit 138 in some modes of operation by means of switches S 1003 and 51004 controlled by the DELAY IN/OUT signal applied to terminal 136. The LB buffer 38 comprises op-amp U1004 with resistors 81043 through 81045 and capacitors C 1004 and C 1005. It is essentially the same as the previously described buffers, with the exception that a shelf filter is created by the additional feedback components C 1004 and 81043 shown. This has the effect of reducing the high frequency gain by about 5 - 6 dB which can be effective in reducing sibilant "splash" from center front dialog, especially in movie soundtracks. 'The equation for the output signal LBO appearing at terminal 48 is:
LBO = 0.699 LT - 0.499 RT
- 0.200 k~ (0.5 LT + 0.5 RT) - 0.226 kBK (0.5 LT - 0.5 RT) + 0.499 kRB RT
+ 0.137 kLB LT

_ ~1~fi0840 Docket No. 13096.20 - 0.488 kLF (0.8676 LT + 0.2875 RT) + 0.206 kRF (0.8676 RT + 0.2875 LT) + 0.274 BF
The RB matrix circuit portion of matrix 30 is comprised of an identical set of resistors 81046 through 81054, also with a delay 140 switched in or bypassed by switches S1005 and S1006, switched by the DELAY IN/OUT
signal at terminal 136, again duplicated for clarity. The same coefficients are used, but the left and right channels are swapped and the opposite polarity of the BK signal is used. The RB buffer 4U is identical to the LB buffer 38, comprising op-amp U1005, resistors 81055 through 81057 and capacitors C 1006 through C 1007. The equation for the RBO signal appearing at terminal 50 is:
RBO = 0.699 RT - 0.499 LT
- 0.200 k~ (0.5 LT + 0.5 RT) + 0.226 kBK (0.5 LT - 0.5 RT) + 0.499 kLB LT + 0.137 kRB RT
- 0.488 kRF (0.8676 RT + 0.2875 LT) + 0.206 kLF (0.8676 LT + 0.2875 RT) + 0.274 BF
FIG. 11 shows a similar matrixing network suitable for use with the second mode of the detector splitter circuit of FIG. 6 and the alternative configuration of FIG. 9 discussed previously. The main differences between FIGS 10 and 11 are the values of the matrixing resistors and corresponding matrixing coefficients.
To simplify the discussion of FIGS. lla and llb, observe that corresponding resistors are numbered correspondingly to those in FIGS. l0a and lOb, as far as possible, except for being Rllxx instead of RlOxx. All of the buffer circuits are identical to those shown in FIGs. l0a and lOb, with the same nomenclature differences.
In FIG. lla, one additional resistor 81158 appears in the LF matrix and a corresponding resistor 81159 appears in the RF matrix portion of matrix 30.
These respectively apply canceling signals for the RT and LT antiphase 2.160840 Docket No. 13096.20 components of the LF and RF matrix, this feature of antiphase blending having been disclosed in an earlier Fosgate patent application. With the coefficients shown to the right of the several resistors in the matrix 30, the equations for the outputs LFO and RFO appearing respectively at terminals 42 and 44 are, respectively, using the second set of equations for the alternative VCA
configuration of FIG. 9 above:
LFO = 0.699 LT - 0.294 RT + 0.113 kLF LT
+ 0.591 kRB (-0.2875 LT + 0.8676 RT) + BF
- 0.393 k~ (0.5 LT + 0.5 RT) - 0.825 kLB (0.8676 LT - 0.:2875 RT) - kBK (0.5 LT - 0.5 RT) + 0..294 kRF RT
+ 0.374 CLK kLB (0.8676 LT - 0.2875 RT) RFO=0.699RT-0.294 LT+0.113kRFRT
+ 0.591 kLB (0.8676 LT - 0.2875 RT) + BF
- 0.393 kF,z, (0.5 LT + 0.5 R'T) - 0.825 kRB (0.8676 RT - 0.2875 LT) +kBK(0.5LT-0.5RT)+0.294kLFLT
+ 0.374 CLK kRB (0.8676 RT - 0.2875 LT) Turning to FIG. llb, this will be seen to be identical to FIG. lOb except that the components previously numbered l0xx are numbered correspondingly llxx, and some resistors in the matrix portion 30 are absent or have different values. Again, the buffers 36, 38 and 40 are identical to those of FIG. lOb.
In particular, the matrix 30 includes no resistors 81138 or 81150 corresponding with 81038 and 81050 respectively, and resistors 81127, 81128, 81130, 81131, 81139, 81141, 81142, 81151, 81153, and 81154 have different values from their counterparts in FIG. lOb. The corresponding equations for the signals CFO, LBO and RBO appearing at terminals 46, 48, and 50 respectively (again using the second set of equations for FIG. 9 VCA's) are:
CFO = 0.591 LT + 0.591 RT
- 0.365 kLB (0.8676 LT - 0.2875 RT) Docket No. 13096.20 - 0.365 kRB (0.8676 RT - 0.2875 LT) +0.183k~(0.5LT+0.5RT) -0.591kLFLT-0.591kRFRT
LBO = 0.699 LT - 0.499 RT
- 0.200 k~ (0.5 LT + 0.5 RT) - 0.226 kBK (0.5 LT - 0.5 R'r) + 0.825 kRB (0.8676 RT - 0,2875 LT) + 0.137 kLB (0.8676 LT - 0.2875 RT) -0.699kLFLT+0.511kRFRT
RBO = 0.699 RT - 0.499 LT
- 0.200 k~. (0.5 LT + 0.5 R'T) + 0.226 kBK (0.5 LT - 0.5 RT) + 0.825 kLB (0.8676 LT - 0.2875 RT) + 0.137 kRB (0.8676 RT - 0.2875 LT) -0.699kRFRT+0.511kLFLT
To further explain the operation of these equations, the results may be tabulated for each of the principal sound source directions. In the table, the first two columns give the values of LT and RT, and the remaining columns represent the output signals, for each of the input directions listed at the left.
In the first mode of operation of F'IG. 6, i.e. with switch 5505 in the first position as shown, and with the VCA's in the first configuration of FIG. 9, and the matrix values according to FIG. l0a and FIG. lOb, Table 1 gives the loudspeaker output signals for each principal source signal direction. The BF
term is ignored, since it is only effective at low frequencies and is not dependent on the logic action.
TABLE 1 Loudspeaker output signals vs. source direction, with values of FIG. 10 and FIG. 6 5505 in first position, ignoring the BF term.
Src Ctrl Dir Sig LT RT LBO LFO CFORFORBO

~isos4~
Docket No. 13096.20 LB kLB 1.000 0.000 O.~g36 0.0000.0000.0000.000 LF kLF 0.924 0.383 0.010 0.6360.1210.0040.005 CF k~. 0.707 0.707 0.000 0.0080.9650.0080.000 RF kRF 0.383 0.924 -0.005 -0.0040.1210.6360.010 RB kRB 0.000 1.000 0.1)00 0.0000.0000.0000.836 CB kBK 0.707 -0.707 O.ti87 -0.0050.0000.0050.687 Thus, relative to the -3dB level expected from the matrix, we have overall levels for each direction of LB +1.45dB LF -0.76dB C;F +2.70dB CB +2.76dB
in the first mode of FIG. 6 (RB and RF are the same as LB and LF, respectively).
In the second mode of FIG. 6, with the second configuration of the VCA's of FIG. 9 and the matrix values of FIG. 11, we can compute a similar table:
TABLE 2 Loudspeaker output signals vs. source direction, with values of FIG. 11 and FIG. 6 S505 in second position, ignoring the BF term.
Src Ctrl Dir Sig LT RT LBO LF'0 CFORFORBO
LB kLB 0.924 -0.383 0.962 0.0060.0000.0000.000 LF kLF 1.000 0.000 0.000 0.8120.0000.0000.005 CF k~ 0.707 0.707 0.000 0.0080.9650.0080.000 RF kRF 0.000 1.000 0.012 0.0000.0000.8120.010 RB kRB -0.383 0.924 0.023 0.0000.0000.0000.962 CB kBK 0.707 -0.707 0.687 -0.0050.0000.0050.687 Thus, relative to the -3dB level expected tom the matrix, we have overall levels for each direction of LB +2.68dB LF +1.20dB CF +2.70dB CB +2.76dB
in the second mode of FIG. 6.
While the equations given above represent the exact relationships between the signals in both preferred modes of operation here described, of - 4i~ _ ~l~pg~p Docket No. 13096.20 course there may be many variations and modifications to the resistor values for broadening or narrowing the presentation, for adapting the system for an additional center back output, for provision of left and right side loudspeakers, for inclusion of special filtering modes far mufti-band operation, or as related to proprietary sound reproduction systems such as Lucas Arts THX and Dolby Surround, as has been described in Fosgate's earlier patents and patent applications.
These and many other modifications will become apparent to those experienced in the art, without departing from the spirit of the present invention.
-4:1-

Claims (21)

1. A surround sound processor for receiving left and right audio signals of a stereophonic source and for processing said left and right signals for presentation on a plurality of loudspeakers surrounding a listening area so as to produce an impression of discrete sound sources surrounding a listener therein, said processor comprising:
a pair of input terminals for receiving said left and right audio signals;
an input matrix circuit which receives said left and right audio signals from said pair of input terminals and comprises at least an inverting amplifier and a non-inverting amplifier of equal gain for each channel to provide said left and right audio signals in either polarity to succeeding circuits;
a detector filter which receives said left and right audio signals from said input matrix circuit and filters each of them with a suitable transfer characteristic to provide left and right filtered signals;
a detector matrix circuit which receives and combines said left and right filtered signals to provide their sum and difference as additional front and back filtered signals respectively;
a direction detector circuit which receives said left and right filtered signals and processes them to provide a left-right direction signal proportionalto the logarithm of the ratio of the amplitude of said right filtered signal to that of said left filtered signal, up to a limiting voltage, and which also receives said front and back filtered signals and processes them to provide a front-back direction signal proportional to the logarithm of the ratio of the amplitude of said back filtered signal to said front filtered signal, up to the same limitingvoltage;
a detector splitter circuit which receives said left-right and front-back direction signals and processes them to provide one or more additional direction signals while modifying said left-right and front-back direction signals to maintain constancy of the sum of all the said direction signals;

a servologic circuit which receives said modified left-right and front-back direction signals and said one or more additional direction signals from said detector splitter circuit, filters each of them with a variable low-pass filter,inverts each resulting signal and half-wave rectifies each signal and its inversion to provide a plurality of smoothly varying control voltage signals of the same polarity;
a plurality of voltage-controlled amplifiers equal to the plurality of said control voltage signals, a different one of said control voltage signals being connected to control the gain of each said voltage-controlled amplifier, each ofsaid plurality of voltage-controlled amplifiers receiving different proportions of said left or right signals or their inversions from said input matrix circuit, and sums said received signals with a variable gain dependent upon the corresponding one of said plurality of control voltage signals applied thereto;
an output matrix circuit comprising a plurality of matrix circuits equal to said plurality of loudspeakers, each said matrix circuit receiving one or more of said left and right signals and their inversions from said input matrix circuit and one or more output signals from said plurality of voltage-controlled amplifiers, and combining them in suitable proportions to obtain a loudspeaker feed signal wherein unwanted directional components are canceled;
a plurality of output terminals equal to said plurality of loudspeakers;
and a plurality of output buffers equal to said plurality of loudspeakers, each said output buffer receiving one of said plurality of loudspeaker feed signals from said output matrix circuit and amplifying it to an appropriate level for driving a power amplifier connected to one said output terminal of said processor, for driving the corresponding one of said plurality of loudspeakers surrounding said listening area.

2. The processor of claim 1 wherein said detector splitter circuit provides one additional direction signal from said front-back and said left-right direction signals, said plurality of voltage-controlled amplifiers is six, and the corresponding plurality of said control voltage signals is also six.

3. The processor of claim 1 wherein said plurality of matrix circuits is five, and the corresponding plurality of said output buffers and of said output terminals is also five.

4. The processor of claim 1 further comprising switching means in said detector splitter circuit for changing the configuration thereof to a firstforward-oriented configuration or to a second backward-oriented configuration.

5. The processor of claim 1 further comprising switching means in said detector splitter circuit for changing the configuration thereof either to a three-axis configuration wherein said detector splitter provides one said additional direction signal, or to a two-axis configuration wherein said detector splitter circuit provides no additional direction signals but inverts said left-right and said front-back direction signals and provides them to different inputs of said servologic circuit.

6. The processor of claim 1 further comprising switching means for disconnecting the outputs of said direction detector circuit from said detector splitter circuit.

7. The processor of claim 1 wherein said detector splitter circuit in a first forward-oriented configuration comprises:
first and second inputs for receiving said left-right and front-back direction signals respectively, said left-right direction signal being responsive to left/right information content of said left and right stereophonic input signalsand said front-back direction signal being responsive to sum/difference information content of said left and right input signals, such that said left-right input signal goes positive when right information predominates and said front-back direction signal goes positive when difference information predominates;
a first inverter for receiving said front-back direction signal and providing at its output an inverted front-back direction signal of equal magnitude and opposite polarity thereto;
first positive half-wave rectifier for receiving said left-right direction signal;
second positive half-wave rectifier for receiving said front-back direction signal, the outputs of said first and second positive half-wave rectifiers beingconnected to a first common point and a first bias resistor connected therefrom to a negative supply voltage, the voltage at said first common point being equalto the more positive one of said left-right and front-back direction signals;
first negative half-wave rectifier for receiving said left-right direction signal;
second negative half-wave rectifier for receiving said inverted front-back direction signal from said first inverter, the outputs of said first and second negative half-wave rectifiers being connected to a second common point and a second bias resistor connected therefrom to a positive supply voltage, the voltage at said second common point being equal to the more negative one of said left-right and inverted front-back direction signals;
third negative half-wave rectifier connected to receive the voltage at said first common point, the output voltage of which goes negative only when both of said left-right and front-back direction signals are negative;

third positive half-wave rectifier connected to receive the voltage at said second common point, the output of which goes positive only when both of said left-right and inverted front-back direction signals are positive;
second inverter for inverting the output voltage of said third negative half-wave rectifier;
first summing amplifier having first, second and third summing resistors of equal value connected respectively to receive said front-back direction signal, the output of said third positive half-wave rectifier, and the output of said second inverter, and having an equal feedback resistor to provide at its output a voltage that is the inverted polarity sum of the voltages presented to said first, second and third summing resistors;
first output terminal connected to the output of said first summing amplifier for providing a first direction signal responsive to front-back information;
second summing amplifier having fourth and fifth equal summing resistors connected respectively to the outputs of said third positive half-waverectifier and said third negative half-wave rectifier, and having a feedback resistor of double the value of said fourth or fifth summing resistors to provide at its output a voltage that is double the inverted polarity sum of the voltagespresented to said fourth and fifth summing resistors;
second output terminal connected to the output of said second summing amplifier to provide a second direction signal responsive to front left-right information;
third summing amplifier having a sixth equal summing resistor connected to said left-right direction signal and seventh summing resistor of double the value connected to the output of said second summing amplifier, and having a feedback resistor equal to said sixth summing resistor to provide at its output a voltage that is equal to the inverted polarity sum of the said left-right direction signal and one half of the output voltage of said second summing amplifier; and a third output terminal connected to the output of said third summing amplifier for providing a third direction signal responsive to back left-right information;
said detector splitter circuit being operative to provide at said first output terminal said first modified front-back direction signal having a maximum negative value when said sum filtered signal is zero, falling to zero when either the left or the right input signal is zero, and remaining zero untilsaid sum filtered signal exceeds the magnitude of the larger of said left or right filtered signals, rising to a maximum positive signal when said sum filtered signal has maximum amplitude and said difference filtered signal is zero; and to provide at said second output terminal said second front left-right direction signal which is zero whenever said sum filtered signal is smaller in magnitude than said difference filtered signal, rising to a maximum positive value when said sum filtered signal equals said left filtered signal in magnitude, falling to zero when said left and right filtered signals are equal in magnitude, and falling to a maximum negative value when said sum filtered signal equals said right filtered signal in magnitude; and to provide at said third output terminal said third back left-right direction signal which is zero through the region where the said sum filtered signal exceeds both of said left and right filtered signals in magnitude, rising to a maximum positive value when said left filtered signal is of maximum amplitude, falling to zero when said sum filtered signal becomes zero, and to a maximum negative value when said right filtered signal is of maximum amplitude, again reaching zero when the right and sum filtered signals become equal in amplitude.

8. The processor of claim 1 wherein said detector splitter circuit in a second backward-oriented configuration comprises:
first and second inputs for receiving said left-right and front-back direction signals, said left-right direction signal being responsive to left/right information content of said left and right stereophonic input signals and said front-back direction signal being responsive to sum/difference information content of said left and right input signals, such that said left-right input signal goes positive when right information predominates and said front-back direction signal goes positive when difference information predominates;
a first inverter for receiving said front-back direction signal and providing an inverted front-back direction signal of equal magnitude and opposite polarity thereto;
first positive half-wave rectifier for receiving said left-right direction signal;
second positive half-wave rectifier for receiving said inverted front-back direction signal from said first inverter, the outputs of said first and second positive half-wave rectifiers being connected to a first common point and a first bias resistor connected therefrom to a negative supply voltage, the voltage at said first common point being equal to the more positive one of said left-right and inverted front-back direction signals;
first negative half-wave rectifier for receiving said left-right direction signal;
second negative half-wave rectifier for receiving said front-back direction signal, the outputs of said first and second negative half-wave rectifiers beingconnected to a second common point and a second bias resistor connected therefrom to a positive supply voltage, the voltage at said second common point being equal to the more negative one of said left-right and front-back directionsignals;

third negative half-wave rectifier connected to receive the voltage at said first common point, the output voltage of which being negative only when both of said left-right and inverted front-back direction signals are negative;
third positive half-wave rectifier connected to receive the voltage at said second common point, the output voltage of which being positive only when both of said left-right and front-back direction signals are positive;
second inverter for inverting the output voltage of said third positive half-wave rectifier;
first summing amplifier having first, second and third summing resistors of equal value connected respectively to receive said front-back direction signal, the output of said third negative half-wave rectifier, and the output of said second inverter, and having an equal feedback resistor to provide at its output a voltage that is the inverted polarity sum of the voltages presented to said first, second and third summing resistors;
first output terminal connected to the output of said first summing amplifier for providing a first direction signal responsive to front-back information;
second summing amplifier having fourth and fifth equal summing resistors connected respectively to the outputs of said third positive half-waverectifier and said third negative half-wave rectifier, and having a feedback resistor of double the value of said fourth or fifth summing resistors to provide at its output a voltage that is double the inverted polarity sum of the voltagespresented to said fourth and fifth summing resistors;
third summing amplifier having a sixth equal summing resistor connected to said left-right direction signal and seventh summing resistor of double the value connected to the output of said second summing amplifier, and having a feedback resistor equal to said sixth summing resistor to provide at its output a voltage that is equal to the inverted polarity sum of the said left-right direction signal and one half of the output voltage of said second summing amplifier; and a second output terminal connected to the output of said third summing amplifier for providing a second direction signal responsive to front left-rightinformation;
a third output terminal connected to the output of said second summing amplifier to provide a third direction signal responsive to back left-right information;
said detector splitter circuit being operative to provide at said first output terminal said first modified front-back direction signal having a maximum negative value when said sum filtered signal is zero, falling to zero when the larger of said left or said right filtered signal is exceeded in magnitude by said sum filtered signal, and remaining zero until said left or right filtered signal reaches a maximum, rising to a maximum positive signal when said sum filtered signal has maximum amplitude and said difference filtered signal is zero; and to provide at said second output terminal said second front left-right direction signal which is zero through the region where the said difference filtered signal exceeds both of said left and right filtered signals in magnitude, rising to a maximum positive value when said left filtered signal is of maximum amplitude, falling to zero when said difference filtered signal becomes zero, and to a maximum negative value when said right filtered signal is of maximum amplitude, again reaching zero when the right and difference filtered signals become equal in amplitude; and to provide at said third output terminal said third back left-right direction signal which is zero whenever said sum filtered signal is larger in magnitude than said difference filtered signal, rising to a maximum positive value when said difference filtered signal equals said left filtered signal in magnitude, falling to zero when said left and right filtered signals are equal in magnitude, and falling to a maximum negative value when said difference filtered signal equals said right filtered signal in magnitude.

9. The processor of claim 7 wherein said detector splitter circuit further comprises a switch for grounding the inputs of both said third positive half-wave rectifier and said third negative half-wave rectifier, thereby causingthe output voltage of said second summing amplifier to be identically zero and causing the voltage at said first output terminal to be equal to the inverse of said front-back direction signal and the voltage at said third output terminal to be equal to the inverse of said left-right direction signal.

10. The processor of claim 7 wherein said detector splitter circuit further comprises a switch for disconnecting the inputs thereof from both said left-right and front-back direction signals, thereby causing the output voltagesat all three of said first, second and third output terminals to be identically zero.

11. The processor of claim 8 wherein said detector splitter circuit further comprises a switch for grounding the inputs of both said third positive half-wave rectifier and said third negative half-wave rectifier, thereby causingthe output voltage of said second summing amplifier to be identically zero and causing the voltage at said first output terminal to be equal to the inverse of said front-back direction signal and the voltage at said second output terminal to be equal to the inverse of said left-right direction signal.

12. The processor of claim 8 wherein said detector splitter circuit further comprises a switch for disconnecting the inputs thereof from both said left-right and said front-back direction signals, thereby causing the output voltages at all three of said first, second and third output terminals to be identically zero.

13. The processor of claim 8 further comprising a multiple pole switch for changing the connections between the component circuits thereof to said second backward-oriented configuration.

14. The processor of claim 1 wherein said servologic circuit comprises:
first, second and third input terminals for receiving respectively said front left-right direction signal, said modified front-back direction signal, and said back left-right direction signal;
first, second and third low-pass variable filters for filtering the direction signals from said first, second and third input terminals respectively with equal variable time constants dependent upon a control signal applied in common to a control port of each said variable filter for varying its cut-off frequency;
first, second and third buffer amplifiers for buffering the voltages developed at the outputs of said first, second and third low-pass filters, respectively;
first, second and third inverters for inverting the voltages at the outputs of said first, second and third buffer amplifiers, respectively;
first and second negative half-wave rectifiers for receiving and rectifying the outputs of said first buffer and said first inverter respectively, to provide first and second control voltage signals at first and second output terminals respectively;
third and fourth negative half-wave rectifiers for receiving and rectifying the outputs of said second buffer and said second inverter respectively, to provide third and fourth control voltage signals at third and fourth output terminals respectively;
fifth and sixth negative half-wave rectifiers for receiving and rectifying the outputs of said third buffer and said third inverter respectively, to provide fifth and sixth control voltage signals at fifth and sixth output terminals respectively;

first, second and third absolute differencing circuits for producing the absolute value of the difference voltage between the inputs and outputs of said first, second and third low-pass filters, respectively;
a summing amplifier for summing the outputs of said first, second and third absolute differencing circuits;
a low-pass filter for smoothing the output of said summing amplifier;
a pulse oscillator for providing high frequency rectangular pulses;
a pulse shaper for providing from said rectangular pulses a train of pulses suitably shaped for pulse width modulation; and a pulse width modulator for generating rectangular pulses and modulating the width of the pulses in response to the output voltage of said low-pass filter;
said servologic circuit being operative to filter the said first, second and third direction signals with a variable time constant inversely dependent upon the rate of change of the said first, second and third direction signals, and toprovide therefrom six negative-going smoothly varying control voltage signals.

15. The servologic circuit of claim 14 wherein said summing amplifier for summing the outputs of said absolute differencing circuits receives said first absolute difference signal derived from said front left-right direction signal at one half the level of said second and third absolute difference signals derived respectively from said modified front-back direction signal and said back left-right direction signal.

16. The servologic circuit of claim 14 wherein said first, second and third low-pass variable filters are of two-pole type, each comprising:
first resistor in parallel with a voltage-controlled switch, connected to the input of said filter;
second resistor in series with said parallel combination of said first resistor and said voltage-controlled switch to which a high frequency rectangular wave of variable duty ratio is applied as a control signal;
first capacitor from the output of said second resistor to ground, forming a first variable time constant with said first and second resistors and said voltage-controlled switch, and varying with the duty ratio of the high frequencyrectangular wave control signal applied to said switch;
third resistor connected from the junction of said first capacitor and said second resistor to the output of said filter; and a second capacitor from the output of said filter to ground, forming a second fixed time constant with said third resistor.

17. The processor of claim 1 wherein said direction detector circuit comprises first and second log-ratio detectors, each comprising:
first and second input terminals;
first and second symmetrical logarithmic amplifier for receiving and amplifying the signal currents applied to said first and second input terminals,respectively, such that the instantaneous amplitude of the output signal voltages vary in proportion to the logarithms of the instantaneous input signal currents;
first and second inverters for inverting the outputs of said first and second logarithmic amplifiers respectively;
first negative full-wave rectifier for generating a negative-going full-wave rectified signal from the outputs of said first logarithmic amplifier and said first inverter;
second positive full-wave rectifier for generating a positive-going full-wave rectified signal from the outputs of said second logarithmic amplifier and said second inverter;
first and second two-pole filter circuits for smoothing the output signals from said first and second full wave rectifiers, respectively; and a summing amplifier for receiving the outputs of said first and second filter circuits in opposite polarities and summing them to produce a direction signal at its output, connected to an output terminal;
said summing amplifier having a series pair of back to back matched zener diodes in parallel with a feedback resistor to provide symmetrical limiting of the output signal as the directional information content of said left and right audio signals varies;
said first log-ratio detector comparing said left and right filtered signal amplitudes, and said second log-ratio detector comparing said sum and difference signal amplitudes, to provide a left-right and a front-back directionsignal at their outputs, respectively.

18. The processor of claim 1 wherein said plurality of voltage controlled amplifiers each comprises:
first and second input terminals for receiving one or both of said left and right signals from said input matrix circuit in either normal or inverted polarity;
voltage-controlled variable attenuator comprising a junction field-effect transistor in conjunction with a series variable resistor and one or more resistors connected to the said first and second input terminals;
inverting amplifier for amplifying and inverting the signal from said variable attenuator;
summing inverting amplifier for summing the signals from said first and second input terminals and the output of said inverting amplifier in suitable proportions such that when said variable attenuator has minimum attenuation the output of said inverting amplifier cancels the signals from said first and second input terminals, the output of said inverting summing amplifier being connected to a first output terminal;
unity gain inverter for inverting the signal at said first output terminal, the output of said inverter being connected to a second output terminal;
control amplifier for providing at its output a positive-going control voltage with a negative bias for controlling the attenuation of said voltage controlled attenuator by applying said output voltage to the gate of said junction field-effect transistor, said control amplifier having a virtual groundinverting input to which a positive bias current is provided, having a feedback resistor connected between its output and said inverting input, and also having a non-inverting input;
voltage divider comprising two resistors in series from the drain of said junction field-effect transistor to ground, the junction between said two resistors connected to the said non-inverting input of said control amplifier, for compensation of distortion due to the square-law nonlinearity of said junction field-effect transistor;

control terminal for receiving one of said plurality of control voltage signals; and linearity correction circuit for providing an input current to the inverting input of said control amplifier that varies in a nonlinear manner with the control voltage signal applied to said control terminal.

19. The voltage controlled amplifier of claim 18 wherein said linearity correction circuit comprises:
first resistor between said control terminal and said inverting virtual ground input of said control amplifier;
zener diode, its anode connected to said control terminal;
second resistor connected between the cathode of said zener diode and ground; and third resistor connected from the cathode of said zener diode to the cathode of a junction diode, the anode of said junction diode being connected tosaid inverting virtual ground input of said control amplifier;
so that for negative voltages applied to said control terminal that are smaller than the breakdown voltage of said zener diode, the current flowing out of said inverting input is proportional to the applied voltage, but for larger negative voltages the current increases at a substantially greater rate.

20. The processor of claim 1 wherein said detector splitter circuit further comprises a switch or switches for reconfiguring the circuitry therein which provides said additional direction signals, and wherein said output matrix circuit comprises:
a plurality of matrixing circuits each composed of a plurality of resistors connected to a common output point, the other terminals of each said resistor being connected to different ones of the outputs of said plurality of voltage controlled amplifiers or to different ones of the outputs of said input matrix circuit; and in one or more of said plurality of matrixing circuits, an additional resistor or additional resistors connected from said common point through a switch or switches operated conjointly with the said switch in said detector splitter circuit to one or to different ones of the outputs of said plurality ofvoltage controlled amplifiers or of said input matrix circuit;
so as to provide alternate configurations of said output matrix corresponding to the alternate configurations provided by said switch or switches in said detector splitter circuit.

21. The processor of claim 1 wherein said detector splitter circuit further comprises a switch or switches for reconfiguring the circuitry therein which provides said additional direction signals, and wherein the proportions ofthe outputs of said input matrix circuit which are received by said voltage controlled amplifiers are changed by means of switches operated conjointly with said switch or switches in said detector splitter circuit;
so as to provide alternate configurations of said voltage controlled amplifiers driving said output matrix corresponding to the alternate configurations provided by said switch or switches in said detector splitter circuit.