CA1310376C - Current gain compensation amplifier - Google Patents

Abstract

RCA 83,957A
CURRENT GAIN COMPENSATION AMPLIFIER

ABSTRACT
A first pair of transistors, Q721 and Q723, operating, each, as an emitter follower, has a relatively low emitter current. The emitter followers drive corresponding base electrodes of transistors of a second pair of transistors, Q148 and Q149, coupled as a differential amplifier. A current gain compensating transistor Q726 has a current gain that tracks current gain variations of each of the transistors of the second pair. A base electrode current of the compensating transistor and a second current iQ727 that is unaffected by current gain variations are summed. The summed current is coupled via transistors Q722 and Q724 of a current mirror arrangement to first and second junction terminals, respectively coupled between the respective emitter electrodes of the emitter followers and the base electrodes of the second pair. Currents generated by the current mirror arrangement compensate for variations in the base currents in the transistors of the second pair so as to maintain the emitter currents in the emitter followers substantially unaffected by these variations.

Description

-1- RCA 83, 957A

CURRENT GAIN COMPE~7~;ATION A~PLIFIE~
The invention relates to an amplifier arrangement that combines an input siynal with a reference signal to produce an output signal that is level shifted relative to that of the input signal. In particular, the invention relates to a level shifter used in an input stage of a power supply regulator of a television apparatus.
In a power supply arrangement o~, ~or example, a television receiver, a voltage representative of a DC, regulated supply voltage B+ is coupled to, for example, an inverting input terminal of a B+ voltage regulator. A
reference voltage is coupled to, for example, a noninverting input terminal of the regulator. The feed-back voltage that is representative, for example, of regulated supply voltage B+ is compared, in the input stagP, with the reference voltage to generate an output voltage that is coupled to a controllable arrangement. The controllable arrangement regulates voltage B+ at a level that is determined by the reference voltage. Voltage B+ may be used, for example, to energize a deflection circuit output stage for a cathode ray tube (CRT).
In one prior art circuit, a voltage representative of, for example, a level of the beam current in the CRT is summed with the reference voltage, such that the sum of both, instead of the reference voltage alone, is applied to the noninverting input terminal of the input stage, a variation of the beam current representative voltage varies voltage B~ so as to maintain constant the raster width in the C~T when the beam current changes.
The summation of the constant reference voltage and the variable beam current representative voltage is done, in such prior art circuit, by coupling the beam current representative voltage via a zener diode to the noninverting input terminal. The zener diode develops between its anode and cathode electrodes the referenc~
voltage. Thus, the reference voltage that level shifts the A :~--2- 1 3 1 0 37 ~CA ~3,9~7 beam current representative voltage is series coupled with the beam current representative voltage.
A disadvantage of such zener diode level shifting is that a change in the zener voltage due to temperature, aging or an inherent noise in the zener diode, causes the level to change.
It may be desirable to sum the beam current repres~ntative voltage with a reference voltage to generate a sum voltage that ls applied to the noninverting input terminal of the input stage without using such zener diode.
In an embodiment of the invention, the reference voltage is produced by a voltage source of, for example, the well known bandgap type. The beam current representative voltage is summed with, or level shifted by, the reference voltage that is produced by the bandgap type source. Advantageously, the bandgap type source is less susceptible than the zener diode to temperature changes, component aging or noise.
In a circuit embodying an aspect of the invention, an input signal, such as a beam current representative voltage, is coupled in series with an emitter electrode of a first transistor that operates as a common base amplifier. A second transistor, operating as a current source, and having its collector coupled to the collector of the first transistor, supplies the collector current of the first transistor. A current mirror arrangement, that includes the first transistor, causes the collector current in the first transistox to be equal to that supplied by the collector current of the second transistor, when the input signal is at a predetermined magnitude such as zero. A first voltage, such as produced by a bandgap type source, is coupled via a resistor to a junction terminal between the collectors of the first and second transistors. The collector currents of the first and second transistors produce a difference current that develops a voltage across the resistor that is summed with the first voltage to produce a second signal that is -3~ t310376 RCA 83,9~7 developed at the junction terminal. The second signal is at a magnitude that is determined by the first voltage and not by the collector currents in any of the first and second transistors, ~hen the magnitude of the input signal is predetermined such as zero. When the magnitude of the input signal is different from the predetermined magnitude, the second signal is different from the first voltage by an amount that is proportional to the magnitude of the input signal.
In accordance with an aspect of the invention, a pQWer supply includes a level shifter level that shifts an input voltage used for controlling an output supply voltage of the power supply. A controllable conductive element is coupled to the input supply voltage for generating, from the input supply voltage, the output supply voltage. The power supply includes a comparator, coupled to the conductive element, for varying, in accordance with an output signal of the comparator, the conduction of the conductive element to control the output supply voltage. A
current mirror arrangement that in~ludes a ~ransistor i~
responsive to a current in a first circuit branch for generating in a first main current conducting electrode of the transistor a current that is the current mirror cf the current in the first circuit branch. The first main curren~
conducting electrode is coupled at a junction terminal to a second circuit branch for conducting at least a portion of a current in the second circuit branch. A source of the input voltage is coupled to the transistor for varying the current in the first main current conducting electrode to produce a difference related current that varies in accordance with the input voltage. The difference current is related to a difference between the current in the first main current conducting electrode and the current in the second circuit branch. A first resistance is coupled to the junction terminal for conducting the difference related current to develop a voltage across the resistance that varies in accordance with the input voltage. A source o~
temperature compensated first voltage is coupled via the ~4~ 1 31 037~
RCA ~3,9~7 resistance to the junction terminal such that the voltage across the resistance is combined with the first voltage for developing a temperature compensated second voltage at the junction terminal. The second voltage varies in accordance with the input voltage. The second voltage is level shifted in accordance with the first voltage. The temperature compensated, level shifted, second voltage is coupled to an input of the comparator for varying the output signal of the comparator to control the output supply voltage~
In the Drawing:
FIGURE 1 illustrates a simplified schematic diagram of a power supply regulator circuit that includes a level shifter, embodying an aspect of the invention;
FIGURE 2 illustrates a detailed schematic diagram of the level shifter of FIGURE 1; and FIGURES 3, 4, 5 and 6 illustrate level shifters embodying different aspects of the invention, respectively.
FIGURE 1 illustrates a simplified schematic diagram of a power supply of a television receiver, not shown in the FI~URES, that includes a regulator 100 that is an integrated circuit that regulates a supply voltage B+.
Voltage B+ may be used, for example, to energize a horizontal deflection circuit or output stage 99 of the television receiver.
A voltage V-~, representative of voltage B+, is obtained from output stage 99. Voltage V+ is coupled to a voltage divider 605 that includes series coupled resistors 601, 604 and 602. Resistor 604 includes a wiper k for developing at wiper k a voltage that is representative of, for example, voltage B+. The voltage at wiper k, that is adjustable by varying the position of wiper k, is coupled to an inverting input terminal 608 of an error amplifier 610 via a resistor 607.
A small voltage that is proportional to the beam current in the CRT of the receiver is coupled from a 1 3 1 0376 RCA ~3,957 -tertiary wlnding of transformer T to a terminal 611 to form a voltage VNINI that is indicative of the beam current.
Voltage VNINI that varies when a variation of the beam current occurs, is coupled via a level shifter 600, embodying an aspect of the invention, to a noninverting input terminal 609 of error amplifier 610 to produce an input voltage VNIN. Level shifter 600 establishes a fixed offset voltage between terminals 611 and 609 that is determined by a voltage VBG. Voltage VBG
is generated in a bandgap type voltage source 699. Bandgap type voltage source 699 advantageously maintains voltage VBG constant such that voltage VBG is affected significantly less by component aging or tolerance than would have occurred had a zener diode been used. As explained later on the feed back arrangement of regulator 100 causes voltage B~ to be such that voltage VIN becomes equal to voltage VNIN.
An integrating filter 612 is coupled between inverting input terminal 608 and an output terminal 618 of amplifier 610 to pr~vide the loop filter of reaulator 100.
A filtered error voltage V0, developed at terminal 618 is coupled to a first input terminal of an adder 613. A
horizontal rate sawtooth generator 98 develops a horiæontal rate signal, having an upramping portion, which is added to error voltage V0 in adder 613. The sum signal that is also upramping, is applied to an inverting input terminal ~614~
of a comparator 615 functioning as a pulse width modulator.
When, during its upramping portion, the output of adder 613 becomes more positive than a constant DC voltage VREF, that is coupled to a noninverting input terminal of comparator 615, a negative going transition at an output terminal 615a of comparator 615 is coupled via a buffer amplifier 616 to a control terminal 617a of a switch 617b of a switch mode power supply output stage 617 to turn on switch 617b of output stage 617.
An input terminal 617c of output stage 617 is coupled to unregulated voltage VuR. Regulated voltage B+

-6-t 3 1 ~37 ~
RCA ~3,9~7 is developed at an output terminal 617d of output stage 617.
The duration, during each horizontal period, H, in which switch 617b conducts is determined by the level of error voltage V0 of error amplifier 610. Thus, regulatèd voltage B+ is determined by voltage VNI~. As indicated before, voltage VNIN is produced by level shifter 600, embodying an aspect of the invention, that is described now in detail.
FIGURE 2 illustrates a schematic diagram of level shifter 600 of FIGUP~ 1 and of error amplifier 610.
Similar numbers in FIGURES 1 and 2 represent similar items or functions. Level shifter 600 of FIGURE 2 is temperature compensated over a wide range of ambient operating temperatures, such as between 0C and 70C, to produce voltage VNIN that is suhstantially unaffected by a change in the temperature within such range.
A temperature compensated current control arrangement 650 generates a control voltage VBR on a rail signal line 900. Rail signal line 900 is coupled to the base electrode of each of transistors Q142, Q725, Q727, Q736 and Q737. The emitter electrodes of the above-mentioned transistors are coupled through corresponding resistors to a fixed DC voltage Vcc. Current control arrangement 650 controls voltage VBR in such a way that the collector current in each of the above-mentioned transistor stays substantially constant when the temperature changes. An example of an arrangement that is similar to current control arrangement 650 is described in detail in U.S. Patent No. 3,886,435, in the name of S. A.
Steckler, entitled VBE VOLTAGE SOURCE TEMPERATURE
COMPENSATION NETWORK.
Level shifter 600 includes transistors Q736 and Q737. The emitter currents in transistors Q736 and Q737 are controlled by resistors R728 and R69, respectively, having the same value, so as to cause the respective collector currents of transistors Q736 and Q737, that are temperature compensated, to be equal. The collector of 1 31 0376 ~CA z3,957 transistor Q737 is coupled to a current mirror arrangement that includes transistors Q733, Q734 and Q735. The collector of transistor Q737 is coupled to the collector of transistor Q734. The emitter of transistor Q735 is coupled to each of the bases of transistors Q733 and Q734.
Transistor Q735 provides the base current drive to each of transistors Q733 and Q734. The emitter of transistor Q734 is coupled to ground via a resistor R732. The P-N junctiun of transistor Q734 be~ween the base and emitter electrodes of transistor Q734, provides temperature compensation that compensates for a temperature related variation of the base-emitter voltage of transistor Q733. The emitter of transistor Q733 is coupled through resistor R731 to terminal 611, where voltage VNINI of FIGURE 1 is developed~.
The value of resistor R731 is equal to that of resistor R732- Voltage VNINI is prevented from exceeding predetermined limits in either polarity by a diode network 675. The collector of transistor Q733 is coupled to the collector of transistor Q736 at a junction terminal 733A.
Assume th2t volta~e VNINI is zero. In this cas~, the current mirror arrangement of transistors Q733, Q734-and Q735, produces a collector current iQ733 in transistor Q733 that is equal to the collectox current iQ734 in transistor Q734 because the base current of transistor Q735 is negligible. As explained before, when voltage VNINI is zero, collector current iQ736 in transistor Q736 is equal to collector current iQ737 in transistor Q737 over a wide.
temperature range. Also, when voltage VNINI is zero, each of collector current iQ733 that is the current mirror of.
curren~ iQ734 is equal to current iQ737 over such wide temperature range. It follows that current iQ733 is also e~ual to current iQ736~
Bandgap type voltage source 699 supplies temperature compensated reference voltage VBG that is coupled via a resistor R729 to terminal 733A. Because, as described before, when voltage VNINI is zero, current i is supplied entirely by current iQ736~ and because the impedance at terminal 733A, that is contributed by the 1 3 1 Q376 RCA 83,957 collectors of tr~nsistors Q733 and Q736 is high, a current iR729 in resistor R729 is zero; therefore, voltage VNIN at tenminal 733A is equal to voltage vB&. Thus, in accordance with an aspect of the inventio~, when voltage VNINI is zero, voltage VNIN is level shifted by an amount that is equal to voltage VBG~
When voltage VNINI at terminal 611 is different from zero, currents iQ736 and iQ737 will not be equal. The difference current between currents iQ733 and iQ736 will cause a voltage to develop across resistor R729 that, i~
turn, will cause a corresponding change in voltage VNIN at terminal 733A. Because transistor Q733 is coupled, relative to voltage ~ INI~ as a common base amplifier, and because resistors R731 and R72g are, illustratively, equal, the gain, or the ratio between voltage VNIN and voltage VNINI, is one, resulting in an amplifier having a unity gain.
In carrying out another aspect of the invention, voltage VNIN, that is level shifted relative to voltage VNI~I by an amount that is equal to voltage VBG, follows v~riations of voltage VNINI that occur in a range betwee~
positive and negative values.
Voltage VB& i5 temperature compensated and has a tolerance range that is narrow relative to, for example, a zener diode. Furthermore, component aging affects voltage VBG substantially less than it affects, for example, the breakdown voltage of a zener diode. Moreover, the level shifting caused by level shifter 600 is, advantageously, less susceptible to temperature, aging and noise when compared with that produced by a corresponding level shifter in the prior art that utilizes a zener diode interposed between a beam current input terminal and a noninverting input terminal of a differential amplifier to perform such level shifting.
Should a temperature change cause a corresponding change in current iQ736~ for example, that, as ind~cated before, would be relatively small, transistors Q737, Q733, Q734 and Q735 will cause a proportional change in current 1 31 0376 RCA 83,957 iQ733 to occur that will prevent even such small change in temperature from affecting the difference current between currents iQ736 and iQ733. Therefore, when voltage VNINI is zero, voltage VNIN is, advantageously, not affected by collector currents iQ736 and iQ737~
detenmined by voltage VBG that is temperature compensated.
It should be understood that temperature compensation may be adequate even when voltage VNINI is significantly different from zero. If temperature compensation, in this case, is inadeguate, a further improvement in temperature compensation may be obtained by coupling the terminal of, for example, resistor R732, that, in FIGURE 2 is grounded, to a voltage that is different from zero and that is related to, for example, voltage VNINI
Advantageously, voltage VBG, as explained before, is maintained at tight tolerances, is temperature compensated and is substantially unaffected by components aging. Therefore, advantageously, no factory temperature burn-in process is required prior-to the installment of regulator 100 of FIGURE 1 in the television receiver.
Furthermore, voltage divider 605 that includes resistors 601, 604 and 602 is required to compensate, advantageously, only for a narrower tolerance range than in prior art circuits in which a zener diode is used for performing the level shifting function of level shifter 600 of FI~URE 2.
Voltage VIN is coupled to the base of a transistor Q721 The clamping operation of a pair of trans~stors Q145 and Q146 prevents voltage VIN from being above voltage VBG or from being below voltage VB~ by more than a predetermined magnitude. Voltage VIN is coupled to inverting input terminal 60~ and voltage VNIN is coupled to noninverting input terminal 609 of error amplifier 610.
Amplifier 610 includes a current source formed by a transistor Q142 that provides the combined emitter currents of a transistor Q148 and of a transistor Q149, coupled as a differential amplifier. The bases of transistors Q148 and Q149 are coupled to the emitters of transistors Q721 and -10- RC~ 83,957A

Q723 respectively. Transistors Q721 and Q723 operate as emitter followers to couple voltages VIN and VNIN to the base of transistors Q148 and Q149, respectively.
A transistor Q722 has a collector electrode that is coupled between the base and emitter electrodes of transistors Q148 and Q721, respectively. A collector current ic of transistor Q722 is equal to a sum of a base current ibQ148 of transistor Q148 and of an emitter current ie f transistor Q721 for supplying both the base current and the emitter current, respectively. The base electrode of transistor Q722 is coupled to the collector of transistor Q730 for varying collector current ic in a manner that compensates for current gain changes in transistor Q148, as described later on. The collector electrode of transistor Q730 is coupled as a diode, that forms with transistor Q722 a current mirror arrangement.
Transistor Q727, having a base electrode that is coupled to voltage VBR generates a temperature compensated collector current that is unaffected hy current gain variation of transistor Q727, as indicated before. A current iQ727 is coupled also to the collector terminal of transistor Q730 such that collector current iQ727 in transistor ~727 provides a first portion of the collector current of transistor Q730.
In accordance with a feature of the invention, the first portion is independent of current gain variations or deviations of transistor Q727. A second portion of the collector current of transistor Q730 is provided by a base current ibQ726 f a transistor Q726 that is summed with current iQ727 to form a sum current iSum that is coupled to the collector terminal of transistor Q730. The second portion that provides current gain compensation is dependent on the current gain of transistor Q726.
The current gain of transistor Q726 follows or tracks in the same sense changes or variations of the A

- lOA- RCA 83,957A

current gain of transistor Q148, occurring due to, for example, temperature or tolerances. This is accomplished, for example, by constructing the two P-N-P transistors r~Jith the same geometry using a similar process and by 5 maintaining the operating temperature of the two transistors the same.
In steady state operation of the differential am~lifier, the emitter current in each of transistors Q148 and Q726 is substantially the same. The nominal emitter currents in transistors Q148 and Q726 are, each, for example, 50 microamperes, as controlled via transistors Q142 and Q725, respectively. As explained before, the collector currents in transistors Q142 and Q725 that are controlled by voltage VBR are temperature compensated. A
deviation in base current ibQ148 will be accompanied with the same sense deviation in the base current ibQ726 of transistor Q726 and, therefore, also in the collector current of transistor Q730. Because of current mirror operation formed by the arrangement that includes transistors Q722 and Q730, a change in base current ibQ726 of transistor Q726 will cause substantially the same sense change in the collector current of each of transistors Q730 and Q722.
Therefore, in accordance with another feature of the invention, the change in base current ibQ148 of transistor Q148 will be, advantageously, compensated by the corresponding equal change of the collector current of transistor Q722. The change in the collector current of transistor Q722 is in the same sense so as to prevent the emitter current of transistor Q721 from changing or deviating from its nominal value when base current ibQ148 of transistor Q148 changes. The emitter current of transistor Q721 will be determined by collector current iQ727. Current iQ727 is independent of current gain variations, as explained before.

t 3 1 0376 -lOB- RCA 83,957A

In accordance with an aspect of the invention, transistor Q722 that is included in the current mirror arrangement generates both a first current portion and a second current portion. The first current portion maintains the emitter current of transistor Q721, advantageously, independent of the current gain of transistor Q148 and the second current portion supplies the base current of transistor Q148. It should be understood that since transistors Q722, Q724 and Q730 are of the N-P-N
type, having a high current gain, their base currents may be ignored in this analysis.
Similarly, a change in the base current of transistor Q149 will be compensated by a corresponding change in a collector current of a transistor Q724. The collector current of transistor Q724 is controlled in the same manner as that of transistor Q722.
The emitter currents of each of transistors Q721 and Q723 are, each, for example, not significantly larger than each of the base currents of transistors Q148 and Q149, respectively. The emitter currents in transistors Q721 and Q723 are, advantageously, maintained substantially unaffected by variation or change of the current gain of transistors Q148 and Q149. As explained before, such changes and variations may occur due to temperature or tolerances. Therefore, advantageously, the offset voltage of amplifier 610 is less affected by current gain changes of transistors Q148 and Q149. It follows that voltage regulation tolerances in the power supply are improved.
Additionally, the current gain compensation will, advantageously, prevent even an excessive current ibQ148, for example, from cutting off transistor Q721.
In accordance with another aspect of the invention, deviations of the current gain characteristic of transistors Q148 and Q149 are sensed in transistor Q726 that is coupled outside the signal path in the differential -lOC- RCA 83,957 amplifier. Transistor Q726 is also coupled outside the current paths in each of transistGrs Q148 and Q149 that form the differential amplifier.
A current mirror arrangement 610 that is coupled to the collectors of transistors Q14~ and Q149 that forms the differential amplifier causes a current i610/ coupled to integrating filter 612 of FIGURE 1, to be equal to the difference between the collector currents in transistors Q148 and Q153. Consequently, current i6lo that is coupled to filter 612 of FIGURE 1, is proportional to the difference between voltages VIN and VNIN- The proportionality factor is determined by the gain of error amplifier 610 that is determined by transistors Q14~ and Q149.
A current mirror arrangement 610b that is coupled to the collectors of transistors Q148 and Q149 causes a current i610/ coupled to integrating filter 612 of FIGURE
1, to be equal to the difference between the collector currents in transistors Q143 and Q153. Consequently, current i610 that is coupled to filter 612 of FIGURE 1, is proportional to the difference between voltages VIN and VNIN. The proportionality factor is determined by the gain of error amplifier 61~.
FIGURES 3, 4, 5 and 6 illustrate level shifters 60Oa, 6~Ob, 600c and 60Od, respectively, embodying other aspects of the invention, respectively. In FIGURES 2-6, numbers and symbols of similar items or functions are similar except that they include the letters a, b, c and d, in FIGURES 3, 4, 5 and 6, respectively.
In FIGURES 3, resistors R731 and R732, that are used in the circuit of FIGURE 2, were eliminated. Without resistors R731 and R732, the gain of amplifier 600a of FIGIJRE 3 is, advantageously, higher than unity.
In FIGURE 3, a diode DCma and a temperature compensated current source Ila cause the collector A

~ RCA 83,957A

currents in transistors Q737a and Q736a to be, for example, e~ual. Similarly, transistor Q734a causes the collector current in transistor Q733a to be, for example~ equal to that in each of transistors Q737a and Q736a when voltage VNINI is zero.
In FIGURE 4, voltage VNINIb is applied differentially between the Pmitters of transistors Q733b and Q734b. Such arrangement provides, advantageously, an improved common mode rejection.
In FIGURE 5 the type, N-P-N or P-N-P, of the corresponding transistors is opposite than that in FIGURE 2 so that voltage VNINIC may, if desired, be reference to voltage Vccl instead of to ground.
In FIGURE 6, the input impedance to voltage VNINId is, advantageously, higher than to voltage VNINI of FIGURE
2 because of the usage of a transistor Q750 that is coupled as an emitter follower.

Claims

-12- RCA 83,957A

The embodiments of the invention in which an exclusive property or privilege is claimed are defined as follows:
1. An amplifier, comprising:
first and second transistors that are coupled to form a differential amplifier;
an input stage coupled at a first junction terminal to a control electrode of said second transistor for establishing a voltage level at said control electrode of said second transistor in accordance with a voltage that is developed at an input terminal of said input stage such that an output current flowing in said input stage is coupled to said first junction terminal;
a source of a first current;
a third transistor having a current gain characteristic that is representative of a current gain characteristic of said second transistor for generating a control electrode current of said third transistor that is combined with said first current to produce a combined second current, said second current having a first portion derived from said first current and a second portion derived from said control electrode current of said third transistor; and a current mirror arrangement responsive to said second current for generating in said current mirror arrangement a third current that is coupled to said first junction terminal, said third current having a first portion derived from said first current of generating said output current of said input stage and a second portion derived from said control electrode current of said third transistor for generating a control electrode current of said second transistor, wherein a deviation of said current gain characteristic of said second transistor from a nominal value thereof -13- RCA 83,957A

that produces a corresponding deviation in said control electrode current of said second transistor is compensated by said control electrode current of said third transistor so as to substantially prevent such current deviation from affecting said output current of said input stage.
2. An amplifier according to claim 1 wherein said input stage comprises a fourth transistor coupled as an emitter follower and said output current of said input stage flows in an emitter electrode of said forth transistor.
3. An amplifier according to claim 1 wherein said first current generating means generates said first current having a value that is independent of said variation in said current gain characteristic of at least one of said second and third transistors.
4. An amplifier according to claim 1 wherein said current mirror arrangement presents said deviation in said current gain characteristic of said second transistor from affecting an offset voltage at said input terminal of said input stage.
5. An amplifier according to claim 1 further comprising, second means coupled to a second junction terminal between corresponding main current conducting electrodes of said first and second transistors for generating a current that varies in the same sense as said first current independently of said current gain characteristic of said second transistor.
6. An amplifier according to claim 1 wherein said first, second and third transistors are bipolar transistors of the same type.
7. An amplifier according to claim 1 wherein said current mirror arrangement includes a fourth transistor that generates said third current, said third current flowing in the same direction as said first -14- RCA 83,957A

current and being equal to the sum of said first current and said control electrode current of said third transistor.
8. An amplifier according to claim 1 wherein said third transistor is coupled outside a signal path formed between an input terminal and an output terminal of said amplifier.
9. An amplifier according to claim 1 further comprising means for generating a temperature compensated control voltage that is coupled to a control electrode of a bipolar, fourth transistor for generating said first current at a collector electrode of said fourth transistor that is unaffected by a variation of a current gain characteristic of said fourth transistor.
10. An amplifier according to claim 9 wherein said current mirror arrangement further comprises a fifth transistor coupled in a diode configuration for conducting therethrough a current that is equal to a sum of said first current and said control electrode current of said third transistor.
11. An amplifier according to claim 10 wherein said current mirror arrangement includes a sixth transistor having a base electrode that is coupled to said fifth transistor and a collector electrode that is coupled to said control electrode of said second transistor.
12. An amplifier according to claim 11 comprising a seventh transistor having a base electrode that is coupled to said fifth transistor and a collector electrode that is coupled to a base electrode of said first transistor.
13. An amplifier according to claim 1 wherein said amplifier forms an error amplifier in a regulator of a television apparatus power supply.