CA1296098C - Phased array antenna with couplers in spatial filter arrangement - Google Patents

Abstract

PHASED ARRAY ANTENNA WITH
COUPLERS IN SPATIAL FILTER ARRANGEMENT

ABSTRACT OF THE INVENTION

A lossless spatial filter having N input ports and N output ports and printed on a single substrate. The filter is used in combination with an antenna system which radiates wave energy signals into a selected angular region of space and into a desired radiation pattern. The aperture of the system includes a plurality of N antenna elements. The antenna elements are arranged along a predetermined path and each element is connected to only one output port of the spatial filter. A beam steering unit controls the direction of radiation. A signal generator supplies a power divider having N output signal ports each connected to a phase shifter controlled by the beam steering unit.

Description

1 BhCKGROUND OF THE INVENTION

2 Field of the Invention 3 This invention relates to array antenna 4 systems and particularly to such systems wherein the antenna element pattern is modified by providing a 6 lossless spatial filter between the antenna input ports 7 and the antenna elements so that the effective element 8 pattern associated with each input port is primarily 9 within a selected angular region of space.
_escription of the Prior Art 11 An array antenna system may be designed to 12 transmit a desired radiation pattern into one of a 13 plurality of angular directions in a selected region of 14 space. In accordance with the prior art designs of such array antennas, each of the antenna elements has an 16 associated input port. By variation of the amplitude 17 and/or phase of the wave energy signals supplied to the 18 input ports, the antenna pattern can be electronically 19 steered in space to point in a desired radiation direction or otherwise controlled to radiate a desired 21 slgnal characteristic, such as a time reference beam 22 scanning pattern. When it is desired to have an array 23 antenna radiate its beam over a selected limited region 24 of space, it is preferable that the radiation pattern of the individual antenna elements also be primarily within 26 the selected angular region. This permits maximum 1 element spacing while suppressing undesired grating 2 lobes.
3 In certain systems, control of the element pattern 4 by modification of the physical shape of the antenna element may be impractical because of a desired element 6 pattern may require an element aperture size which 7 exceeds the necessary element spacing in the array. A
8 practical approach to overcome the physical elements size 9 limitation is to provide networks for interconnecting each antenna input port with more than one antenna 11 element, so that the effective element pattern associated 12 with each input port is formed by the composite radiation 13 of several elements. These networks can be realized by 14 printed circuit techniques using a single substrate layer.
16 One prior art approach to this problem has been 17 described by Nemit in U.S. Patent No. 3,803,625. Nemit 18 achieves a larger effective element size by providing 19 intermeidate antenna elements between the primary antenna elements and coupling signals from the primary antenna 21 element ports to the intermediate element ports. This 22 tapered multielement apertyure excitation produces some 23 measure of control over the radiated antenna pattern.
24 A more effective prior art antenna coupling network is described by Frazita et al. in U.S. Patent No.
26 4,041,501 assigned to the same assignee as the present , ~,,, ,.,, ~

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1 invention. According to the technique oE Fra7.ita, the 2 antenna elements are arranged in element modules, each 3 module is provided with an input port. Transmission 4 li.nes are coupled to all of the antenna element modules in the array. The transmission lines couple signals 6 applied to any of the ports to selected elements in all 7 the antenna element modules of the array. This antenna, 8 herein referred to as a COMPACT antenna, provides an 9 effective element aperture which is coextensive with the array aperture.
11 Still another effective prior art antenna coupling 12 ne-twork is described by Wheeler in U.S. Patent No.
13 4,143,379 assigned to the same assignee as the present 14 invention. According to the technique of Wheeler, cross coupling ports are employed to couple wave energy signals 16 to modules which are contiguous to each module.
17 Yet, another technique is shown in U.S. Patent No.
18 4,168,503 which describes an antenna array with a printed 19 circuit lens in a coupling network. A radiated signal, received by each one of a plurality of spatially 21 separated antennas forming a directive array, is 22 coherently recovered by the lens. The lens comprises a 23 plurality of verticlaly standing and circularly arranged 24 printed circuit panels, each of which includes a conductor strip connected at one end to each antenna. A

1 pluralit,y of semi-el~iptical circuit panels are affixad 2 to the vertical panels at a predetermined angle. Metal 3 strips plated on the semi~elliptical panels provide the 4 desired time delay to the antenna signa]s. A combining strip couples the time delay strips and provides a 6 combined output signal at one end of the semi-elliptical 7 pattern. The angle at which the semi-elliptical boards 8 are a~ixed to the vertical boards corrects for time 9 delay distortion caused by the placement of the combining strip. This configuration cannot be 11 implemented using printed circuit techniques on a single 12 substrate layer.
13 U.S. Patent No. 4,321,605 describes an array 14 antenna system ~laving at least a 2:1 ratio of antenna elements to input terminals interconnected via primary 16 transmission lines. Secondary transmission lines are 17 coupled to and intersecting a selected number of the 18 primary transmission lines. Signals supplied to any of 19 the input terminals are coupled primarily to the elements corresponding to the input terminal, and are 21 also coupled to other selected elements.
22 In time reference scanning beam systems such as 23 microwave landing systems (MLS)~ there may be a 24 linearity requirement for the glide path guidance i.e., :

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-1 the difference between the actual and indicated angle 2 must be within a limited range. There is also a 3 requirement to minimize the field monitor distance for 4 the glide path antenna. Particularly in MLS, this invention provides a non-thinned or fully filled array 6 which may be used to achieve linearity and minimize the 7 field monitor distance.

9 It is an object of the present invention to provide an alternate array system having an antenna 11 element pattern formed by a spatial filter between the 12 antenna element input ports and the antenna elements.
13 It is another object of this invention to provide 14 a non-thinned antenna system i.e., an antenna system wherein the number of antenna input ports equals the 16 number of antenna element output ports so that there is 17 no reduction ratio in the number of radiators to the 18 number of phase shi~ters.
19 It is another object of this invention to provide an antenna system which does not generate grating lobes.
21 It is still another object of this invention to 22 provide a lossless spatial filter having a 1:1 23 input/output ratio which employs a minimum number of 24 couplers and terminations.
It is another object of this invention to provide ; 26 a lossless spatial filter having flexibility in ~ -6-: ::

1 controlling the spatial filter radiation pattern, 2 meeting linearity requirements and minimizing field 3 monitor distances.
4 In accordance with the invention, the antenna system radiates wave energy signals into a selected 6 angular region of space and into a desired radiation 7 pattern. The system includes a lossless spatial filter 8 having N input ports and N output ports. The aperture 9 of the system comprises a plurality of N antenna elements. The antenna elements are arranged along a 11 predetermined path and each element is connected to only 12 one output port of the spatial filter.
13 A beam steering unit controls the direction of 14 radiation and includes N phase shifters and means for controlling of phase shif-ters. Each phase shifter has a 16 phase shifter input port and a phase shifter output port 17 which is connected to only one input port of the spatial 18 filter. The antenna also includes a supply means for 19 supplying wave energy signals. The supply means includes a signal generator supplying a power divider 21 having N output signal ports, each output port connected 22 to only one phase shifter input port.
23 For a better understanding of the present 24 invention, together with other and further objects, reference is made to the following description, taken in 26 conjunction with the accompanying drawings, and its 27 scope will be appointed out in the appended claims.

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1 BRIEF _ SCRIPTION OF THE DRAWINGS
2 Figure 1 i9 a conceptual diagram of an antenna 3 system including a three level spatial filter wherein L~ signa]s applied to an antenna input port are provided to 5 the antenna element associated with the port and to the 6 antenna elements adjacent to the associated element.
7 Figure 2 is a plan view of a printed circuit 8 coupling network of the three level spatial filter 9 illustrated in Figure 1.
Figure 3 i9 a conceptual diagram of an antenna 11 system in accordance with the present invention 12 including a three level spatial filter cascaded with a 13 four level spatial filter.
14 Figure 4 is a plan view of a printed circuit coupling network of the cascaded spatial filters 16 illustrated in Figure 3.
17 Figures 5A, 5B and 5C are antenna patterns for 18 antennas according to the invention employing spatial 19 filters having two level, three level and four level coupling, respectively.
21 Figure 6A illustrates a schematic diagram of a 22 coupler and its relative inputs and outputs.
23 Figure 6B is a listing of the formulas which 24 define the coupler values and the termination values.
Figure 6C illustrates a schematic diagram of a 26 series coupler network.

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l Figure 6D is a generalized schematic 2 representation of a five level spatial filter.
3 Figure 7 illustrates a prototype network for an 4 infinite spatial filter antenna to be employed with the invention.
6 Figure 8A is a schematic diagram of an antenna 7 system of two cascaded 8-coupler spatial filters 8 according to the invention.
9 Figure 8B is a table of the optimum excitations for an 8-port spatial filter according to the invention.
11 Figure 8C is a schematic diagram of a unit cell 12 of a modular antenna system of two cascaded 4-coupler 13 spatial filters according to the invention.
14 Figures 9 and 10 illustrate a computed antenna pattern for the zero-thinned spatial filter shown in 16 Pigure 8A.
17 Figure 11 is a graph illustrating the linearity 18 requirements which limits the deviation from the ideal 19 linear relationship of the MLS guidance angle and the actual angle.
21 Figure 12 illustrates the geometry and formulas 22 of a model of a flat horizontal surface used to quantify 23 the effects of sidelobe radiation on the performance of 24 an automatic flight control system.
Figures 13, 14, and 15 summarize the simulation 26 rasults of vertical acceleration, vertical velocity, and g_ . - , , .,. , , ;.

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1 vertical attitude, respect;vely, with regard to the peak 2 MLS guidance error for 10 feet and 20 feet elevation 3 antenna phase center heights when passenger comfort is 4 considered.

DETAILED DE~CRIPTION OF l'HE INVENTION
6 Figure 1 is a schematic diagram illustrating an 7 antenna system in accordance with the present 8 invention. The diagram of Figure 1 includes a plurality 9 of antenna elements 1~8 arranged in a predetermined path which, in this case, is a straight line. Each antenna 11 element is connected to one and only one output port 12 9-16 of spatial filter 17. The spatial fllter is 13 comprised o~ a plurality of modules A through H, one 14 module for each antenna element. Spatial filter 17 includes 8 input ports, 18-25 each connected to the 16 output of one and only one phase shifter 26-33. The 17 array of phase shifters 26-33 form beam steering unit 18 34. The inputs 35-42 of the phase shifters are 19 connected to one and only one output of power divider 43 20 which is fed by signal generator 44. The power divider 21 and signal genera'~or form a supply means for supplying 22 wave energy signals. Although filter 17 has been 23 illustrated as symmetrical, it is contemplated that 24 spatial filters according to the invention may be unsymmetrical.
26 Referring to the signal path of wave energy ~:
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1 signal supplied by signal generator 44, the original 2 signal is provided via line 45 to power divider 1l3 which 3 divides the signal into eight equal components. Fach l~ component is provided via lines l~6-53 to only one input of beam steering unit 3L~. For example, referring to the 6 left-most portion of the antenna system, line 46 7 provides the signal component to input 35 of beam 8 steering unit 34. The component then passes through 9 phase shifter 26 which may shift the phase of the component according to instructions received from 11 control uni~ 54 via control line 55. The output of 12 phase shifter 26 is provided to input port 18 of spatial 13 filter 17. The signal component provided to input port 14 18 is provided to output port 9 which is connected to antenna element 1 and is also provided by a coupling 16 arrangement to element 2 which is adjacent to antenna 17 element 1, 18 Spatial filter 17 couples component signals which 19 are provided to any input to the antenna element associated with the input and to elements adjacent to 21 the associated element. 50uplers 56-62 couple signals 22 which are provided to an associated antenna element to 23 the antenna element which is to the left of the 24 associated antenna element. The component signal provided to an input is transmitted to the antenna 26 element associated with the input by transmission lines 27 64-71. For example, the component signal pro~ided by :
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branch 39 of the power divider 43 is fed through phase 2 shifter 30 and provided to input 22 of spatial filter 3 17. Input 22 is connected by transmission line 68 to 4 its associated output 13 and antenna element 5. The component signal is also coupled by coupl,er 59 to 6 antenna element 4 which is to the left of and adjacent 7 to antenna element 5. Similarly, component signals 8 provided to an input are also coupled to antenna 9 elements adjacent and right of the associated antenna element by couplers 72-80. For example, the co~ponent 11 signa.l provided by branch 49 of the power divider to 12 input 38 of phase shifter 29 passes through phase 13 shifter 29 and is provided to input 21 of the spatial 14 filter 17. The component signal is then provided to output 12 by transmission line 67. Output 12 is 16 directly connected to antenna element 4. Element 5 is 17 adjacent to and to the right of antenna element 4 and 18 receives a portion of the component signal via coupler l 9 76 . Element 3 is adjacent to and to the left of antenna element 4 and receives a portion of the component signal 21 via coupler 58.
22 Spatial filter 17 is shown in modular form. As a 23 result, the input to coupler 72 is terminated by 24 termination 81 because there is no antenna element to the left of antenna element 1. Similarly, the output 26 from coupler 56 is terminated by termination 82 because 27 there is no antenna element to the left of antenna . .. ....

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element 1 to receive the component signal provided to 2 input 18. On the right side of spatial filter 17, 3 coupler 80 is terminated by termination 83 and coupler 4 ~3 is terminated by termination 84 because there is no antenna element to the right of antenna element 8 to 6 receive the couple signal from coupler 80 or to provide 7 a coupled signal via coupler 63.
8 Figure 2 illustrates a plan view of a printed 9 circuit coupling network useful as the spatial filter 17 10 OI Figure 1. Network 17 includes input ports 18-25 11 connected to the outputs of beam steering unit 34.
12 These input ports are connected to a first series of 13 couplers C1 shown in detail in Figure 2A. Coupler C1 as 14 well as all other couplers may be standard microstrip 15 network couplers having a predetermined coupling 16 ratio. The specific coupling ratio depends on the 17 width, length and on the thicknesLs of the transmission 18 lines within the coupler. ~y convention, signals 19 provided to the inputs 101 and 102 of coupler C1 are 20 coupled to the outputs 103 and 104 according to a 21 predetermined ratio. In the case of coupler C1, input 22 102 is terminated by termination 105 resulting in any 23 component signal which is supplied to input 101 being 24 distributed to outputs 103 and 104 such that C 1 2 + T 1 2 = 1 .
26 Following the firs-t array OI couplers C1 is a 27 second array of couplers C2 illustrated in more detail ': ~ ` ' , '';

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1 in Figure 2~. Signals provided to inputs 105 and 106 2 are combined and transmitted to output 108 at a ratio T2 3 and coupled to output 107 at a ratio C2 such that 4 T22 + C22 + 1. Completing the three level spatial filter 17 is a third series of couplers 109-116.
6 According to the invention, these couplers have the same 7 configuration as coupler C1. Couplers 109-116 work in 8 the same manner as coupler C1 as shown in Figure 2A by 9 combining signals provided to their inputs to the outputs 9-16 o~ spatial ~ilter 17.
11 As specified by the invention, spatial filter 17 12 is ideally lossless (except for dissipative losses) and 13 for that reason the relationships 14 Cl + T12 = 1 and C22 + T22 = 1 must apply to the power (voltage) passing through each 16 coupler C1 and T1, respectively. The following 17 relationship ensures the lossless condition for the 18 network:
19 C1 = 2 (1 + ~1 - C2 ~ (1) This relationship can be derived by setting the 21 inputs at 18-25 equal to unity and the inputs to the 22 terminations 117-124 equal to zero.
23 As used in rega~d to the invention, a non-thinned 24 spatial filter is a filter formed by an array of couplers. The array is essentially lossless in that the 26 power dissipated within terminations is minimized.
27 Figure 3 is a schematic diagram of an antenna .

1 system in accordance with the invention including a 2 three/four level cascaded spatial filter 300. In 3 general, this spatial filter may be used in combination 4 with the antenna system as shown in Figure 1 by replacing spatial filter 17 with spatial filter 300.
6 Each antenna element 1-8 would then be connected to one 7 and only one output port 301 of the spatial filter 8 300. Spatial filter 300 is comprised of a plurality of 9 modules A through H, one module for each antenna element. Spatial filter 300 includes input ports 302 11 each connected to one and only one of the outputs of a 12 phase shift network.
13 Figure ll is a plan view of a printed circuit 14 coupling network of the cascaded spatial filter 300 illustrated in Figure 3. Network 300 includes input 16 ports 302 connected to the output ports of a beam 17 steering unit. These input ports are connected to a 18 first series of couplers C1 shown in detail in Figure 19 2A. Following the first array of coupler C1 is a second array of couplers C2 illustrated in more detail in 21 Figure 2B. Following the second array of couplers C2 is 22 a third array of coupler C2. Completing the four level 23 spatial filter 300 is a fourth series of couplers C1.
24 According to the invention, for symmetrical excitations, couplers C1 at the beginning and end of the array and 26 intermediate couplers C2 have the same configuration.

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l The following relationship ensures the lossless 2 condition for the networks 3 C12 = - + C2 ~1 - C2 (2) 4 Figure 5A illustrates an ideal antenna pattern for an antenna according to the invention,employing 6 spatial filters having a two level coupling.
7 Essentially this coupling creates lobes 501, 502 and 8 503. Figure 5B illustrates a typical antenna pattern employing a three level spatial filter which for~s a single lobe 504. Figure 5C illustrates a typical 11 antenna pattern for a four level spatial filter 12 generating a more well defined single lobe 505.

13 Synthesis Procedure For Five 14 Level Non-Thinned Spatial Filter Step 1: Referring to Figures 6A, 6B, 6C, and 6D, 16 determine initial values for couplers C1--C5 17 (a) specify desired excitations A1-A5 18 (b) specify C1 19 (c) compute C2-C5 Using Figure 6C
Step 2: Compute actual excitations A1'-A5' according 21 to the following formulas:
22 (a) A1' = T5C4C3C2C1 23 (b) A2' = C5T4C3C2C1 - T5T4T3C2C1 :
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1 (c) A3' = C5C4T3C2Cl - T5C4T3C2T1 2 - T5T4T3T2Tl - T5T4C3T2C1 3 - C5T4C3T2Tl - C5T4T3T2Cl 4 (d) A4' = C5C4C3T2C1 - T5T4C3C2T1 - C5T4T3C2T1 - C5C4T3T2Tl 6 (e) A5i = C5C4C3C2T1 7 Step 3: Adjust values for couplers C2-C5 8 (a) adjust C5 such that A2' A2 A1' A1 (b) adjust C4 such that 11 A3' A3 A2' A2 12 (c) adjust C3 such that 13 A4' A4 A3' A3 14 (d) adjust C2 such that A5' _ A5 ~, _ ~
16 Step 4: Recompute actual excitations A1' - A5' 17 (see Step 2 for formulas for A1' - A5') 18 Step 5: Normalize actual excitations by computing 19 A1" - A5"
(a) Let A1" = 1 . Then, '' ":
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(b) A2" = A1, 2 (c) A3n = A3 3 (d) Al~n = A4 4 (e) A5" = A1 5 Step 6: Compute deviation S between normalized 6 actual excitations A1" - A5" and desired 7 excitations A1 - A5 8 S ~ (AN" - AN) N = 1,2,.......... ,5 9 Step 7: Repeat steps 3-6 until deviation S is within an acceptable limit 11 Step 8: Repeat steps 1--7 until ratio of power in 12 terminations PT to radiated power PR is a 13 minimum i.e., minimize PT
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PT ~(TN) ; PR = ~[AN)2 N = 1,2,...,5 :: ~
15~ For example, consider the case o~ a five 16 element aperture as illustrated in Figure 6A. Assuming : :
17 the desired excitation (from step 1a) is:
18 A1 = 1.0000 .~., A2 = 1.6086 2 ~3 = I.93156 3 A4 = 1.6086 4 A5 = 1.0000 Let C1 = 0.979 (from step lb); then, the 6 values of the other couplers (from step 1c) are:
7 C2 = 0.9502 8 C3 = 0.9366 9 C4 = 0.9600 C5 = 0.9852 11 The normalized actual excitations (steps 12 2-5) result in:
13 A1 = l 14 A2 = 1.3755 A3 = 1.6478 16 A4 = 1.5449 17 A5 = 1.1957 18 The db loss (from step 8) between the 19 normalized actual excitations (from step 5) and the ::
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21 LOSS = 7.12db 22 Table 1 below continues the synthesis 23 procedure.

~;~ T l C1 C2 C3 C4 C5 A11' A2'' A3'' A4'' A5'' loss .979 .9225 .8401 .9042 .979 1 1.6061 1.932 1.6061 1 6.72db 2 .98 .9285 .857 .9132 .98 1 1.608 1.932 1.611 1 6.59db 24 3 .985 .953 .9155 .9461 .985 1 1.608 1.933 1.604 1 6.69db 4 .99 .971 .9523 .9685 .99 1 1.608 1.931 1.609 1 7.68db 5 .981 .9343 .8718 .9212 .98 1 1.6085 1.932 1.6085 1 6.53db , ~ .

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~ 3 l Table l 2Five Coupler Synthesis 3As shown in table l, trial 5 illustrates an 4 optimum arrangement with minimum power loss. As shown in table 2, trial Ll illustrates an optimujm arrangement 6 for a five coupler structure where the symmetry of the 7 excitation is invoked to set C5 = Cl and C4 = C2.

Trial C1 C2 C3 A1 A2 A3 loss l .981 .91506 .85575 1 1.6086 1.932 6.93db 8 2 .979 .8823 .7849 1 1.6086 1.9318 7.90db 3 .982 .92425 .8739 l 1.6086 1.932 6.80db 4 .984 .93866 .90095 l 1.6086 1.9321 6.75db .986 .95011 .92131 1 1.6086 1.932 6.90db 9 Table 2 Five Coupler Synthesis, C5 = Cl, C4 = C2 11 Although the above procedure has been applied to 12 develop a symmetrical filter, the procedure is general 13 in nature and can also be used to develop non-14 symmetrical filters. Symmetry is generally preferred to maintain simplicity and reduce complexity. Symmetrical 16 filters usually employ redundant couplers and other 17 structures which minimizes design efforts.
18 The design of a spatial filter involves the 19 determination of coupler values ~or a multilayer circuit. No closed form solution is readily apparent to 21 the synthesis of a network that produces a specified 22 output voltage distribution. However, analysis of any :
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a~'~9~3 1 network is possible. There~ore, synthesis involves the 2 iterative trial and error procedure described above in 3 which coupler values are gradua1ly adjusted until the 4 desired outputs are achieved.
Since the analysis of a complex network requires 6 significant computer time, it is desirable to formulate 7 an iterative algorithm that converges to the desired 8 solution within a reasonable time. Analysis of every 9 possible combination of coupler values could take weeks or months to e~aluate on the computer. Furthermore, an 11 infinite number of solutions exist that produce the 12 desired amplitude distribution. The difference in 13 solutions is the insertion loss of the resulting 14 network. Therefore, it is necessary to determine by theoretica] means the minimum possible 16 loss, so that it will be known when an optimum solution 17 has been achieved.
18 The theoretical loss of a spatial filter network 19 is determined by conservation of power considerations.
The network prototype is shown in Figure 7. The network 21 is symmetrical and continues to infinity in both 22 directions. Each input excites a sub array with N
23 outputs. The sub array outputs, resulting from adjacent 24 inputs, overlap. The network shown in Figure 7 has an equal number of inputs and outputs. Therefore, the 26 input and output spacings are equal and, when all inputs 27 are excited, each output port will be the sum of 1 contributions from N input ports. There must be an 2 internal termination for each output port.
3 The output excitation that results from input 1 4 is designated A1(N), whereas the output excitation resulting from input 0 is designated AO(N?. Because the 6 networ~ is symmetrical, A1(N) = AO(N) = Aj(N).
7 Similarly, the power terminated, designated as Bj(N), 8 must also be equal.
9 The network is realized with N layers of directional couplers. To achieve the desired symmetry, 11 all coupler values in a given layer must be equal.
12 Furthermore, a symmetrical output excitation (Aj(1) =
13 Aj(N), Aj(2) = Aj(N-1), etc.), requires that the coupler 14 values in the first layer be equal to those in the Nth layer, etc. Therefore, as an example, an 8-output 16 network has 8 layers of couplers. If the 8-element 17 excitation is symmetrical, C1 (coupling value for all 18 couplers in first layer) must equal C8, C2 = C'7, C3 =
19 C6, and C4 = C5. Therefore, there are only 4 different coupler values or unknowns that must be determined for 21 an 8-output network.
22 When input power is delivered to port one, 23 conservation of power dictates the sum of powers in 2ll A1(N) added to that internally terminated (B1(N)) must equal the input power. A normalization to an input 26 power of 1 watt yields the equation:

~ A1(N) + ~ B1(N) = 1 1 i=1 i=1 2 The A~s and B's are voltage coefficients. The 3 power at each output port is equal to the square of the 4 voltage coefficierlt when the system imped,ance is normalized to one ohm.
6 ~hen all input ports are excited with equal power 7 and in phase, the output at each port is the sum of N
8 voltages. From symmetry and conservation of power, the 9 sum of the power at one output port and its internal termination must equal one watt. All output ports will 11 be equal.

12 ~ A1(N) + ~ B1(N) = 1 (4) i=1 i=l 13 A co~bination of equations (3) and (4) gives:

14 ~ A1(N) + B1(N) = A1(N) ~ ~ B1(N) (5) i=l i=1 i=1 i=l If the network is to be lossless when a single 16 input port is excited, no power can be delivered to the 17 internal terminations (all B~s = O). If that condition 18 exists, 19 A1(N) = ~ A1(N) (6) ~ ~ i=l i=l :~;
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1 There are few output excitations that satis~y 2 equation 6. The least loss occurs for an excitation 3 that does not satisfy equation 6 when 1~ ~ Bl(N) = 0 or ~ B1,(N) = 0 (7) i=1 i=1 When that condition is met, the network will be 6 lossless when all input ports are excited with equal 7 amplitude and phase. The 105s, when a single input port 8 is excited and the sub array pattern has a maximum in 9 the in-phase direction, is given by:

loss = ~ A1(N) ~ ~1(N) (8) i=1 i=1 11 ~hen the sub array pattern has a maximum in a 12 direction other than the in-phase direction, the lower 13 bound on the loss is increased by the difference in the 14 sub array gain in the two directions. The optimum networ~ is one that provides the least loss. The loss 1:6 that can be expected is the difference between the 17 computed network loss and the theoretical value. Thus, 18 if one computes the theoretical minimum loss to be 3.1 19 dB when a single input port is excited using equation 8, and the least loss that can actually be achieved with a 21 realizable network is 4.6 dB, it will be found that the 22 loss, when all inputs are excited in phase, is 1.5dB.
23 This 1.~ dB loss results from the consideration of the .
' 1 center of the sub array pattern. When the array is 2 scanned to the sub array peak the theoretical loss is 3 reduced to zero.
Il The basic spatial ~ilter network topologies are well-known. A preferred implementation requires 17 6 layers and is nearly impossible to synthesize. A
7 practical network, that closely approximates the 8 per~ormance of a l7-layer network, uses two cascaded 9 8-layer networks as illustrated in Figure 8. The pattern characteristics for this network are shown in 11 Figures 9 and 10 for a radiating element spacing of 0.79 12 wavelengths.
13 Figure 11 describes the linearity requirement for 14 MLS glide path guidance. The discussion of linearity concentrates on the ele~ation guidance per~ormance, 16 however, linearity is also a requirement for the azimuth 17 guidance. Linearity is a subject that has generated 18 much discussion in the MLS community. The invention 19 provides a phased array antenna which meets the elevation linearity requirement. The spatial ~ilter 21 network is a practical way to satis~y the low effective 22 sidelobe requirement which is directly related to the 23 linearity requirement.
24 The linearity ~autopilot) requirement limits the deviation ~rom the ideal linear relationship of the MLS
26 guidance angle and the actual angle (see Figure 11). It ~: :

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1 specifies the transverse accuracy characteristic of the 2 angle guidance signal as opposed to the longitudinal 3 characteristics of PFN and CMN. The longitudinal 4 characteristic causes the aircraft to deviate from the glide path (bends~ or generates noise-lik,e action of the 6 controls. The transverse characteristic is capable of 7 causing instability in an automatic flight control 8 system.
9 After several years of discussion within the MLS
community it is now generally accepted that PFN, CMN and 11 linearity for the EL guidance equipment are all 12 dependent on the effective sidelobe level of the 13 antenna. The issue has been which one of the three 14 characteristics (PFN, CMN or linearity) is the driver with respect to the specification of the effective 16 sidelobe level. The Path Following Noise (PFN) relates 17 to the path following mean course error and is caused by 18 any frequency component that an aircraft can follow.
19 The Control Motion Noise exists in situations where there is no PFN but the scanned MLS signal indicates a 21 bounce or deviation which an aircraft cannot follow.
22 initially it was argued that PFN was the driver. The 23 effective sidelobe level required to ensure that the PFN
24 for a 1.5 beamwidth antenna does no~ exceed 0.083 is -25 dB (a 0 dB ground reflection coefficient is assumed, 26 the 0.0830 PFN limit is derived t'rom the ICA0 standard 27 that the PFN shall not be greater than plus or minus l.3 1 feet). After some analysis by the FAA, it was 2 recogni2ed that with the antenna phase center 20 feet.
3 above the reflecting ground, CMN could be generated when 4 the aircraft was within 2000 feet. of the runway threshold. Consequently, in the draft sp,ecifications 6 for the FAA second MLS procurement, the effective 7 sidelobe level is specified such that the CMN does not 8 exceed 0.0ll5. This requires an effective sidelobe 9 level of -30 dB for a 1.5 beamwidth antenna.
Based on the results of simulations of an actual 11 automatic flight control system in service it has been 12 concluded that linearity is the most stringent 13 requirement with respect to the specification of the 14 effective sidelobe level. The results of the simulations indicate that the angle error limit must not 16 exceed 0.02~ to ensure performance of an automatic 17 flight control system within passenger comfort levels.
18 This error limit corresponds to a -36 dB effective 19 sidelobe level for a 1.5 beamwidth antenna.
The discussion on the linearity requirement has 21 raised the issue of the measurement methodology for 22 determining compliance with specifications. With regard 23 to this issue it should be recognized that effective 24 sidelobes can be measured on an antenna range and that design approval by an authority can be based on these 26 antenna range measurements.

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1 The ~idelobes radiated by the elevation antenna 2 in the direction of the ground are folded back on the 3 main beam because of specular reflection. The sidelobe ll radiation distorts the beam and causes PFN, CMN and linearity errors. The specification of P,FN and CMN
6 limits the magnitude of the angle guidance error. The 7 linearity error, however, depends on the product of the 8 maximum angle guidance error and the height of the 9 antenna phase center above the reflecting ground surface. A large error-height product is capable of 11 causing substantial degradation of the guidance loop 12 gain of an automatic flight control system to the point 13 where the automatic flight control system becomes 14 unstable. For example, a maximum error of 0.045 and a phase center height of 20 feet can cause the loop gain 16 to vary between +6 dB and less than -40 dB (at the "max 17 gain spot" and the "dead spot", see Figure 11).
1~ The model of a flat horizontal surface is used to 19 quantify the effects of sidelobe radiation on the performance of an automatic flight control system. The 21 geometry and formulas are presented in Figure 12. For 22 the case of a constant glide path, the magnitude of the 23 error remains essentially constant and the phase 24 variation is that attributed to the path difference between the direct signal and the indirect signal 26 emanating from the ground image of the EL antenna.

~ ' ' , ' ' ' ' ' ' '' 1 The model was used as a perturbation input to a 2 simulation of an automatic glide slope control systeM
3 for a small jet aircraft. The criteria for the ll acceptability of the automatic flight control system is passenger comfort. Figures 13, 14 and 15~provide a 6 summary of the simulation results with respect to the 7 allowable peak MLS guidance error, elevation antenna 8 phase center height and passenger comfort. The 9 simulations start at a distance of 3 r~M from the elevation antenna.
11 Figure 13 shows that for a 20 feet phase center 12 height and a peak error of 0.0830 the automatic control 13 system is unstable. The vertical accelerations exceed 14 the passenger comfort level by a factor of 2. 4 :1 . For peak error of 0.045, the system is marginally stable;
16 for larger phase center heights, say 37 feet, it is 17 expected that the system would be unstable (the error-18 height product, 0.045 X 37', is equal to that of the l9 0.083~ maximum error and 20 feet height case). Figures l 4 and 15 exhibit the same trends; they show that for a 21 20 feet phase center height and a peak error of 0.0830 22 the vertical velocity and attitude exceed the passenger 23 comfort levels by factors of 4:1 and 2:1 respectively.
24 The following conclusions are based on a study of the available information, with respect to the 26 specification of the effective sidelobe level and 27 autopilot performance within passenger comfort levels:

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, 1 1. the present PFN error limit (0.083) is not 2 acceptable;
3 2. the present CMN error limit (0.045) is 4 marginal (especially if higher than 20 feet antenna phase center heights ,are 6 contemplated);
7 3. a limit of 0.0240 appears to be acceptable 8 ~or tlne case studied;
9 4. linearity is the dominant system requirement with respect to the speci~icaticn of the 11 effective sidelobe level; and 12 5. the error-height product should not exceed 13 0.45 degrees-feet.

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Claims (19)

1. Claim 1. An antenna system for radiating wave energy signals into a selected angular region of space and in a desired radiation pattern comprising:
   a lossless spatial filter having N inport ports and N output ports;
   an aperture comprising a plurality of N
   antenna elements arranged along a predetermined path, each element connected to only one output port of the spatial filter, a beam steering unit comprising N phase shifters and means for controlling the phase shifters, each phase shifter having a phase shifter input port and a phase shifter output port which is connected to only one input port of the filter; and supply means for supplying wave energy signals, said supply means including a signal generator supplying to a power divider having N
   signal output ports, each output port connected to only one phase shifter input port;
   whereby when wave energy signals are supplied by the signal generator through the power divider, signal supplied by a signal output port of the power divider are coupled to the antenna element associated with said output port and to adjacent antenna elements, to cause said aperture to radiate said desired radiation pattern primarily within said selected region of space without grating lobes.
2. Claim 2. The system of claim 1 wherein said spatial filter comprises:
   a plurality of N first coupling means each having a first input port, a first coupled output port and a first transmitted output port, said first coupling means for distributing without loss wave energy signals applied to the first input port, such applied signals being distributed to the first coupled output port and to the first transmitted output port according to a first predetermined ratio, said N first input ports being the N input ports of the spatial filter;
   a plurality of N second coupling means interspersed between said N first coupling means, each having a second left input port associated with the first coupled output port of the right adjacent first coupling means and a second right input port associated with the first transmitted output port of the left adjacent first coupling means, said second means having a second coupled output port and a second transmitted output port, said second coupling means for combining and distributing without loss wave energy signals applied to the second left and second right input ports, such applied signals being distributed to the second coupled output port and the second transmitted output port according to a second predetermined ratio; and a plurality of N third coupling means interspersed between said N second coupling means, each having a third left input port associated with the second coupled output port of the right adjacent second coupling means and a third right input port associated with the second transmitted output port of the left adjacent second coupling means, said third coupling means having a third output port, said third coupling means for combining without loss wave energy signals supplied to the third left input port and to the third right input port, such applied signals being combined and provided by the third combining output port according to a third predetermined ratio, said N third output ports being the N output ports of the spatial filter.
3. Claim 3. The system of claim 2 wherein said first predetermined ratio equals said third predeter-mined ratio.
4. Claim 4. The system of claim 3 wherein said second predetermined ratio (C2) is associated to said first predetermined ratio (C1) according to the following:
   
   
5. Claim 5. The system of claim 2, said spatial filter further comprising a plurality of N
   fourth coupling means located between said second means and said third means, each of said N said fourth means interspersed between said N second coupling means, each having a fourth left input port associated with the second coupled output port of the right adjacent first coupling means and a fourth right input port associated with the second transmission output port of the left adjacent first coupling means, said fourth means having a fourth coupled output port associated with the third right input port and having a fourth transmitted output port associated with the third left input port, said fourth coupling means for combining and distributing without loss wave energy signals applied to the fourth left and fourth right input ports, such applied signals being distributed to the fourth coupled output port and the fourth transmitted output port according to a fourth predetermined ratio.
6. Claim 6. The system of claim 5 wherein said first predetermined ratio equals said third predetermined ratio and said second predetermined ratio equals said fourth predetermined ratio.
7. Claim 7. The system of claim 6 wherein said second predetermined ratio (C2) is associated to said first predetermined ratio (C1) according to the following:
   
   
8. Claim 8. The system of claim 1 wherein said spatial filter comprises:
   distribution means having N distribution input ports and 2N distribution output ports for distributing without loss wave energy signals applied to said distribution input ports, to such applied signals being distributed to the distribution output ports according to a first predetermined ratio, said N distribution input ports being the N input ports of the spatial filter;
   first transmission means having 2N first transmission input ports, each associated with only one of the 2N distribution output ports, and having 2N first transmission output ports, said first transmission means for combining and distributing without loss wave energy signals applied to said first transmission input ports, such applied signals being combined and distributed to the first transmission ports according to a second predetermined ratio;
   combining means having 2N combining input ports, each associated with only one of the 2N
   first transmission output ports, and having N
   combining output ports, said combining means for combining without loss wave energy signals applied to said 2N combining input ports, such applied signals being combined at the combining output ports according to a third predetermined ratio, said N combining output ports being the N
   input ports of the spatial filter.
9. Claim 9. The system of claim 8 wherein said first predetermined ratio equals said third predetermined ratio.
10. Claim 10. The system of claim 9 wherein said second predetermined ratio (C2) is associated to said first predetermined ratio (C2) according to the following:
    
    
11. Claim 11. The system of claim 8, said spatial filter further comprising a second transmission means located between said first transmission means and said combining means, said second transmission means having 2N second transmission input ports, each associated with only one of the 2N first transmission output ports, and having 2N second transmission output ports, each associated with only one of the 2N combining input ports, said second transmission means for combining and distributing without loss wave energy signals applied to said second transmission input ports, such applied signals being combined and distributed to the second transmission output ports according to a fourth predetermined ratio.
12. Claim 12. The system of claim 11 wherein said first predetermined ratio equals said third predetermined ratio and said second predetermined ratio equals said fourth predetermined ratio.
13. Claim 13. The system of claim 12 wherein said second predetermined ratio (C2 is associated to said first predetermined ratio (C1) according to the following:
    
    
14. Claim 14. The system of claim 1 wherein said filter comprises first and second cascaded lossless spatial filters having N input ports and N output ports.
15. Claim 15. The system of claim 2, 3 or 4 wherein said spatial filter comprises a printed circuit located on a single substrate.
16. Claim 16. The system of claim S, 6 or 7 wherein said spatial filter comprises a printed circuit located on a single substrate.
17. Claim 17. The system of claim 8, 9 or 10 wherein said spatial filter comprises a printed circuit located on a single substrate.
18. Claim 18. The system of claim 11, 12 or 13 wherein said spatial filter comprises a printed circuit located on a single substrate.
19. Claim 19. The system of claim 14 wherein said spatial filter comprises a printed circuit located on a single substrate.