CA1229896A - Detection logic and signal processing method and apparatus for theft detection systems - Google Patents

Abstract

DETECTION LOGIC AND SIGNAL PROCESSING
METHOD AND APPARATUS FOR THEFT DETECTION SYSTEMS
ABSTRACT
Improvements in theft-detection or surveillance systems of the type in which an alternating electromagnetic field is established across a doorway or other portal and is monitored to detect the presence within the field of a marker or tag member comprising a small strip of permalloy or like material of high permeability hidden or otherwise carried on merchandise or other articles and objects to thereby "mark"
such merchandise or objects, i.e., to make them readily detectable even though hidden from view. The improvements reside in signal-processing electronic circuity which increases detection sensitivity and accuracy while at the same time reducing erroneous detection results. In par-ticular, the circuitry utilizes summing and differencing techniques to improve signal-to-noise ratios and eliminate previously-unsuspected sources of error, and additionally utilizes the concept of frequency spectrum-content ratios as a determinant in distinguishing between apparent detection of true markers from other objects or structures whose response to the alternating interrogation field closely resembles that of the true markers and would normally produce erroneous detection indications. In doing so, detection signals are processed by use of sampling techniques representative of both marker-presence and marker-absence, comparison of these samples through summing, differencing and peak-integrating techniques, as well as other more particular approaches for reducing or eliminating the effects of error and/or noise signals resulting from or introduced by causes not previously understood or appreciated.

Description

2~ 196 DETECTION LOGIC AND SICNAL PROCESSING
~ETHOD AND APP~RATUS FOR THEFT DETECTION SYSIE~
TECHNICAL FIELD
This invention relates in a broad sense to detection systems and apparatus, and more particularly to de~ection systems principally used in "anti-pilfering", i.e., theft-prevention, systems; more particularly still, the invention relates to that type of detection system in which an alternating electromagnetic field is monitored to unobtrusively and invisibly detect the presence ~ithin the field of a small strip of permalloy or like highly-magnetizable (ultra-ls~Y
coercivity) metal foil which is hidden upon or in articles such as consumer merchandise whose theft or othenYise-impermissible ta~in~
is to be detected and prevented.
BACKGROUND OF THE INVENTION
Many prior efforts have been made toward deterring or pre-venting ~hefts in the nature of shoplifting~ or other undesired removalof "contraband" articles or goods, or example, unchecked librar~
books or the like, and such prior efforts have given rise to a variety of different systems and approaches, based upon dierent technological phenomena including, for e~amplel detection o~ permanent magne~ pieces, a variety o electromagnetic field applications, micrswave systems, infrared or ultraviolet, etc. These rather extensive prior efforts have, quite understandably, advanced the general state of the art in these diferent fields, and have in general enhanced the degree of success available; however, ~he desired end is exceedingly difficult from a technological point of view, since the areas ~o be moni~ored (in general, doon~ays or like points o~ egress~ are large in a physical sense, whereas the articles ~mder surveillance are usually relatively small, requiring a proportionally ~iny detection element or "marker".
Generally speaking, this requires e~ceedingly high system sensi-tivity, but it is not only important to d~tect the illicit passageo contraband material; it is almost equal~y as important to a~oid "~alse alarms", in which bona fide customers or other innocen~ persons are wrongly pointed out as carrying stolen or contraband goods through the portal, since this not only leads to i~m~diate wTongful embar-rassment of the individual involved, but also is likely to cost themerchant or other proprietor the loss of sllbstc~ntial goochYill and, -2- ~L~2~3~

potentially, possible litigation by those claiming to be damaged by such incidents.
Accordingly, real progress satisfying both of the afore-mentioned requirements of high-sensitivity attended by great selectivity has been di-fficult to achieve and slol~ in coming. This conclusion is evidenced by the issuance of various patents over a long period of years, each asserting the achievement of improvement~, but each followed in time by another patent directed to still a further improvement in a seemingly continuous sequence. By way of example, perhaps the most frequently-employed, and probably the most success-full system concept, relates bacX to the often-noted French Patent of P. A. Picard, No. 763,681, issued in 1934, in which the technological phenomenom is described as involving electromagnetic field perturba-tions resulting from the insertion or presence within the field of a piece of magnetic material. In particular, Picard noted the field effects created by the presence of highly-magnetic (high penneability) material such as pe~nalloy, which creates the presence of a number of the higher-order odd hannonics of the fundamental frequency of the applied field (e.g., Picard referred to the presence of the ninth and eleventh harmonic). While a period of almost 50 years has elapsed since the appearance of this pa~ent to Picard, various patents con-tinue to issue from time to t~ne asserting advances in Picard's theories and findings in the area of '~ilferage detection" systems of the type noted hereinabove; foT example, reference is made to a number of paten~s issued to Edward Fearon (including U. S. Nos. 3,631,442, 3,754,226,

3,790,945, 3,820,103, 3,820,104) and to Peterson (U. S. No. 3,747,086), Elder et al. (U. S. Nos. 3,665,44~ ~d 3,765,007~ as well as U. S.
Patent No. 3,983,S52 to Bakeman. Indeed, a very recent such patent is that issued to Robert Richardson, U. S. No. 4,300,183, which is directed to and describes various attributes of the underlying-concept relating back to Picard.
As stated above, the seemingly continuous advance in the general state of the ar$, as evîdenced by the aforementioned patents, has undoubtedly provided new insights ~nd improvements in the general level o the art, the requirements of truly satisfactory detection systems are very severe and demanding, and the need therefore continues to exist, and in some ways becomes even more pronounced, for truly reliable systems ~hich will unerringly detect relatively small '~arker"
elements or indicia, while at the same time being essentially immune to a practically endless number of widely-varying metal devices, objects, S articles, and components, all of which cause perturbations in the magnetic interrogation ield, with resulting detection-actua~ing results being inevitably present.
BRIEF SU~MARY OF THE INVENrION
The present invention provides new and highly significant improvements in electromagnetic field-type detection systems of the type noted above, whirh improvements substantially enhance both the sensi-tivity and the selectivity of such a system, pursuant to which pre-viously-unappreciated detrimental effects such as ield-perturbing metal structural components in the environment of the egress ~ortal (e.g., field-perturbing eeiling grids overhead and/or field-perturbing rein-forcing rods or mesh in structural concrete nearby, etc.) are substan-tially eliminated as error sources. The improved system provided hereby thus makes it possible to accurately, consistently, and reliably detect ~he presence of tiny markers or ~ags of magnetic mateTial and reject, or not detect, the presence of other field disrupting metal elements or components as, for example, keys, pocXetXnives, wristwatches~
metal containers such as beYerage cans or the like~ baby strollers and shopping carts, and a host of other widely-differing apparatus and objects In accordance with the disclosure a detection system and method is provided with greatly enhanced processing of the marker-detection signals, incorporating a summing and diferencing procedure for substantially impro w d signal-to-noise ratios, in accordance with which a co~paratively low frequency component band and a comparatively high requency component band are separately det0rmuned, and utilized in a multiple-step comparative manner to dynamically control the detection alarm threshold. In this manner, a balancing of the frequency components or bands is utilized, to produce alarms only when the ratio of requency bands is in the appropriate order, representatiYe of the actual marker indicia, thereby avoiding false alar~s produced by prior systems in response to metal articles whos~ field-perturbation .. . ~

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effects happen to be very similar to those o the authen~ic marker, even including those articles which produce similar frequency components but ~Yhich are distinguishable by the relative amounts of diferent frequency bandsa i.e., the ratio of the signal strength representa-tive of different frequency component bands.
Further in accordance with the disclosurethe method and apparatus provided operates to additively, or construetively, sum representations of detection signals indicative of marker presence within the field, regardless of and continuously consistent with, field alternation phase changes and differences; additionally, the method and apparatus provided dif~erences or subtracts signals representative of non-marker presence (i.e., nois~) in order to accurately portray non~
marker effects. In this manner, the marker-charac~erizing signals are comparatively analyzed by reference to the non-marker signals, thus substantially enhancing selectivity.
Somewhat more particularly, in accordance with the prese~t disclosure detection logic and processing methods and apparatus ar~
provided for examining representations of electrical signals which ar~
produced initially by receiver means that monitor an alternating elec-tromagnetic interrogation field in which a predetermined marXer may bepresent, so as to accurately and reliably determine the presence, and/or the non-presence, o such 2 marXer in such a ~ield, In accord-ance herewith, such signal-examining is carried out in a manner having the efect of determining, and using, a first and second composite analysis signal, the first being used to set a first value of a com-parison operation and the second such signal being used to set a second such comparison value, such two cornparison values being ef~ectively compared in such comparison operation. Preferablyl this is accom-plished by developing an integrated ambient-representative signal and using the latter to vary a preset threshold in comparison stages whose primary comparison inpu~s are signals representative of marker presence and marker absence, with the results of such comp~risons being summed against each other, and the resultant su~ation level used to trigger an alarm or lndicator showin~ the verified presence of the marker within the interrogation field. In particular, the inventor provides pre~erred methods and appara~us for selective pre~mplification and processing of ,~ .

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the signals produced initially by the receiver means9 prior to the opera~ion of the preferred detection logic and processing methods and apparatus, both of which are further disclosed in the following de-scrip~ion of particular preferred embodiments.

In accordance with one aspect of the invention there is provided, a me~hod for determining the presence of a predeternlined marker member within an alternatlng electromagnetic int~rrogation field, o~ the type using one or more monitors ~hich produce ~ecerical signals o containing indicia indicative of the pres~nce o'~ such a marker withinsuch field, the improveme~t comprisin~: subjecting the slgnals frQm different ones of said monitor means to preamplification and subtracti~e processing before the signal anal~sis which determines and in~icates marker presence or absencej said preamplifica~ion compr~sing the separate application of the signals ~r,om different monitor means to separate preamplification channels, and said subtractive processing comprising subtracting a monieor mea,ns signal produced by a monl~or means relatively further ~rom a mar~er within ~he interrogation field from a different marker means signal produc~d by ~ moni~or means IYhich is relatively closer to that marker, thereby at least partially cancelling out ambient noise and/oF other undesired effects from the monitor mean~ signals prior to urthe~ processing thereaf.

In accordance w1th a second aspece of the ln~ention there is a method for det~cting the presence of a partlcular marker me~ber within In a mothod of det~ctlng thc pres~c~ o~ a partlcul~r m~r~or ~e~bc~ wlthln ~n a1tern~tln~ electroma~netlc lnt~ro~tlon fl~ld ~st~b1lah~d ~croas a portal, whercln sald ol~ctrom38notlc Pl~ld 13 monltor~d by at lca3t flr~t and second recelver mean3, eAch for accessln~ the ~leld ~o~ a dlf~3s~nt poaltlon and ~or producln~ an alectrlcal u~gna1 roprasontatl~a of ~lgna1 ,indlcla cau~qd ~y such a markcr ln ~aspons~ to thc alto~natlons o~ the flelt, th~ l~prove~3nt for use in detectln2 s~d mar~or whSch co~prl~s~ the ~teps o~: ~roducin~ a first proccssin~ si~nal for marker-pro30nce ans1ys~3 by uso Oe ~sld alactrlcsl si~nals produced by at 1cast ona o~ ~ald recelver~; producln~ ~ socond procossln~ slBnal eOr U9~ ln ~arkor-pr0senca ~n~1ys~ by dle~ar3ncln~ th~
c10ctricQ1 slgna1s producod by ~ald flrst ~nd sacond recel~ars, to roduce common-mode nols~ or othor unda~l~ed sl~na1 char~ctsrl~tlc~ ln that o1ectrlc~1 si~nal; Bnt usln~ tho second procaq~in~ sl~nal In co~pcratlvo an~1ysl3 wlth ~aid flrst procassln8 al~ctrlc~ ns1 to doteraln4 pr~3ance and ~b~enc3 o~ 8 msrk~r ~lthln said ~i31d.

- 5a A number of additional improvements and adYan~ages are pro-vided in accordance herewith, as described in more detail hereinafter in conjunction with certain preferred embodiments of the invention as depicted in the attached drawings and specifically noted in conjunction therewith for a more meaningful disclosure.
BRIEF DESCKIPTION OF ~HE ~RAl~INGS
-Fig. 1 is a simplified, schematic-fonm block diagram of the overall detection system in accordance with the invention;
Fig. 2 is an enlarged and schema~ic circuit diagram of the preamplifier portion of the system shown in Fig. l;
Fig. 3 is an enlarged schematic circuit diagram of the inhibit amplifier portion of the system shown in Fig. l;
Fig. 4 is an enlarged, simplified system block diagram il-lustrating the preferred detection logic and processing circuitry;
Fig. 5 is a schematic circuit diagram showing a first portion of the detection logic and processing circuitry of Fig. 4; and Fig. 6 is a schematic circuit diagram showing the second portion of the detection logic and processing circuitry of Fig. 4.
DETAILED DESCRIPTION OF PREFERRED Eh~ODIMENTS
. . . _ . .
The general nature o~ the overall system is illustrated in Fig. 1, in which a typical two-portal system is depicted. Generally speaking, such "portals" should be understood as being egress passages, e.g., doo~aysl on the opposite sides of each of which are maintained electromagnetic interrogation field sources (e.g , induction coils constituting part of an oscillating L-C tank cirouit) together with a receiving antenna which monitors the electromagnetic field from that particular side of the portal. As will be understood, many of the prior patents referred to hereinabove depict and discuss systems ~ing such interrogation coils and antennae; for example, Richardson ~U. S. No.

4,300,183) depicts a system and components whose general nature may be taken as more-or less standard, dating back to the work of E. Fearon whose prior patents are also noted above~ As illustrated in the ~ 3''3~;

aforementioned ~ichardson patent, wherein the "portals" are designated "doorways", and whèrein one of a number of different possible coil shapes and orieneations are illustrated, the field-inducing coils are physically large~ such that the interrogation field which they produce occupies a physical area which is more than sufficient for a human being to readily pass through. Inasmuch as the general characteristics, attributes, and parameters of such systems, including their field-inducing coils and receiving antennae, have long been known and have, in effect, resulted from the work of a number of individuals working at various points in time, the afore~entioned prior patents of Fearon, Elder, Richardson, et al. should be consldered as portraying the known state of the art and describing both general system characteriatics and circuitry, componentry, and the liXe; consequently, these patents should, to the extent dee~ed necessary or desirable for environmental disclosure or otherwise, be considered.
Referring further to Fig. 1 herein for a very general illustration of the overall syste~, it will be noted that "portal 1" has a pair of oppositely-spaced sides, designated "side 1" and "side 2", and the same is true with resp~ct to ~ortal 2, which may be considered a substantial duplicaee of portal 1. E~ch side 1 of each portal preferahly receives the same type of drive, i ~., interrogation coil-excitation or drive current and each side 2 coil i5 also driven like its counterparts, although as explained hereinafter the side 2 excitation preferably changes in phase periodically whereas the side 1 excitati~n does not~
With contlnued reference to Flg. 1, it will be seen that the signal path from each side of each portal is indlvidually coupled (via channels or paths A and A', B and B') to a prea~p 1, from whlch two outputs are separately processed, one being directed to an amplifier/
filter 2 and the other to an inhiblt amplifier 3, whose respecti~e outputs are coupled on paths E and F to a detection lo~ic and processlng module 9~ whlch also receives control signals on path G from a ti~in~
generator 4. Generally speaking, the detection logic module 9 funccions to provide indicAtor and/or alarm signal~ indicative of the presence, within the alternating interrogacion field maintalned between the respec~lve sides of a glven portal, of the desired field-affecting marker member, such indicator or alarm outputs being depicted in Fig. 1 as coupled along a pa-th H to an alarm module which is so marked. Power supply paths are indicated in Fig. 1 as being directed from an outside "power in" source and along a con~on bus 11 to a power supply 8. The latter provides various power levels and types to the preamp 1, the detection logic module 9, the amp/filter 2, the inhibit amplifier 3, the timing generator 4, the phase driver control 5, and the phase driver 6.
The outputs of the phase driver 6 are coupled along the aforementioned paths C and D to portals l and 2, to drive the oscillating interrogation coils located there. ~Yhile state of the art circuits and components which are generally usable as the foregoing functional units are cer-tainly knol~n and available at the present point in time, and are re-ferred to in the aforementioned prior patents, for e~ample~ certain preferred new versions or improvements o such are disclosed hereinafter lS It should be understood that in accordance with the present invention the interrogation coils at the portals are preferably driven at a nominal oscillation frequency o 10 kHz, and that in order ~o maximize detection capabilities in a broad sense, it i9 desirable to drive the two interrogation coils on opposite sides of ~he same portal in an alternating in-phase and out-of-phase sequence, in "bursts" which continue over a desired number of cycles. Thus, for a first such period both sides of portal 1 and portal 2 will be driven in phase, wher0as or the next ensuing such period side l of each will be driven with the same phase as before but side 2 of each will be driven with directly out-of-phase excitation, the effect of which will be to re-direct the resultant direction of the interrogation field by gO~, thus affording detection c~pabilities for particular marker crientations within the field which might possibly be missed or produce very weak detection signals i by chance oriented essentially orthogonal with respect to the direction of flux within the interrogation field. A particular example of a pre-ferred phase-reversal sequencing comprises alternating bursts of 160 cycles ~i.e., 16 msec) of the nominal 10 kHz fundamental alternation, separated by a "dead time" or "inter burst gap" of 4 msec, with the first such 160 cycle burst applied with the s~ne phase condition ~e.g., "phase A") on both sides l and 2 of each portal (i.e., "A-A" phasing), and ~he second such burst applied with l'phase Al' on side 1 and the 8- ~L~ 3~

opposite ("phase B") applied to side 2 of each portal ci.e., "A-B"
phasing). Accordingly, the an*enna at each portal side (constituting the ini~ial "receiving means" herewith) will return in-phase signal components for marker-present conditions within the portal during the first such in-phase drive condition, and out-of-phase marker-present signals during the next such drive condition.
As indicated in conjunction with Fig. 1, the detection signals from the antennae at the various portal sides are coupled along signal paths A, A' and B, B' to the preamplifier 1, a detailed illustration of a preferred embodiment of which is set forth in Fig. 2, to which refer-ence is now made. In the preamp 1, signal pa~hs A and A' from the receivers, or antennae, which monitor side 1 of both portals l and 2, are coupled respectively to preamp inputs P-l and P-2. Conversely, signal paths B and B' from side 2 of both portal l and portal 2 are coupled, respectively, to preamp inputs P-3 and P-4. As may be observed, each such preamp input feeds into an identically-configured amplifying and filtering network branch, located generally within the circuit portion on the left, designated l-A, and each of the four such pre-amplifier/filter network portions feeds into a summation circuit portion on the right, designated l-B.
Generally speaXing, the interrogation field~generating coils are driven, in the alternating-phase sequence noted above, with current pulses on the order of magnitude of approximately 50 amps, preferably once every several oycles of tank circuit oscillation (e.g., every fourth cycle of oscillation), resulting in an oscillation of approx-imately three hundred volts (peak to peak) in amplitude. Each receiving antenna, therefore, would nominally detect a very strong 10 ~J~ si~nal, and for this reason the antennae are pre~erably figure-eighted in winding configuration, so as to null out as much as possible of the 10 l~l~ component. The field perturbations caused by the presence within one of the portals of the permalloy strip or other such marker element are miniscule in comparison to the tan~ drive level, thus presenting very substantial signal-processing difficulties in order to achieve lligh sensitivity, to avoid missing contraband-carried m~rkers, while at the same time achieving a high degree of selectivity, to avoid erroneous contraband or theft-indicatiYe ala~ brought about by any o~ a variety ~ ~ .
.

9~ V,~%~

of metallic objects or articles which also cause perturbations in the interrogation field.
Toward the foregoing end, certain characteristics of marker detection have become knonn which greatly facilitate the sensitivity-selectivity requirements, for example, the drive excitation pulses applied to the field-inducing coils are highly disrLtptive in and of themselves, and it is thus desirable to blank out all or part of the recei~ing circuitry during the time such drive pulses are being applied to the interrogation coil. Furthermore, the actual permalloy or other such low-coercivity markers create field perturbations by switching their magnetic domain orientation each half-cycle of alternation of the interrogation field, i.e.> on each positive-going half-cycle as well as upon each negative-going half-cycle, ~ith magnetic domain switching occurring during the first 90 of current flow in the coils for each such half-cycle. Accordingly, if the antenna ~"receiYer means") signals are examined to deternune the presence of a marker within the field only during the current-rise portion of the cycle (i.e., the first 90~>
other non-marker perturbations may be screened out. Furthermore, i a sample of the antenna/receiver means signals is exanuned during other por~ions of the cycles of interro$ation field alternation, i.e., when marker perturbations are not anticipated (i.e., during the current-falling portion of each half-cycle), a representative ~mbient field condition may be established for comparison with the receiver means signals obtained or examined during those periods when marker signals are to be anticipated if indeed a marker is present within the portal, i.e., within the interrogation field.
In addition to the Eoregoing, the treatment afforded the receiver means signals prior to actual analysis efforts, whose purpose is to deterntine whether or not a marker is present, beco~.es very im-portant to successful processing. That is, while it has heretofore beenreco~tized that the interro~ation field fundamental frequency (here, 10 kHz~ must be eliminated to the fLtllest extent possible, the counter-vailing consideration is to maintain the integrity (fidelity) o the actual signal rom the antenna to the ~reatest extent possible. This desired result is greatly facilitated by the circuit configuration shown in Fig. 2 for the preerred fo}~ of preamp l, in which each separate ~.

9~3~6 pre~mp circuit path proceeding from the differen-t antenna inputs P-l; P-Z, etc., is identical, thus mc~king a description of only one such path necessary. Referring to path P-l, it may be seen ~hat the signals first encounter a Pi-type RC filter 40, l~hich applies an initial attenuation i"f 5 f 6 DB centered upon the lO kHz drive frequency, but does not introduceany appreciable noise content as other filtering might. Next, the receiver signals encounter an amplifying stage 42, which is preferably a low-noise voltage amplifier having a gain on the order of about ten, designated UlO0. In a particular preferred embodiment, the latter is implemented by use of an integrated circuit operational amplifier such as that designated as IC5534 coupled into the circuit in the manner indicated, with resistive feedback. Accordingly, the receiver signals from the antennae essentially encounter strong low-noise amplification prior to operational filtering. Such filtering occurs after the first stage of amplification 42, in the t~in-T notch filter 44 comprised of resistors RlO7, Rl08, RlO9, and capacitors ClO4, Cl05, and Cl06, in which it will be observed that resistors Rl08 and RlO9 are variable in na~ure, to provide for precise setting of the notch characteristics.
Notch filter 44 is centered upon the lO kHz interrogation field fre-quency, and supplies at least 40 DB of rejection for such frequency.
Follol~ing the notch filter 44 in the preamp circuit pa~hs is abuffer stage of amplification 46, which may be implemented by another integrated circuit No. 5534 operational amplifier, conigured to provide unity gain. It is to be noted that both amplifiers 42 and 46 should be wide band amplifiers, so as to accommodate all of the frequencies ~ithin the range e.Ytending from the fundamental of the interrogation field out to at least the fifteenth harmonic thereof. IYhile set forth more fully hereinafter, it is to be noted that the high-order and low-order harmonic content of these antennae signals are utilized as important determina-tive factors in accordance here-~ith by observing the ratio or relative amounts of these bands of frequencies. The lo~er-fre~uency band com-prises primarily the third and fith harmonic range, and to some exten~
the seventh, since this range is highly representative of interrogation field perturbations brought about by non-marker metal objects of many and different particular natures. That is, the actual permalloy or like marker produce5 a significantly different ratio of the higher-order ~9~

harmonic band with respect to the lo~er-order harmonic band, even though both the authentic marker and other non-marker objects may produce varying amounts of both frequency bands in the responses detec~ed from the field perturbations which they cause. Indeed, some particular and relatively unusual metal objects (such as certain plated keys and certain loop-form or mesh-type metal objects) may to a considerable degree "mimic" (that is, resemble) the response of an authentic perm-alloy or like marker, although in essentially every instance the actual ratio of the high order harmonic band to the lo~ order band (as defined above) will be at least somewhat different than those brought about by the authentic marker element.
Each of the preamp circuit paths commencing at the inputs P-l, P-2, etc., thus produces a relatively noise-free and significantly amplified version of the antennae/receiver means signals, with the fundamental 10 kHz si.gnal substantially reduced but ~ith harmonics of this signal present. Each such circuit path has a pair of output resistors Rlll/R112, R211/R212, etc., which are coupled into the summing portion of the preamp l-B in the following manner. First, it will be noted that the Rlll-R211 outputs are ganged together and fed to the inverting side of a differential amplifier V102. This same circuit path is coupled to the output side of an upper switch portion Sl in a four-stage C~IOS analog switch S100, through which signals from preamp path P-3 output resistors R311 and R411 may also be coupled, upon appropriate actuation (excitation) of s~tch control terminal SCl/4, which also controls switch stage S4 of the ~S switch S100.
Output resistors R112 and R212 in preamp paths P-l and P-2 are ganged together and coupled to the inverting input of a second differ-ential amplifier U202, and that signal path is also coupled to the output side of a third switch stage S3 of switch S100. Similarly, output resistors R312 and R412 of preamp paths P-3 and P-4 are ganged together and coupled to the input side of switch stage S3 in switch S100. These same two output resistors, R312 and R412, are also coupled to the input of the fourth switch stage, S4, of the CMOS switch S100, just as output resistors R311 and R411 are additionally commonly-coupled to the input side of the second switch stage, S2, of switch S100.
From the foregoing, it may be seen that the output of commonly . i:

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coupled, preamp paths P-l and P-2, which repres nt side 1 of both portals 1 and 2 (Fig. 1), are coupled to the inverting (i.e., "-") side of both differential amplifiers V102 and U202, and that signals from preamp paths P-3 and P-4, representing side 2 of both poTtals 1 and 2, will also be applied to this same side of differential amplifiers Ul02 and U202 ~as outputs from resistors R311 and R411) when either the first stage (Sl) or the ~hird stage (S3) of switch S100 are actuated, through energization of their respective different control terminals (i.e., SCl/4 or SC2/3). At the same time, the non-inverting (i,e., "~") side of di~ferential amplifier U102 is coupled to receive the outputs from side 2 of portals 1 and 2 (preamp paths P-3 and P-4, from resistors R311 and M 11) whenever the second switch stage (S2) of s-~itch S100 is triggered by a signal on s~itch control terminal SC2j3, and the analogous (non-inverting) side of differential amplifier U202 will receive the side 2 ~paths P-3 and P-4~ output signals (from resistors R312 and R412) upon actuation of the fourth stage (S4) of switch S100, by an appro-priate signal on switch terminal SClJ4. For purposes of this specifi-cation, these control signals applied to switch terminals SCl/4 and SC2/3 may merely be considered as comprising appropriately-timed gating signals produced by the timing generator 4 of Fig. 1 which are closely synchronized to the frequency and phase of the oscillations actually present at the interrogation field.
Accordingly, by supplying the aforementioned gating signals to the switch terminals of CMOS switch SlO0, the antenna signals from the opposite sides of the two portals will be constructively s~m~ned (magni-tudes instantaneously added) in one path and con~ersely, "destructively summed" or differenced in another path, to provide t-~o quite different but nonetheless related signal outputs cn preamp output terminals P-5 and P-6, leading from differential amplifiers U102 and U202, respectively.
hlore particularly, when both sides of each portal are being driven in-phase with one another, the outputs from their respectively-associated antennae are in phase and are directly added (summed) by operation of the first st~itch stage Sl of switch S100, an appropriate control signal being suppled at that time to control terminal SCl/4. Under these conditions, differential amplifier U102 has all four such amplified, in-phase antenna signals applied to its inverting input, and none applied '' ~2~

to its non-inverting input. ~he control signal applied to terminal SCl/4 also actuates the fourth switch stage, S~, thus applying the Yery same output signal from paths P-3 and P-4 to the second differential amplifier, U202, but in the reverse manner, i.e., applying such signals to the non-inverting inputs, whereas the other two antenna signals are applied to the inverting inputs, so that these two sets o signals are differenced, or subtracted, at differential amplifier U202.
When the opposite phase relationship occurs at the inter-rogation fields (i.e., side 2 of both portals driven out-of-phase with side 1 thereof), a similar end result is obtained through opposite switching of the analog swi~ch S100. That is, additive summation occurs at differential amplifier U102 and subtractive sun~ation at amplifier U202, i.e., signals from side 2 of both portals are arithmetically added together at amplifier U102 (directly out-of-phase signals applied to opposite inputs of the differential amplifier~ while the same signals are arithmetically subtracted at amplifier U202 (i.e.~ directly out-of-phase signals resistively combined and applied to the inverting input, and no signal applied to the non-inverting input). Accordingly, the output appearing at preamp ouput terminal P-5 represents the algebraic difference but arithmetic sum of the antenna signals from sides 1 and 2 of the two portals, whereas the output at preamp terminal P-6 represents the algebraic summation but arithmetic difference of the antenna signals from opposite portal sides, taking into effect the alternating phase conditions present within the interrogation field. In this connection, it is to be noted that the control signals applied to terminals SCl/4 and SC2/3 and the C~S switch S100 are preferably o sufficient duration to maintain switch actuation througilout tlle entire time interval from the initiation of one phase condition t~Dr example A-A) to the initiation of the next succeeding phase condition ~continuing the example, A~B).
~lat is, the "on" time for the s~itch should preferably continue through the aforementioned dead space or interburst gap, rather than ending at the immediate conclusion of the ongoing phase condition, since by continuing the summing and differencing operation on into the "dead space", additional signal information will be obtained with respect to interrogation field perturbations which will contribute meaningfully to system sensitivity and selectivity.

Quite clearly, ~he t~o signals appearing on preamp output terminals P-5 and P-6 will have substantially different çharacteristics, the first such output representing combined antenna outputs providing the highest possible signal-to-noise characteristics, for maximum S sensitivity, whereas the output on the second such terminal has had eliminated from it "common-mode~ noise and other such undesired fre-quency components. The first aspect is particularly important ~ith respec~ to the weakest likely marker-present conditions, i.e., a marker whose particular metallurgy and/or physical characteristics produce very wea~ perturbations of the interrogation field, and ~hich is located in the middle of the portal, midway between the two sides where the re-ceiver means antennae are located, a set of conditions likely to be missed in prior systems. Of course, by use of a separate preamp circuit or path for each portal side receiver~ i.e., antenna, sensitivi~y is optimized in any event, and even this basic actor has been lost upon certain of the prior systems; this is all the more true when the par-ticular preamp circuit path configuration, as described above, is taken into consideration, since this optimizes the desired signal ~i.e.9 hannonic frequency bands with the least introduction of noise). Of course, the summing (additive and subtrac~ive) described above is a further and very substantial enhancement for systems such as those in use or proposed heretofore.
Some of the reasons underlying the above-described differ-encing of the co~non-mode noise signals from the two different sides of a single port~l result from the fact that, in contrast to the true or real ~e.g., pel~alloy) marker, most other objects or materials which would have a low enough coercivity to generate harmonics of in~erest (i.c., tending to mimic or mask a true marker) also have lo~ permea-bility and only cause a perturbation in the interrogation field when in close proximity to the portal sidesl i.e., to an interrogation ~ield-generating coil or to an antenna or rcceiver. lhus, a non-marker object carried through a portal causes a pertur~ation in the field as the Gbject passes near the interrogation coil. At the same time, there are many objects or structures ~hich may be present beneath the floor, or example, wire mesh, concrete reinforcing rods, etc..., or in the ceiling ~e.g., ceiling grids) whicll cause receiver signals in the 30 to 50 kH2 . r, !r ' ., '` -15- 9L~ 3~3~

region, i.e.~ the third and fifth harmonic of the 10 kHz drive fre-quency fundamental. Addi~ionally, even distortion in the capacitor geometry and the coil geometry of the L-C field drive circuit are likely, due to the high magnetic fl~Y densities, ~o produce receiver signals in the 30 to 50 kHz region. I~ is desirable to lower this background or ambient noise level, so tha~ field perturbations caused by non-marker items passing through the portals very near the interrogation field coil at one side would have the greatest detectable effect, i.e., would be more easily and more reliably detected, and thus discriminated out of alarm-causing effect.
To this end, the receiver tantennae) signals from opposite sides of the same portal are "summed" (i.e., combined) in accordance herewith such that during each particular interrogation field phase condition such signals are arithmetically subtracted from one ano~her, or differenced, that is, they will (at least partially) cancel out one another. The signals coming from the two ante~lae are much alike, and if the two signals, properly phased, are summed together so that one cancels with the other, the result will be to lo~er the signal portion attributable to "background" or environmental noise, thereby enh~ncing, or highlighting, perturba~ion effects from objec~s located near one portal side. On the other hand, a true marker causes a significant perturbation even as it passes down the center of a portal, and it is desirable to enhance those perturbation effects. To accomplish such enhar.cement, the antenna signals from opposite sides of the same portal are constructively ~algebraically~ added in the preamp and processor to produce a dif~erently-constituted second composite or resultant signal for subsequently processing, containing all of the perturbation effects present at either antenna and therefore maximizing the result produced by a real marker.
Thus, the present invention provides an appreciation and realization that the only time a non-marker object i5 likely to produce perturbations with a ha~onic response fairly closely mimicing that caused by an actual marker is when the object is close to one of the portal sides. Tha~ is, because of the comparatively low permeability of t` 35 most non-marker objects, they do not have as large an ~ fect on the interrogation field as the permalloy strip o a true marker does, and so ~- -16~ 9~

only generate significant harmonics when interrogated with ~ very strong field. When so interroga~ed, a non-marker may ac~ually generate some of the same harmonics as a marker, but not to the same extent, and not in the same ratio of harmonic orders, and non-marker objects Yill thus be S detected to a greater ex~ent when the object is close to one portal side. It is important to realize, moreover, that perturbations caused by non-markers do not have the same distribution or ratio of harmonicsJ
and that is why it is desirable to produce, and compare, the two dif-ferent pre~np output signals, as done in accordance herewith. Further-more, while a non-marker will theoretically produce the same distribu-tion of hanmonics, or the same harmonic content, ~hether it is in the middle of a portal or close to one side, the permeability of such an object is likely to be such that its magnetic domains do not even undergo switching by the interrogation field if the object is near the center of the field, whereas a true marker will still undergo substan-tial saturation and domain-s~itching under such circumstances. That is, the strength of the interrogation field does differ across its width~
but the perturbation effect of any object is really a function of two factors: first, coercivity, which for non-markers is mos~ likely not as low as that of the real marker, requiring a stronger field to cause domain-switching; second, non-marker objects do not have as high a permeability as real markers, ~Id non-marker objects do not disrupt the field as much when they do undergo switching. ~IUS, there is a double effect as an object moves away from a sid~ of the portal, and the effec~s caused by non-markers fade away very quickly.
The above-described separate outputs from preamp terminals P-5 and P-6 are separately and respectively appli.ed to the i~libit amplifier 3 and the ~nplifier/filter 2 noted above in conncction s~ith Fig. 1, w~lere each such output is separately processed (basically, amplified and frequency-shaped), and the resulting outputs are then separateIy supplied to the detection logic and processin~ ~it 9.
~ lore particularly, the output signal from pre~p ter~inal P-6 is applied to input terminal A-l of the inhibit amplifier 3J a preferred embodiment of which is illustrated for convenience in Fig. 3. BasicallyJ
inhibit amplifier i is preferably a tlio-stage band pass ~mplifier whos~
pass ba~d encompass~s primarily the third and fifth ha~onic, and -17- ~L~ 98~3~

preferably the seventh as well, of the alternating interrogation field frequency ~in the preferred embodiment already noted, 30 to 50, and up to abou~ 70 kHz), which generally characterizes the lower frequency spectr~n in the ratio used to critically identify the particular marker element within the interrogation field~ as e~plained more fully herein-after. It will be noted that both the input terminal A-l and the output terminal A-2 are subject to switching by being coupled through the complementary halves of a Ch~S analog switch S300, which may advan-tageously be implemented by a single four-stage such switch, the two complementary halves of which are shown for purposes of illustration at different positions in Fig. 3 (i.e., one at the input and one at the output). Additionally, the input tenninal A~l, after being switched through swi~ch portion S5 of Ch~S switch S300(a), is coupled to a twin-T
notch filter 310 preferably having a variable resis~ance in both its series-connected and parallel-connected branches. Like the twin-T notch filter 44 noted above in connection with the pre~np 1) notch filter 310 is used for the purpose of further removing, i.e., diminishing, ~he lO
kHz fundamental frequency o~ the interrogation field, since the effects of the latter are very strongly present in the receiver signals picked up by the various antennae, and require substantial effort ~o properly filter out for optimum sensi~ivity and selectivity in marker det~ction.
By the variable-resistance twin-T filtering CQncept no~ed, another 40 DB
of rejec~.ion in the residual level of the 10 kHz signal may be accom-plished, with desirable results.
As noted, the input to the inhibit ~nplifier 3, and the output from such amplifier and circuit, are both subject to switohing by the c~nalog switch S300. ~lis switching is provided for bl~nking purposes, during which the inhibit ~nplifier may be effectively removed from operation at certain critical points in the operation of the system, i.e., when the interrogation field-generating coils are receiv~ng their drive pulses. Such blal~ing is accomplished by appropriately timed inputs on C~S s~itch control tenninals SC-10 and SC-12, the first o ~hicll blanks the input cmd the 5econd of whicil bl~ks ~he output.
Signals for these two control te~inals are provided from the timing generator 4 noted in connection with Fig. 1~ and may generally be considered as pulse-t~e blanking signals whose pulse-w~dth determines ;~, iL~ 9 ~3~

the time of circuit shutdo~n, the timing of the bl~nking signals being synchronized to the application of the aforementioned excitation or drive to the field-producing coils. In a more particular sense, the blanking signal applied to control terminal SC-12, at the output of the inhibit amplifier, is preferably about 50~ longer in duration than the signal applied to switch control terminal SC-10, which blc~nks the input of -this amplifier. In a particular sense, where the interrogation field f~mdamental frequency is 10 kHz and one quarter-cycle ~during which time the drive pulse is actually applied) has a duration of 25 microseconds, a preferred input blanking period is on the order of 100 microseconds, and a preferred output blanking period is 150 microseconds, both signals synchronized to the drive pulse. By so doing, transients produced in the amplifier as a result of switching will have been avoided, and both the amplifier and the LC oscillating circuit will have undergone sub-lS stantially complete settling, thus avoiding distortion effects which canbe very significant.
The output from the inhibit ~nplifier 3 comprises carefully-timed bursts of the frequency range representing primarily the third, fif~h and seventh harmonic of the interrogation field fund~nental, as noted above, and this output from terminal A2 of the inhibi~ amplifier is applied ~o input terminal DL-4 of the de~ector logic circuit 9 ~Figs.
4, 5 and 6), to be described further hereinafter.
The second output from the preamp l, namely that appearing on its output terminal P-6, is applied to the amplifier/filter 2, noted previously in connection with Fig. l. Although not illustrated specifi-cally in the drawings, it should be understood that the ampffiltcr 2 has an input terminal to which the preamp signal (from output terminal P-6) is applied. Preferably, the amp/filter 2 i5 a three-stage band-pass device, having a single-ended output which is coupled to the detector logic circuit 9. With respect to the preferred characteristics of the amp/filter 2~ the three stages of amplification may all be implemented by use of an L~-318 integrated CilCUit operational c~mplifier, connected in a multiple-pole c~mplifying configuration with appropriate frequency-shaping capacitance, centered upon the desired pass band comprising the fifteenth hannonic of the fundamental frequency at which the interrogation field is driven, in the embodiment contemplated here T

~ -19~ 9~3~

approximately 140 kHz. In a preferred configuration, the first stage is a high pass stage, Lhe second stage is a band-pass stage, and the third stage is essentially a gain stage with both high and low alts. l~here integrated circuit amplifier stages are used, each succeeding stage is preferably coupled in complementary conductance configuration, ~ h appropriate positive-negative-positive reference or biasing voltages.
As already indicated, the output from the amplifier/filter unit 2 is coupled to the detector logic network 9, where it is inputted on termi-nal DL-l.
Referring now to the detection logic net~ork 9, and initially to Fig. 4 which illustrates the general nature of a preferred form thereof, it ~ill be observed that this system has five discernible branches, designated by the numerals 900, 910, 920, 930, and 940, which are set apart from one another in this figure by dashed lines, for purposes of illustration. 0f these, branches or sectors 900 and 930 are essentially the same as one another from the standpoint of componentry, although having very definite operational differences ~o be noted subsequently. That is, both branches 900 and 930 embody a control switch 10, 1?, respectively, a reference control ancl threshold compara-tor set 14, 1$ and 16, 22, respectivelyJ and a driver, timer, and indicator unit or circuit portion 22 and 74, respectively, each of the latter having respective output terminals 21 and 25 as well as LED
signal elements ("~FD 2" and "LED 3", respectively). As further seen în Fig. 4, the respective outputs from the threshold comparators 18 and 22 are also ~irected to an integrator 34, and thus are seen to be summed with respect to one another; however, the particular manner in ~Ynich such summing is carried out is an importc~nt aspect and is described in much greater detail hereinafter ~ith continuing reference to the block diagram of Fig. 4, and to the general at-tributes of detector logic unit 9, the center.circuit portion 920 includes an ampli-fying and integrating, or integrating-de-tector, circuit portion 30, ~hich receives an input from teI~linal DL-4 ancl has an output directed to a comparator and alarm 32 having ~n LED
indicator ("LED 1") as one oE its outputs The output from this alanm is also fecl as an input to the lo~er circuit branch 940, more particular-ly, to a driver, timer and alarm ~mit 38, whicll as indicated provides ~2~39~
an "Alarm Output No. 2". This same input terminal of the alarm unit 38 receives control signals on an input lead 39 connecting to the "signal gate" and "noise gate" inputs fed to control switches 10 and 12 from circuit input terminals DL-2 and DL-5. The second (upper) inpu~ terminal of the driver, ~imer and alarm unit 3S is coupled back to the input side of a discharge clamp 26 in path 910, whose primary input is f~om circuit terminal DL-3. The output of the discharge clamp 26 is coupled to, and directly affects, ~he integrator 34, and the integrator is coupled to, and actuates, a comparator, timer and alarm 28 having a primary alarm output directed to a lamp driver 29, which also provides a s~itched alarm Output, labeled Alarm Output ~ \lso, timer and alarm 2~ controls an indicator LED ("LED 4") coupled to its output.
Referring not~ in more detail ,o the detector logic circuitry as depicted in Figs. S and 6, it will first be noted that the upper and lower portions of the circuit, comprising channels 900, 910, 93Q and 940 in Fig. 4, are depicted in Fig. 5, ~hereas the central portion of the circuit, comprising the path designated by the numeral 920 in Fig. 4, is depicted separately in Fig. 6. In the preferred embodiment shot~n in these Figures, the elements identified as "control switch 1" ~nd "control switch 2" in Fig. 4, and desic~nated by the numerals 10 and 12 therein~
are sho~n to comprise input s~ntchin~ transistors Ql and Q2, whose bases receive control inputs through resistors R8 and R9, respectively, from circuit input terminals DL-2 and DL-5. Also, the bases of switching transistors Q, and Q2 are coupled together through resistors R4 and R10, and the junction of the latter two resistors is coupled to the low voltage side of a pull-up resistor R21, and then through conductor 39 to the positive or non-inverting side of an amplifier Ull in path 940. The primary signal inputs to be switched by transistors Ql and Q2 are received Oll circuit input terminal DL-l, which is coupled to the collector of each such transistor through resistors R6 and R5, respectivelyr The "reference control" componeIlts or units 14 and 16 of Fig.
4 are seen in Fig. 5 to comprise switches, e.g. transistors, Q4 and Q3, respectively, ~hich are connected in emittel-follower configuration, and whose bases are coupled together by a lead 49 so as to re`ceive a common input, to be described subsequently. Ihe respective outputs from transistors Q4 and Q3 are coupled as reference inputs to threshold -21- ~L~ 3~39 6 comparators U-la and U-14a, and it is to be noted ~ha~ the circuit arrangement of paths 900 and 930 is of an inverted configuration, i.e., the output ~rom transistor Q4 in path 900 is applied as an inverting input to comparator U-la, whereas the output from transistor Q3 in path 930 is applied to the non-inverting input of comparator U-14a. Each such comparator input also receives a particularly-set reference voltage obtained from ~he junction of voltage-divider resistors R14 and Rl, and applied through input resistances R16 and R3, respectively. The respective opposite input terminals of threshold eomparators U-la and U-14a receive inputs from the collectors of switching transistors Ql andQ2, respectively. These inputs are also supplied to comparators U-lb and U-14b (which may be half of the same double integrated circuit amplifier comprising comparators U-la and U-14a, respectively, for example, an integrated circuit comparator No. 339). In essence, the second comparators U-lb and U-14b are used as drivers for ensuing timers and indicators U-2a and U-2b, whQse primary function is merely to time out or an indicator drive signal of desired duration on respective signal lamps LED 2 and LED 3, as described hereinafter. As indica~ed, the two timers U-2a and U-2b are interconnected to one another9 and they may in fact be implemented as the complementary halves of an IC5S6 timer, which is a double unit.
The lowermost circuit portion 940 of the detector logic network 9 comprises in effect a comparator, acurrent source which drives a ganged double-timer, and an amplif;ed l~np-driver output for alarm signal purposes. More particularly, the initial comparator comprises the aforementioned comparator unit U-ll, which may be implemented by use of a 339 integrated circuit component. The comparator output is diode-coupled to a transistor QS disposed in grounded-collector configuration to act as a timed current source whose timing cycle is detennined by the charge rate on capacitor C14. This current source drives the double-timer U-i2a and U-12b, w}lich may advantageously be the two halves of a No. 556 inte8rated circuit timer whose terminals are ganged in the mam~er illustrated~ The first half of the timer, U-12a, is diode-coupled ~D7) to a final amplifier or driver U-13 and lamp driver Q6, driver U-13 being a further comparator component which may be imple-mented by use of a 339 IC whose non-inverting input is supplied by the 22- ~L~ 3~3~

same signal ~ihich is applied to the inverting side of the first-stage amplifier U-ll. Further, the first timer stage U-12a is connected to (diode OR'd with) the second timer stage U-12b such that the first stage, upon its initial excitation, immediately co~nences a continuous lamp-driving operation of amplifier U-13 and switch Q6, as a "pot~er on"
indicator; ho~ever, whenever the current source comprising transistor Q5 and its ~iming capacitor C14 reaches full charge, the second timer (U-12b) is gated in and assumes control of the output signal, causing a blinking of the signal lamp driven by driver Q6, for purposes noted subsequently.
Generally speaking, the operation of that portion of the detector logic circuitry described above is as follo~Ys. Input tenninal DL-l receives the above-described output from the amp/filter 2, IYhich as already pointed out comprises the arithmetically-summed antelma signa:Ls from both sides of a given portal, or group of portals, after band-pass amplification centered upon the fifteenth harmonic of the interrogation field fundamental frequency. This signal is supplied equally to the control switches Ql and Q2, whose switching operation determines whe~her or not any portion of the supplied signal is gated ~hrough the switching transistors to either path 900 or path 93Q. The latter two channels are gated into and out of operation by timing signals applied to inputs DL-2 and DL-5, as supplied from ~he timing generator 4. The first such input, to transistor Ql, is representative of the "signal gate" or "marker signal ~indow", i.e., those particular increments of time represen~ing an increasing-current condition ~both positive-going and negative-going) in the interrogation field drive coils; conseq-lently, these gate signals represent times l~hen a marker-present signal ~s likely to be present in the signals from the portal antennae, if a marker is in fact present ~ithin the interrogation field. Conversely, the gating signals applied to terminal DL-5 and transistor Q2 represent the opposite portion of the interrogation field alternations, i.e., when marker-present signals are not likely to occur in the antemlae sign31s even il a marker is present in the portal. Conse~uently, the gate signals applied to te~ninal DL-5 define a "noise gate"~ i.e., a period of time during which the signals received by the portal antennae~ on an instantaneous basis, represent an actual measure of the e~is~ing noise ~_ -23-~g~

level in the antennae signals.
It should be noted tha~, in accordance with this invention, the duration o the aforementioned "noise gate" is shorter than the duration o the "signal gate", and that there is a gap or interval between the two. More particularly~ assuming the interrogation field fundamental frequency to be 10 kHz, so that the duration of each quarter-cycle is 25 microseconds, the marker-present signals are likely to occur during the quarter-cycles when the current is increasing, eitheT posi-tively or negatively, whereas the current-decreasing quarter-cycles represent the condition when marker-present signals are not likely to occur in the antennae signals. By maintaining the "signal gate" for a full 25 microseconds but maintaining the "noise gate" for only approxi-mately half that time, thus providing a gap of approximately 12 micro-seconds between each noise gate and ensuing signal gate, distortion and transients which otherwise would "ring through" the circuit will be eliminated, thus further enhancing sensitivity and reliability. Of course, the particular timing and synchronization of such signals are also highly important. While the general state of the art includes circuits and components well able to provide representative gating or blanking signals of this type, a preferred form of timing generator is a digital clock and divider, synchronized to the actual oscillation conditions of the interrogation field.
Since the inputs to terminals DL-2 and DL-S occur at different points in time, and in effect represent an alternating sequence, circuit paths 900 and 930 of the detector logic 9 in effect alternate in opera-tion, and during the period each is in operation it applies an input to the aforementioned thresholcl comparators U-la (and U-lb) tin channel 900) and U-14a (and U-14b) (in channel 930). In so dGing, each such circuit path functions to alter the charge state of an integrating capacitor C9, and it is important to note that the two circuit paths act oppositely from one another in that regard. That is, the switched input from transistor Ql to comparator U-la in path 900 is applied to the non-inverting (i.e., positive) input, whereas the opposite is true in path 930, where the signals gated through by switch Q2 are applied to the inverting side of comparator U-14a. Consequently, the two such CiTCUit paths act to raF)idly and sequentially apply increments of charge to, and --24- ~L~ 9 ~3~

draw increments of charge from, integrating capacitor C9, on an alterna-ting, increment-by-increment or pulse-by-pulse basis. As will be seen hereinafter, these added and subtracted increments of charge are not necessarily equal in magnitude, and the resultant charge state on the integrating capacitor is thus cumulative with respect to time during each "burst" of pulses, so long as they are of the same phase, as described more fully hereinafter.
It is very i~nportant to note, in conjunction with the alter-nating operation of circuit paths 900 and 930 noted just above, that the signals gated through by transistors Ql and Q2 to comparators U-la and U-14a work against variable reference levels, and ~hat these variable reference levels are applied to the opposite-polarity input terminal of each such comparator. That is, in circuit path ~00 the inverting input of comparator U-la receives the variable reference level~ ~hereas in circuit path 930 it is the non-inverting input of comparator U-14a which receives the other such variable reference level. As already indicated above, these variable reference levels both operate f~om the same nominal or steady-state reference levelJ obtained from the junction of voltage-divider resistors R14 and Rl~ through identical series resistors R16 and R3. This steady-state reference level is subject to variation, ho~ever, by the operation of transistors Q4 and Q3, which constitute the "reEerence controls" 14 ~nd 16 noted in connection with Fig. 4. ~lat is, in channel 900 the base of transistor ~4 ls controlled, in a manner described more particularly hereina~ter, so as to vary the resulting reference level applied to the inverting te~ninal of diferential amplifier U-la. In channel 930, the steady-state re~erence level is applied to the non-inverting input of comparator U-14a, and this nominal reference level is made subject to variation by reference control 16, i.e., transistor switch Q3, ~}liCh receives the same varying input as transistor ~4, i.e., the base of each of these transistors is cor~nonly coupled to receive the same control input signal (frcm the output of the amplifier, peak-detector and integrator 30 in pat}l 920, sho~nn in Fig.

5). In the case of both circuit paths '900 and 930, the second-stage comparatorsU-lb and U-14b, respectivelyl may be considered as in cssence duplicative of the first such stage, insofar as inputs are concerned, except that instead of applying and subtracting charge from integrating -25- ~ g~9~i capacitor C9, they are utilized to drive indicators LED 2 and LED 3, which are pulsed by timer units U-2a and U-2b, to indicate the opera-tional status of each such circuit path.
The second portion or channel 910 of the detector logic network 9 is also illustrated in detail in Fig. S, and will be seen to include a pair of inputs, a first one of which is provided by circuit input terminal DL-3 which is coupled to the inverting input of a compara-tor U-3, comprising the "discharge clamp" 26 noted in connection with Fig. 4. The outpu~ of this comparator connects to the conductors 21 and 23 by which charge is applied to and removed from integrating capacito~
C9. Therefore, when an appropriate gating signal is applied to the inverting side o~ comparator U-3, under general system conditions to be noted subsequently, this comparator/amplifier will clamp integrating capacitor C9 to ground, thus f~ly discharging the integrator. This in lS effect terminates, and dissipates, the incrementally-accumulated charge effect carried on for the duration of each different phase condition present in the interrogation ield, as noted above. Therefore, each time the interrogation field-inducing coils are to be switched from one phase condition to another (for example, from an in-phase or phase A-A
condition to an out-of-phase, or phase A-B condition), an appropriate pulse supplied from the timing generator 4 is applied to input terminal DL-3, to fully discharge the inte~rating capacitor C9. Durin~ the time each opposite phase condition exists (described previously as preferably on the order of 16 msec, representing lS0 cycles of altema~ion) the charge state existing on integrating capacitor C9 is continuously subject to pulse-by-pulse change, depending upon the operational levels of circuit paths 900 and 930, described above.
Animportant function of the detection logic and processor 9, involving that portion thereof designatet generally as channel 910, and particularly of that portion of the circuitry disposed to the ~ight of portal point 911, is the production of a desired alarm or signal upon the detected presence of the particular marker within the interrogation field~ hlore particularly, it will be noted that the node or junction 911 ~here comparator U-3 interconnects with conductors 21 and 23, which lead to the integration capacitor C9, comprises the signal input to a comparator U-4, which receives a predetermin~d bias or steady-state -26- 3L~ 3~3~

reference on its positive (non-inverting) input terminal from voltage-divider resistors R13 and Rl5. Therefore, at any time the charge level on integrating capacitor C9, representing the relative proportions o higher-order harmonic content in the antenna signals versus lower-order harmonic content therein, caused by an object producing perturbation of the interrogation field, rises to a predetermined level~ established by the reference applied to comparator U-4, this comparator triggers and applies an alarm-causing output signal to tlle timer U-5 (which may be an IC No. 555). Cne output of timer U-5 actuates an alarm signal ~LED ~) and is also coupled to an output driver Q7, which may be used to drive a signal lamp, sound an audible alarm, or the like, utilizing an ou~put taken at terminal 912 connected to ~he collector of transistor Q7.
Furthers a switched output signal of timer U-5 which is representative of the control signal applied to the base of transistor Q7 is available on the output terminal designated 914.
Perhaps the most important of the many important functions of the detection logic and processing network or unit 9, is carried out on circuit path 910, sho~ in more detail in Fig. 6. As sho~n there, this circuit path receives an input on terminal DL-4, which input comprises the amplified, frequency-selective output from the inhibit ampliier 3, noted generally in connection with Fig. 1 and more particularly described in connection with Fig 3. This signal from the inhibit amplifier 3 comprises sequential, time-gated, synchronized bursts of the subtracted (differenced) signals from the portal antennae, afler low-pass selective amplification thereof in the inhibi~ ~nplifier. Consequently, this input to the~detection logic circuit is representati~e o~ the low-frequency component band (in the range of the third, fifth, and Up to the seventh harmonic of the interrogation field, here on the order of 30 to 50, and approachir.g 70, ~Iz), which signal is attributable to an object ~ithin the interrogation field. Whether that object is-an actual marker, or merely some non-marker element causing perturbations in the interrogation field, remains to be determined, but as already indicated, the true or actual markers will have a relatively unique ratio or balance of the high frequency component band with respect to ~he low frequency band. This low frequency band is used in the detection logic and processing unit 9 as a determinant which must be satisfied by the lZ2g89S

magnitude of the high frequency band produced by ~he same object within the interrogation field before a marker-present signal or alarm is sounded; i.e., the amount (magnitude) of the low frequency band actually encountered, as represented by the magnitude of the input applied to terminal DL-4, is used to determine the required level ~hich the high frequency band produced by the same object in the field must equal or exceed if it is indeed an actual marker; the ratio or balance of these frequency components for true markers being relatively uni~ue.
To achieve the above result, the aforementioned input on terminal DL-4 is coupled to one end of a variable resistance or poten-1:iometer R2, whose movable contact is coupled through a series resistor R56 to the inverting input of a differential amplifier U-6 coupled into the circuit as an inverting amplifier, whose gain is thus set by po-tentiometer R2. Inverting amplifier U-6 ~which is preferably imple-mented by use of a 3240 integrated circuit operational amplifier) formspart of the amplifier, detector and integrator unit 30 no~ed brie1y al)ove în connection with Fig. 4; thusy the output of inverting amplifier U-6 is diode-coupled through a series resistor R48 to the parallel combination of a second inverting ampli~ier U-7 and an R-C integrating network consisting of resistor R50 and capacitor C28. This over~ll network in effect comprises a combination amplifier, peak-detector and integrator, or in effect an integrating detector. That is, the charging time-constant for capacitor C28 is a function of the voltage drop across series resistor R48. Thus, the charge on capacitor C28 ~ill build during the continuation of each burst of input signals applied to te~inal DL-4 and passed by inverting amplifier U 6, with capacitor C28 integrating only the peaks of the negative excursions of the incoming signals ~i.e., that portion of a cycle wllich exceeds the preset refer-ence ]evel).
The peak-integration or inte~rating detector effect just noted is preferably accomplished by maintaining the integration time constant or capacitor C28 of very short duration, for example by utilizing a .1 microfarad capacitor for ~28 ~nd a 4.6 K-ohm resistor for R48. This will produce a very fast~acting integrator which will operate in the manner of a current source, i.e., integrating for only the irst few excursions and tracking the applied signal very accurately and clo5ely~

-2~ 3~3~

yet reclucing the effects of narrow, high-amplitude spikes due to switching tr~tsientS from the blanking ~gate-generating) and other related circuitry. This type of detector is preferred since the band-pass stages preceding it allow some of tile higher order components ~for example in the range of 14n k}lz) to pass through, usually in the form of spikes. Additionally, spikes may be created by the blanking circuitry, as just indicated. If a more conventional peak-detector was used, such spikes would result in a high level of detected signal, whereas the preferred integrating detector responds more to the average value above the diode (D-10) voltage drop. The level of the signal so integrated appears on conductor 48, on the output side of inverting amplifier U-7, and this signal level is not only coupled forward to a ccmparator U-8, but is also reflected back (on conduc~ors 50, 47 and 49) as a threshold-changing signal to transistors Q3 and Q4, (i.e., "reference contro j" 14 and 16) noted above in connection with Figs. 4 and 5.
The forwardly-coupled signal from inverting amplifier U-7 is applied to comparator U-8 and, when this signal rises to a predetermined level constituting a system override condition, comparator U-8 switches, thereby energizing an indicator labeled "~FD 1", through a series resistance RS2 and a level-setting resistor R53. This pTovides a visual indication that the charge level on integrating capacitor C~8 has - reached the override threshold voltage determined by comparator U-8.
Furthermore, the output of comparator U-8 is coupled to the inverting input o differential amplifier U-9, to whose output is also coupled the anode side of LED 1, and the resulting output from ~mpli~ier U-9 is coupled to one input of a second inverting amplifier U-lO. ~le output of amplifier U-10 is coupled back, on conductor 901, to the non-inverting input of the aforementioned amplifier U-ll in path 940 (Fig. S~, whose function has been described previously, and also coupled back ~on conductor 39) to the bases of switching transistors Ql and Q2,-to bring about system override, or lockoutJ as will bs noted subsequently.
Accordingly, it will be seen that the input to terminal DL-4 of channel 920, representing the lower-fre~uency spectrum produced by the interrogation field-monitoring antennae, is utilized, with appro-priate processing, to accomplish two distinct purposes. First, thepeak-detected and integrated reflection of this input is coupled back to -29- ~ 9~3~

the bases of threshold reference-setting transistors Q3 and ~4, to change the threshold level of comparators U-la and U-14a as a direct function of the instantaneous level of the low frequency spect~m produced by an object detected in the portals. Of course, the effect of this is to chan~e in a very significant way the amo~ts of charge applied to and accumulated on integrating capacitor C9. This directly changes the relative conditions under which a logical decision is made to either produce or not produce a marker-detection alarml through ~hat portion of circuit path 910 coupled to node 911 and including comparator U-4, timer U-5, and output driver Q7. That is, in the manner already described in a qualitative sense above, the determination that an object within the interrogation field is a genuine marker is made to be directly dependent upon the relative proportion of the high frequency spectrum (in the neighborhood of the fifteenth hannonic) with respect to the low frequency spectrum (primarily third and fifth harmonic~ which that object is producing in the interrogation field. In this manner, by using the ratio of the detected frequency component bands as the re-quisite detection criteria, substantial and accurate discrimination is accomplished betwe~n actual markers and the myriad of other objects which produce more-or-less analogous interrogation field perturba~ions and which, if detected and indicated as being real markers, would provide a false and erroneous output indicating the~t~ pilfering, or the like where none was in fact taking place.
In accordance with the foregoing, it will now be appreciated that the present detection system provides a multiple-step or multi-layered approach for highly sensitive and yet hi~hly selective detection of the lo~-coercivity permalloy or other such marker within the interro-gation field, based upon the inevitably characteristic ~nd relatively unique balance or ratio of low-order hanmonics versus high-order har-monics caused by the magnetic domain-switching of the ma~ker in response to each ensuing half-cycle of alternation of the interrogation field.
IYhereas many or even most metal objects will have some of the low-order harmonic band, and may even have an appreciable quantity of the high-order band, ew if any non-marker objects will have the same charac~er-istic ratio of high order to lo~ order harmonic bands or componentgroupings; generally speaking, the higher-frequency harmonic b~ld -30- ~L~ 9~3 will be deficient in objects and articles which are not true markers, even through in a general sense substantial quantities of the higher-order harmonics may indeed be present, particularly in objects and articles ~hich provide multiple magnetic paths or loops and which include at least some arcing points, i.e., gaps in the magnetic circuits.
l'hus, the invention provides a method and means to determine the low-frequency components or band and the high-frequency components or band of an object within the interrogation field, and these lo-~-frequency components are used to dynamically control the detection threshold of the high-frequency components which produce an alarm signal. In so doing, the signals from the antennae monitoring the interrogation field are carefully processed to p~oduce two different types of signal output: one which represents the summation of the signals from the antennae, for maximum sensitivity, and the other of which represents the differencing of the signals from opposite sides of the interrogation field, for maximum selectivity. These two signals are separately processed to emphasize their respective high- and low-order harmonic content9 and the signal with the high-order harmonic band is time-sampled in a manner such that the resulting samples are likely to accurately portray marker-presence signals on the one hand and marker-absence or ambient-level (noise-level) signals on the other hand. The resulting s~mples are then separately compared to a varying threshold reference ~hich is pro~ided by a peak-integrated si~nal representative of the detected object's low-order harmonic band, such that the higher ~he level o tlle latter signal, the higher the level which the marker-present signal must have in order to bring about a threshold-crossing in either of the two marker-present or marker-absent signal channels.
I~hatever threshold crossings do result from the foregoing process are then in effect differenced and the result integrated cumu-latively over the repeated cycles o~ the interrogation frequenc~y duringeach successive phase-related burst thereof. Should the resulting integrated level exceed that indicative of the presence of a genuine marker t~ithin the interrogation field, an alann is sounded~ Conver5ely, if the peak-integrated signal representative o~' the low-order harmonic band becomes sufficiently large to exceed a predetermined threshold, indicating that the variable reference to ~ihich the high-order frequency i` --31- ~L~2t38~3~

samples are compared has become prohibitively large and is, in effect, blocking the detection channels, an indication of that status is given.
Initially, this indication results from energizing an indicator light and, should the condition exist for a time period exceeding that at-tributable to some unusual but nonetheless expectable occurr~nce, aflashing alarm is enabled (via detector logic channel 940).
The condition just described, indicative of an unusual and undesirable situation prevalent within the interrogation field which is causing a substantial overbalancing of the detection circuitry by way of excessive levels of the low-frequency harmonic band, is a severe aber-ration in the detection circuit parameters, and thus indicates the advisability of fully inhibiting the detection circuitry, in addition to the flashing lamp indication just noted which shows the existence of the condition. Thus, the signal indicative of the low-frequency overbalance which is coupled back to channel 940 for the purpose o~ enabling and driving the flashing lamp indicator is also coupled back, on conductor 39, to each of the control switches 10 and 12 (transistors Ql and Q2) such that ~hey latch out and bloc~ the input from terminal DL-l, thcreby preventing any build-up on integrator 34 (capacitor C9) which might otherwise result in an erroneous detection alarm.
In connection with the function and operation of detector logic channels 900 and ~30, it is to be noted that the level of the instantaneously-variable detection threshold or reference on comparators U-la and U-14a, set initially by voltage divider R14 and Rl, and varied by proportional or relative conduc~ion of transistors Q3 and Q4, is in effec~ stored for a short time interval on capacitors C7 and C8, coupled to the emitters of transistors Q4 and Q3, respectively, through a time constant-setting resistor R16 and R3~ respectively. That is, the peak levels of threshold variation due to conductance o transistors Q3 and Q4 in response to elevated inhibit signals from integrating capacitor C28, are stored on capacitors C7 and C8 between phases; thus, these threshold peaks ~ill be held briefly when the interrogation field switches its resultant fluY direction in response to reversal in the phase o~ the drive excitation applied to one of the interrogation field-inducing coils. Thus, the system will not be susceptible to error as a resul~ of detection harmonic content levels which vary substantially ~rom one ~ _,, .
. ' .

'~ -32- ~L~ 33~;

phase condition to the ne~t. What is desired is to have enough stoTage in the system so that relatively high inhibit levels built up during one phase condition, which have effectively raised the threshold at the comparators to a substantial degree, will be maintained after the change in phase condition for at least the first half-cycle of the ne~t inter-rogation field alternation and resulting detection signal, representa-tive of a change in phase condition. For example, assuming the inter-rogation field fundamental frequency to be the aforementioned 10 XHz, utilizing a bleed-off time constant on the order of about 100 msec for C capacitors C7 and C8 (the combined resistance of resistors R16 and Rl for C7 and R3 and Rl for C8), storage will be provided for an interval reasonably representative of the aforementioned period.
Of course, it is to be understood that the above is merely a description of certain preferred embodiments of the invention, and that various changes and alterations can be made without departing from the underlying concepts and broader aspects of the invention as set forth in the appended claims, which are to be interpreted in accordance with such underlying concepts and broader aspects and by application of a full range of equivalents.

Claims

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE PROPERTY OR PRIVILEGE
IS CLAIMED ARE DEFINED AS FOLLOWS:

In a method for determining the presence of a predetermined marker member within an alternating electromagnetic interrogation field, of the type using one or more monitors which produce electrical signals containing indicia indicative of the presence of such a marker within such field, the improvement comprising: subjecting the signals from different ones of said monitor means to preamplification and subtractive processing before the signal analysis which determines and indicates marker presence or absence, said preamplification comprising the separate application of the signals from different monitor means to separate preamplification channels, and said subtractive processing comprising subtracting a monitor means signal produced by a monitor means relatively further from a marker within the interrogation field from a different marker means signal produced by a monitor means which is relatively closer to that marker, thereby at least partially cancelling out ambient noise and/or other undesired effects from the monitor means signals prior to further processing thereof.

The improved method as recited in claim 1 , wherein said subtractive processing comprises subtracting from the signals produced by at least certain of said monitor means at least some of the low-order harmonics of the fundamental alternation frequency of the inter-rogation field prior to further processing of the monitor means signals.

The improved method for detection systems as recited in claim 2, comprising the steps of subtracting out low order harmonics which are produced by non-marker causation.

The improved method as recited in claim 3, further comprising the step of maintaining the presence of low-order harmonics of the fundamental field frequency resulting from marker-caused field perturbations while subtracting out said low-order harmonics produced by non-marker causation.

In a method of detecting the presence of a particular marker member within an alternating electromagnetic interrogation field established across a portal, wherein said electromagnetic field is monitored by at least first and second receiver means, each for accessing the field from a different position and for producing an electrical signal representative of signal indicia caused by such a marker in response to the alternations of the field, the improvement for use in detecting said marker which comprises the steps of: producing a first processing signal for marker-presence analysis by use of said electrical signals produced by at least one of said receivers; producing a second processing signal for use in marker-presence analysis by differencing the electrical signals produced by said first and second receivers, to reduce common-mode noise or other undesired signal characteristics in that electrical signal; and using the second processing signal in comparative analysis with said first processing electrical signal to determine presence and absence of a marker within said field.

The improvement for a marker-detection method as recited in claim 5, including the step of subjecting said second processing signal to particularly-timed blanking to remove selected signal intervals prior to said step of using said second processing signal in comparative analysis with said first processing signal.

The improvement for a marker-detection method as recited in claim 6, including the steps of periodically energizing said interrogation field by applying a drive pulse thereto, and synchronizing said blanking with said periodic drive pulse application.

The improvement for a marker-detection method as recited in claim 7, including the step of carrying out said blanking for a period longer than that of drive pulse application.

The improvement for a marker-detection method as recited in claim 6, including the step of integrating at least portions of said blanked second processing signal before carrying out said comparative analysis to determine marker presence within the interrogation field.

The improvement for a marker-detection method as recited in claim 9, including the step of restricting said integration of said second processing signal to signal excursion peak portions thereof.

The improvement for a marker-detection method as recited in claim 10, including the step of using an integration time constant which is on the order of about five times the cycle period of the fundamental frequency of alternation of said interrogation field.

The improvement for a marker-detection method as recited in claim 5, including the steps of subjecting the signals from different particular ones of said receiver means to preamplification and subtractive processing before comparative signal analysis to determine marker presence or absence, said preamplification comprising the separate application of the signals from different receiver means to separate preamplification channels, and said subtractive processing comprising subtracting a receiver means signal produced by a receiver means relatively further from a marker within the interrogation field from a different receiver means signal produced by a receiver means which is relatively closer to that marker thereby balancing the said further and closer receiver means signals with respect to one another and at least partially cancelling out ambient noise and other undesirable effects from the monitor means signals prior to further processing thereof.