CA1109516A - Direct current ballasting and starting circuitry for gaseous discharge lamps - Google Patents

Abstract

DIRECT CURRENT BALLASTING AND STARTING
CIRCUITRY FOR GASEOUS DISCHARGE LAMPS
Abstract of the Disclosure Direct current ballasting and starting circuitry for efficiently operating a gaseous discharge lamp on direct current.
Series-pass switching means in one of a pair of input lines alternatively switches between on and off states to periodically supply pulses of energy from a source of direct current voltage.
During steady-state operation, current sensing means limits the maximum current conductable by said switching means such that the output of the circuitry is current regulated. Filter means, in series with the switching means, smoothes the pulses of energy delivered by the switching means into direct current with a com-paratively small alternating current component. Starting means, in series connection between the filter means and an output termin-al, senses the nonionized state of the lamp and provides a voltage pulse of sufficient magnitude and duration to initiate ionization in the lamp. The starting means permits uninterferred passage of current from the filter means to the output terminal. Various forms of control means or drive means for controlling the con-ductive state of the switching means are disclosed, including a phase-corrected oscillator which is momentarily excited by a starting pulse delivered at the output terminal and oscillation is thereafter sustained by interaction with the switching means;
a resonant transformer capacitor circuit with a secondary winding of the transformer controlling the conductive state of the switch ing means; and a pulse width modulation drive means. Operation of the circuitry in a current-limited output mode is accomplished by either sensing current delivered by the switching means or by sensing the current level through a commutating diode.

various starting means are disclosed, including one embodiment in which a single starting means senses the nonionized state of one of a plurality of lamps, with each lamp operating from a separate ballasting circuit, and provides a starting pulse for the nonionized lamp.
An A.C. to D.C. power conversion circuit, for opera-ting a plurality of ballasting and starting circuits therefrom, utilizes a portion of a primary winding of a transformer for connection to the A.C. voltage source. A stepped up A.C. voltage across the entire primary winding is applied to a rectification means, which rectifies the A.C. voltage and supplies the same to a filter means. The filter means provides a D.C. voltage for input to the ballasting and starting circuits. A center-tap on a secondary winding of the transformer is referenced to the second of a pair of input lines of the ballasting and starting circuit, and the ends of the secondary winding provide a small A.C. voltage to the cathodic heater element of the lamp.

Description

DIRECT CURRENT BALLASTIN~ AND STARTING
CIRCUITRY FOR GASEOUS DISCHAR~E LAM~S

Background Of The Invention This invention relates in general to ballasting and starting circuitry for operating gaseous discharge lamps from D.C, and more particularly to such circuitry with a current-regulated output characteristic wherein series-pass switching means is alternatively switched between on and off conductive states and to A.C. and D.C. power conversion circuits for supplying direct current to a plurality of ballasting and starting circuits.
At the present time, use of A.C. power sources to power gaseous discharge ~ub~s,especially those used in 1uorescent lighting, by far exceed the use of D.C. power sources. This is not particularly surprising because AoC~ power sources are usually more readlly available than D.C. power sources. However, operation of fluorescent lamps from A.C. power sources has a number of dis-advantages. One of these problems is that fluorescent lamps generate and radiate radio frequency interference (RFI). RFI is a form of electro-magnetic radiation, which among other things is known for interferring with the performance of communications systems, e.g. radio and television. The RFI is generated because, as the A.C. changes or reverses polarity during a portion of each cycle, the arc between the electrodes of the gaseous discharge tube extinguishes~ The tube must then be restarted for current flow in the opposite direction and much of the RFI is generated when the arc between the electrodes of the gaseous discharge tuhe begins to restrike.
The constant polarity reversal of voltage and current in an A.C.`power source also requires that heating be provided at both electrodes of the gaseoas discharge tube. To condition a gaseous discharge tube for the striking of an arc between the electrodes, it is neces'sary to he'at the cathodic electrode to facilitate electron emission. However, in an A~C. system, the electrode of the tube'which is the' cathodic electrode is con-stantly changing as the polarity of the voltage and current change. This necessitates the heating of both terminals.
Because the arc between the terminals of the gaseous discharge tube operating from an A.C. power source is constantly being extinguished and then reignited, lighting from a fluores-cent lamp is not continuous. Instead, the lamp actually flickers.This flickering phenomenom is not noticeable to the unaided eye because the frequency of most A.C. power sources is somewhat above a frequency level which is perceptible. Nonetheless, recent behavorial and physiological studies have indicated that the inher-ent flickering has undesireable side-effects. Behavior and activity of children tending to be hyperactive are believed to be aggravated by the flickering. The flickering is also believed to hasten fatigue, which is a serious problem to medical per-sonnel when attempting to differentiate between the various shadings ' of x-ray films.
The flickering phenomenom further causes stroboscopic effects when any movement is related to a harmonic of the A.C.
power source frequency. This can present a safety hazard because - the stroboscopic effects cause rotating machinery to appear to be either stationary or slowly rotating. '`
Even when operatiny from an A.C. power source, ~luores-cent lamps require circuitry to power and control the lamp because of the unusual load characteristics of gaseous discharge tubes.
To achieve arcing between the electrodes of a gaseous discharge tube, the str1king voltage of the tube must be exceeded. The striking voltage is often twice the voltage at which the tube will '$~

operate once striking of the arc between the terminals of the tube occurs~ Circuitry must be provided to generate a voltage pulse of sufficient magnitude and duration to achieve striking. However, when striking of the arc occurs, current to the tube must then be limited. The current limiting function is often provided in A.C.
circuits by a high leakage reactance transformer, which may con-stitute the bulk of the weight and expense in a fluorescent system.
Gaseous discharge tubes are not susceptible to voltage regulation once arcing between the electrodes thereof is initiated because of the negative impedance characteristic of the tube. The tube will conduct an excessive amount of current to the point o~ sel~-destruction Therefore the current to a fluorescent lamp must be limited and the particular load charac-teristic of the lamp will determine the voltage at which it operates at a regulated current level. For a given current, the operating voltage of the tube is a function of the lenyth of the tube, its diameter, the types of gases within the tube and a number of other factors.
Some prior art efforts have been concerned with operating gaseous discharge tu~es in conjunction with D.C. ballasting and starting circuits. These efforts have been generally centered aro~md biasing a series pass semiconductor, usually a transistor, such that the current therethrough is li~ited ~o the d~ d ~urre~t through the gaseous discharge tube. TO compensate ~or a number of variables in such circuit deslgn and the expected variations in the D.C. power source, a relatively lar~e volta~e is usually drop-ped across the series pass transistor. Hence,the series-pass transistor must dissipate a significant amount of power. This power is wasted energy and Ieads to a low eficiency of operation for the circuit. The power diss1pation also requires the use of larger and more expensive semiconductors for the series pass element.

This further requires a heat sink to ~issipate the hea~ from ~he transistor, and some inskances, forced air ventilation thereof.
Thus, such prior art ballasting and starting circuits have not met with much acceptance or commercial success, except in quite limited or specialized applications.

Summary Of The Invention The direct current ballasting and starting circuitry of the present invention employs a series-pass switching means in one o a pair of input lines. The circuit provides a regulated output current. The switching means or semiconductor is alter-nately switched between on and off conductive states. During the off state, no current is conducted through the series-pass semi- `~
conductor an~ during the on stage the voltage drop thereacross is very small. The power lost in the series--pass semiconductor is very minimal. Thus there is no need for massive heat sinks or `
~or large semiconductors capable of withstanding and dissipating higher power losses, as in the prior art circuits. Means of limiting the current conducted by the switching means is accom-plished by current sensing means in series connection with switch-ing means. Filtering mean~ are in series with the current sensing means for smoothing pulses of energy delivered by the switching means ~or the D.C. voltage source. The filter means has a direct current output with a comparatively small alternating current component thereon.
~ The starting circuitry is in se~ies connection ~etween the filter means and an output terminal of the circuitry for sensing the nonionized condition of a lamp connectible to the output terminal. The starting circuitry further provides a voltage pulse of sufficient magnitude and duration to initiate ionization within the Iamp. Once ionization is achieved, the starting -, .

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circuit becomes inactive and does not impede -the supply of direct current from the filter means to the lamp.
Various means for controlling the on and off conduc-tive states of the series-pass switching semiconductor may be utilized. In one embodiment, an oscillator is momentarily res-ponsive to excitation provided by the starting voltage pulse at one of the output terminals and therea~ter interacts with the switching means. The oscillator has a resonant tank circuit for controlling the series pass semiconductor and the tank cir-cuit is also phase-corrected to provide the series pass semi-conductor with a 70 per cent duty cycle to insure that sufficient power is available ~rom the circuit.
Another embodiment of the control means utilizes a ~ree-running multivibrator; the output of which is pulse-width modulated to control the on-off states of the series-pass semi-conductor.
Where a plurality of b~llasting and startiny circuits are used, a single starting circuit may be used to generate the striking voltage pulses for all of the ballasting circuits.
Secondary windings of pulse transformers are in series connection with an output line o~ each ballasting circuit. Primary windings are connected in series between an energy storage means and a voltage responsive means. A diode from the output line of each ballasting circuit is poled in a logic "or" configuration such a nonionized condition in any lamp associate with its ballasting circuit will charge the energy storage means to a voltage level which wlll render the voltage responsive means conductive, thereby discharging the energy storage means through the primary windings of the pulse transformers and starting the desired lamp.
A power conversion circuit for converting A.C. voltage to a suitable D.C. voltage for operation of the ballasting and -5~

starting circuitry is also disclosed. A pOrtion of a primary win~ing of a transformer is tapped for applying the A.C. voltage source thereacross. The entire primary winding is applied to rectification means for rectifying the A.C. ~oltage and supply-ing the same to a filter means. The filter means supplies an elevated D.C. voltage level to a plurality of D.C. ballasting and starting circuits. A secondary winding of the transformer supplies a considerably lower A.C. voltage level -to the cathodic electrode of the lamp associated with each ballasting and start-ing circuit for heating the same. The secondary winding of thetransformer is center-tapped, with the center-tap referenced to a second of the input lines of the ballasting and starting cir-cuitry. Thus the cathode of each lamp is heated with a small A.C. voltage which is balanced with respect to the second input line thereby eliminating A.C. modulation of the D~Co current supplied to each lamp.
Various other objects, features and advantages of the invention will become apparent from the following detailed dis-closure when taken in conjunction with the drawings.

Brief Description Of The Drawings In the drawings:
Figure 1 is a schematic diagram of the A.C. to D~C~
power conversion circuit for supplying D.C. power to a plurality of ballasting and starting circuits, each of which supplies regulated direct current to a gaseous discharge tube;
Figure 2 is a schematic circuit diagram, mostly in block form, of a D.C. ballasting and starting circuit as illus-trated in Figure l;
Figure 3 is a schematic circuit diagram of the preferred embodiment of a D.C. ballasting and starting circuit for supply-ing power to a gaseous discharge tube;

Figure 4 is a schematic circuit diagram of a D.C.
ballasting circuit with an alternative embodiment oE con-trolling the conductive state of a series-pass semiconductor;
Figure 5 is a schematic cixcuit diagram of an alter-nate pulse-width modulation technique for controlling the con-ductive state of the series-pass semiconductor;
Figure 6 is a schematic circuit diagram illustrating an alternate embodiment of a starting circuit;
Figure 7 is a schematic circuit diagram illustrating use of a single starting circuit in con unction with a plurality of ballasting circuits.

Description Of The Preferred Emhodiments Referring to Figure 1, there is shown an A.C. to D.C.
power conversion circuit, generally designated as 10. A single circuit 10 is capable of supplying D.C. power of a suitable vol-tage level to a plurality of D.C. ballasting and starting circuits ll. As will be hereinafter presented, the ballasting and start-ing circuit ll supplies regulated current to at least one gaseous discharge tube or fluorescent lamp 12.
The power conversion circuit lO has a power transformer, generally designated as 13, with a core 14 of iron or other suit-able magnetic material. A primary winding 15 of the transformer 13 has a tap, at a point 16. A pair of leads ~7, 18, one of which is connected to the tap at point 16, supply voltage from an A.C. power source to a portion of the primary winding 15.
A pair of leads 19, 20 interconnect opposite ends of the primary winding 15 to opposite ~erminals 21, 24 of a rectifi-cation means, i.e. a diode rectification bridge 22. A fuse 23, in series with the lead 19, protects the circuit 10 and the cir-cuits ll from overload or malfunction which could result in ~
excessive current demand~ -To minimize the amount of filtering required to filter the rectified A.C. voltage, the diode bridge 22 is preferably of the full-wave type. The diodes in the bridge 22 are poled to provide a positive D. C. potential at a terminal 25 with respect to a common terminal 26. The potential between the te~minals 25, 26 will typically be in the range of 145 to 190 volts D.C.
Connected in parallel across a pair of leads 27, 28, which are respectively connected to the terminals 25, 26, is at least one capacitor 29 for filtering the rectified A.C. vol-tage from the bridge 22. Because of the magni~ude of D.C. voltage between the leads 27, 28, it may be more economical to provide a plurality of capacitors 29.
A bleeder resistor 30 is usually provided in parallel with the capacitors 29 to reduce the voltage across the leads 27, 28 within a specified period of time a~ter A.C. voltage has been removed from the leads 17, 18. The phosphor coating in some fluorescent lamps 12 will continue to fluoresce at a reduced level of illumination until the voltage across the lamp 12 is insufficient to maintain arcing between an anodic electrode (not shown) and cathodic electrode (not shown) of the lamp 12.
The bleeder resistor 30 insures that within a specified period of time the vol-tage levels within the circuit 10 and hence the circuit 11 will be reduced to a level at which the fluorescent phenomenom will terminate.
The positive D.C. voltage lead 27 is connected to an input line 32 o~ each o~ the ballasting and starting circuits 11.
l'he common lead 28 is similarly connected to a second input line 33 of the circuits 11~ An ou~put terminal 34 of each of the circuits 11 provides suitable power, sensing and controlling ~unctions to power and control at least one fluorescent lamp 12 of the circuits 11, as is hereinafter described.

The power transformer 13 also has a secondary ~7inding 35 with an A~C. voltage thereacross which is considerably lo~es in magnitude than that across the primary winding 15. The secondary winding 35 is center-tapped and connected by a lead 36 to the common line 28 of the circuit 10 and to the terminal 26 of the diode bridge 22. A pair of leads 38, 39 are connected to opposite ends o~ the secondary winding 35 and to terminals 40, 41 to heat the cathodic electrode within the lamp 12 to pro-vi.de electron emission therefrom. Because the secondary winding 35 is center~tapped and referenced by a lead 36 to the common line 2g, the average voltage on the cathodic electrode of any lamp 12 will be zero. Thus, ~.C. voltage modulation of the potential across any lamp 12 is minimized or eliminated. Use of a center-tapped secondary winding 35 avoids the need for supply-ing a small D.C. voltage ~or heating ~he cathodic electrode in the lamps 12, which would re~uire additional com~onents such as rectifying diodes and filtering capacitors to eliminate ripple.
The secondary winding 35, while providiny an A.C.
voltage for heating the tube 12 also ~orms part of the retun ~a path for the D.C. current for the lamps 12. D.C. current supplied by the ballasting and starting circuitry 11 at the output terminal 34 flows through the tube-12 and returns to the power conversion circuit 10 through both of the leads 38j 39, the secondary winding 35, and the lead 36 to the-terminal 26 of the diode bridge 22.
As previously noted, in operating fluorescent lamps 12 from an A.C. power source, both ends of the lamp lZ must be heated because the cathodic electrode in khe lamp 12 changes as the polarity of the voltaye and current in the lamp 12 reverse.
3Q However, in operating a lamp 12 from a D.C. power SOurGe, the polarity o the voltage and current applied to the lamp 12 ~' X ' g_ .

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emains constant. Thus, only the end of -the lamp 12 which is to be the cathodic end needs to be heated. As illustrated in Figure 1, the output terminal 34 of the ballastiny and starting circuit 11 need only be applied to one terminal on the anodic end of the lamp 12 ~o provide electrical connection there-to.
Turning now to Figure 2, there is shown a circuit dia-gram, mostL~ in block form, of one of the D.C. ballasting and starting circuits 11 of Figure 1. A pair of input lines 32, 33 supplies a source of D.C. voltage to the circui~ ll. An output terminal 34 supplies power to the gaseous discharge tube 12, and controls and senses the condition of the gaseous discharge tube.
In series with one of the input lines 32 is switching means in the form of a switching series-pass transistor 44. The switching transistor 44 is alternately switchable between on and off con-ductive states for periodically supplying pulses of energy from the sources of D.C. voltage. An oscillator circuit 45 in comhina-tion with base drive 46 provides a means of controlling the con--ductive state of switching transistor 44. Current sensing means 47 limits the maximum current through the transistor 44. One means of limiting current through the transistor 44 is by divert-ing the base drive 46 therefrom, to immediately switch the trans-istor 44 to an off conductive state. Filter means 55 includes an inductor 48 and a capacitor 49. The inductor 48 receives pulses of energy from the switching transistor 44 and in combination with ~ -the capacitor 49 filters the pulses of energy into direct current with a small alternating current component, in the form of ripple, superimpos-ed on the direct current at a junction 50. The cap-acitor 49 is connected between the junction 50 and the second input line 33. A commutating diode 51 is connected between the second input 33 and a junction 52 with the diode 51 poled to main-tain current continuity in the inductor 43 when the switching transistor 44 is in an off conductive state.
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sd/~ ~10-, A starting pulse circuit 53 is biassed between the output terminal 34 and the second input 33. Because the ballast-ing and starting circuit 11 does not regulate voltage at the output terminal 34, but only regulates current deliverable thereto, the potential at the output terminal 34 will rise to a level similar to that at the input line 32 when the circuit 11 is first energized. The starting pulse circuit 53 senses this higher voltage level as indicative of a nonionized or nonconductive condition of a lamp connectible to the ou~put terminal 34 and thereupon generates a voltage pulse of sufficient magnitude and duration to initiate ionization in a gaseous discharge lamp.
Once striking of the arc within the lamp has occurred, the vol-tage at the output terminal 34 drops and the output voltage is controlled and determined by the load characteristics of the particular gaseous discharge tube connected thereto. The circuit 11 will then begin operating in a current-regulated output mode.
As the potential at the terminal 34 drops, the startiny pulse circuit 53 will become inactive. However, should some occurrence cause a loss of arcing within the lamp, the above starting pro cess will automatically repeat.
The oscillator 45 is connected by a lead 54 to the output terminal 34 such that the starting voltage pulse initially excites the oscillator 45, and thereafter the oscillator 45 interacts with the transistor 44 to remain in an oscillatory condition. The oscillator circuit 45 alternately aids and opposes the base drive 46 to the switching s3ries-pass transistor 44, thereby alternately switching the transistor 44 between on and off conductive states.
Figure 3 illustrates the preferred embodiment of the D.C. ballastlng and starting circuit 11 of Figures 1 and 2. A
series-pass switching transiStGr 44~ of the NPN type, is connected 5~

in series between the input line 32 and a junction 56. The collector terminal of the transistor 44 is connected to the junction 56. A resistor 57 is connected from the input line 32 to a junction 58. The base terminal of the transistor 44 is also connected to the junction 58 such that the transistor receives base drive from the resistor 57 to normally bias the transistor 44 in an on conductive state.
Connected in series between the junctions 52, 56 is a resistor 59, of low ohmic value, for sensing the current delivered by the transistor 44. A series combination of a zener diode 60 and a rectifying diode 61 are connected between the junction 52 and the junction 58. The zener diode 60 has its cathode terminal connected to the junction 58 while the rectifying diode 61 has its cathode terminal at the junction 52. When the current through the current sensing resistor 59 establishes a potential thereacross which exceeds the zener voltage of the zener diode ~ .
60, the zener diode 60 begins conduction and diverts base cur- :~
rent drive delivered by the resistor 57 and by an oscilla-tor ~.
winding 86 away from the base of the transis~or 44. The diode ~
61 compensates for both the potential drop by the forward-biassed ~ase-emitter junction of the transistor 44 and also for tempera-ture variation thereof. Thus, the combination of the resistor 59, the zener diode 60 and the rectifying diode 61, limits the maximum current which the switching transistor 44 may conduct.
In fact, in steady-state operation of the circuit 11, the switch-ing transistor 44 is repeatedly switc~ed ~o the off conductive state upon reaching a predetermined current level. The ~ransistor 44 however receives sufficient hase drive that it operates near : the saturation region before delivering the limited current.
Thus, although the transistor 44 delivers significant power to the junction 56 during its on conductive states, because of the -12~
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low collector to emitter drop across the transistor 44, power dissipation in the transistor 44 is minimal. Of course, during off conductive states of the transistor 44, no current is con~
ducted therethrough and no power di~sipation therein occurs.
A series combination of another zener diode 62 and another rectifying diode 63 are connected between the junctions 56, 58 across the base-emi~ter junc~ion of the transistor 44.
The zener diode 62 has its cathode terminal connected to the emitter of the transistor 44 while the rectifying diode has its cathode terminal connected to the base of the transistor 44.
l'he diodes 62, 63 pre~ent reverse voltage breakdown of the base-emitter junction of the transistor 44 due to signals applied to the base-of the trans.istor 44 by the oscillator circuit. The zener diode 62 also reduces oscillator loading during off con-ductive states of the transistor 44.
An inductor 48 is connected in series with the current .:
sensing resistor S9 between the junctions 5~ and 50. A capa-citor 49 is connected between the junction 50 and the second .
input line 33. The combination of the series inductor 48 and the parallel capacitor 49 comprise a filter to smooth the pulses of energy delivered by the switching transistor 44 at the junction 52. The voltage at the junction 50 is primarily D.C. with a small amount of ripple superimposed th~reon~ The small voltage ripple at the point 50 is due to the fact that the filter com-prised of the inductor 48 and the capacitor 49 is not an ideal filter.
Connected between the junction 52 on the opposite side of the inductor 48 and the second input lead 33 is a commutating diode 51. The commutating diode 51 has its cathode terminal connected to the junction 52. Continuity of current through the inductor 48 during the off conductive state of the transistor 44 is provided by the commutating diode 51. However, duriny the on conductive state of the transistor 44 the commutating diode 51 is reverse-biassed and non-conductive.
Due to consideration5 of power ef~iciency and rapid switching, as are more fully discussed hereinafter, the commutat-ing diode 51 must have fast recovery times when switching between conductive states. A suitable diode is commercially available from Varo, Inc., Garland, Texas 74040, as part number V334X
and has 3 ampere, 400 volt ratings.
ld Connected in series between the junction 50 and the output terminal 34 is a secondary winding 64 of a pulse trans-former 65. The secondary winding 64 does not interfere with passage of direct current therethrough to th~ lamp 12. However, the secondary winding 64 is capable of delivering a starting pulse of sufficient magnitude and duration which, when added to the potential already at the junction 50, provides a sufficient potential at the output terminal 34 to initiate ionization and the stxiking of an arc between the electrodes o the lamp 12.
The starting circuit further has a pair of voltage dividing resistors 66 and 67 connected between the junction S0 and the second input line 33. ~nergy storage means in the form o~ a capacitor 68 is connected in parallel with the resistor 66. A
primary winding 69 o~ the pulse transformer 65 has one end con-nected to the junation 50. Another end of the primary winding 69 is connected through a spark gap 70 to another junction 71 between ~he voltage dividing resistors 66 and 67. Connected in parallel across the spark gap 10 is a second capacitor 72 which is of greater capacitance than the energy storage capacitor 6~.
The spark gap 70 is a v~ltage threshold sensitive device, which is nonconductive for voltages across the resistor 66 which are below its threshold voltage. Upon exceeding its threshold voltage, .

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the spark gap 70 assumes a very low impedance characteristic if provided with su~ficient current during initial conduction. The capacitor 72 initia].ly provides suf~icient current to insure that the spark gap 70 assumes a low impedance condition to completely discharge the energy storage capacitor 68 through the primary winding 6V of the pulse transformer 65, thereby generating a starting pulse across the secondary winding 64. After discharging the capacitor 68 to a low voltage level, the arcing in the spark gap 70 will extinguish whereupon the spark gap 70 will resume its high impedance, nonconductive sta~e. A suitable spark gap 70 with a threshold voltage level of approximately 90 volts is commercially available from the Siemens Corp, Iselin, New Jersey 08830, as part number BI-F90.
If the starting pulse generated across the secondary winding 64 is unsuccessful in striking an arc in the lamp 12, the voltage at a junction 50 will nearly equal that at the input lead 32. The energy storage capacitox 68 will again recharge in approximately one second to the point at which the voltage there-across exceeds the threshold voltage of the spark gap 70. Thus the starting circuit will continue to generate starting pulses until ionization is established in the lamp 12. Unless the lamp 12 is defective or some other circuit malfunction is present, the starting circuit will usually energize the lamp 12 when the first starting pulse is generated. When arcing in lamp 12 com-mences, the D.C. potential at the junction 50 and at the output terminal 34 will drop to a potential which is determined by the load characteristics of the lamp 12. That is, the ballasting and starting circuit 11 beyins to operate in a current-regulated out-put mode.
: 30 The means for controlling the on and off states of ~.
the series-pass transistor 44 is provided by an oscillator circuit.
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As shown in Figure 3, the oscillator circuit has a resonant tank circuit consisting of a trans~ormer 75 and a capacitor 76. A primary winding 77 of the transformer 75 is connected in series with the capacitor 76 between the junction 56 and the second .input line 33. A transistor 78 momentarily excites the resonant tank circuit when the starting pulse is generated at the output terminal 34. Thereafter, the tank circuit interacts with the transistor 44 to remain in a self-oscillating condition.
The transistor 78 has a collector terminal connected to ajunction 79 between the primary winding 77 and the capacitor 76~ An emitter terminal of the transistor 78 is referenced to the second input line 33 through a resistor 80. A diode 81 is connected between the emitter and base terminals of the transistor 78~ with the cathode of the diode 81 connected to the base of the tran-sis~or 78, to prevent reverse voltage breakdown of the base-emitter junction of the transi.stor 78 and to recharge a capacitor 83 through the resistor 80 to prepare the circuit for a subsequent starting pulse, if necessary. A series comb.ination of a resistor 82 ana a capacitor 83 are connec~ed hetween the base of the tran- ;
sistor 78 and the output terminal 34. The oscillator circuit is thus insen~itive to tke D- C. level at the output terminal 3~, but is instanteously excited when ~he starting voltage pulse appears at the output terminal 34. The transistor 78 momentarily conaucts current through winding 77 thereby generatin~ base drive for the transistor 44 across the winding 86. Resistor 80 limits peak current conducted through the winding i7. Transistor 7g only conducts during the starting pulse which, however, has the after~
effect of causing the~resonant tank circuit to ring for a suffi-cient number of cycles to cause transistor 44 to begin self-os-:cillatin~ at the natural resonant frequency of the tank circuitincluding winding 77 and capacitor 76. A parallel combination of 5~
a capacitor 84 and a resistor 85 are connec~ed in series with a secondary winding 86 of the resonant transformer 75, whieh is in turn eonneeted across the base and emitter terminals of the serles-pass switching transistor 44. The secondary winding 86 applies the oseillator output signal aeross the base-emitter ~unetion of a series-pass switching transistor 44, thereby alternately driving the transistor 44 into an on conductive state. When the level of the transistor output signal across the secondary winding 86 rises, the oseillator will reverse bias 10 the base-emitter junetion of the transistor 44 whieh will eause the transistor 44 to assume an off conductive state. At the same time, the secondary winding 86 will conduct any base drive through the resistor 57 away from the transistor 44.
A parallel combination of the capacitor 84 and the resistor 85 provide an R~C phase-shifting network in series with the secondary winding 86. The R-C network phase-eompensates the oseillator output signal for phase-shifts eaused by other eireuit eomponents, and under normal operatin~ conditions pro-duces 70 per cent on period and 30 per cent off period for the switehing transistor 44.
The resonant frequency of th~ oseillator cireuit will determine the frequency at whieh the switching resistor 44 operates. Higher operating frequencies are preferred because the induetor 48 may be of less induetanee and the capaeitor 49 of less eapaeitance and still maintain the peak~o-peak rippIe voltage appearing at the output terminal 34 below permissible levels. Lower inductive and capacitive values mean that the induetor 48 and the eapaeitor 49 of the filter means will be of smaller physieal ~size and usually of a lower cost. However, 3Q Iimitation on the maximum frequeney which the oseillator should operate is imposed by power loss eonsiderations in the switching transistor 44 and in the commutating diode 51. As previously noted, very little power dissipation occurs in the series-pass transistor 44 when the transistor 44 is in an on conductive state because it is operating in the saturation region, i,e.
a very low collector to emitter voltage drol~. No power dis-sipation occurs in the series-pass transistor 44 when it is in an off conductive state, because there is no curren~ passing therethrough. Similarly, the commutating diode 41 experiences no power dissipation when in an off conductive mode, and very little power dissipation when in an on conductive mode because the voltage drop thereacross is only that of a forward-biassed diode junction.
However, both the transistor 44 and the diode 51 have finite turn-on and turn-off time periods. When the operating frequency of the circuit becomes high enough that the turn-on and turn-off times of the transistor 44 and the diode 51 become an appreciable portion of the time period associated with the operating frequency, the power losses in both the translstor 44 and the diode 51 also become appreciable. ~ suitable operating frequency at which the size and cost of the inductor 48 and the capacitor ~9 are minimized but which is also low enough to avoid appreciable switching losses in the transistor 44 and in the diode~Sl is in the vicnityof 20 kiloHertz.
Various'other forms of control and drive means for the series-pass switching transistor 44 will become apparent to those skilled ln the art besides the oscillator circuit illustrated in Fiyure 3. Shown in Figure 4 is a resonant circuit of the tuned~T -configuratlon which is interposed between the current sensing resistor 59 and the junction 52, at which the commutating diode , 51 is connected to one side of the inductor 48. A resonant transformer 90 has a pair of primary windings 91, 92 connected in '~ ~

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, series between the current sensing resistor 59 and the junction 52. A capacitor 93 is connected to a junction 94 between the two windings 91, 92 and to the second input line 33. A second-ary winaing 95 of the transformer 90 is connected in series with a resistor 96 across the base and emitter terminals of the tran-sistor 44. The resistor 96 limits the amount of current deliver-able by the secondary winding 95 to the transistor 44. As the polarity markings associated with the windings 91, 95 would in-dicate, the secondary winding 95 provides more drive for the transistor 44 when the transistor 44 begins to assume an on con-ductive state.
At a later time during the on conductive state, when current through the primary windings 91, 92 begins to decrease , the secondary winding in 95 will experience a voltage reversal which will reverse bias the base-emitter junction of the transistor 44, thereby switching the transistor 44 to an of conductive state.
At the same time, base drive for the transistor 44 through the resistor 57 will be divertea through the secondary winding 95.
Thus, the time periods of the on and off conductive states of the series-pass transistor 44 will be determined by the frequency of the tuned resonant circuit comprising the transformer 90 and the capacitor 93. Otherwise, operation o~ the circuik in Figure 4 is similar to that in Figure 3. It is, of course, realized tha~ the circuit in Figure 4 will require a starting circuit for initiating ionization in a gaseous discharge lamp. The start-ing circuit will be connected across a junction 50 and the second input line 33 in a manner similar to the circuitry in Figure 3.
A pulse-width modulation technique for control]ing and driving the series-pass transistor 44 is illustrated in Figure 5. A zener diode 100 has an anode terminal connected to the second input line 33 and a cathode terminal connected through a resistor 101 to the first input line 32. The zener diode 100 ':

~ .

provides both reference voltages and biasing voltages a-t a terminal 102 which is connected to the cathode terminal of the zener diode 100. The terminal 102 is electrically the same as other terminals 103 and 104.
Terminal 103 is referenced to the second input line 33 through the series combination of a resistor 105, a lead 106, a resistor 107, a lead 108, and a resistor 109. A base terminal of a transistor 110, which is of the NPN type, is connected to the lead 108. An emitter terminal of the transistor 110 is referenced to the second input line 33. A collector terminal of the transistor 110 is connected to a base terminal of a PNP
transistor 111 through a resistor 112. The transistor 111 has an emitter terminal connected to the first input line 32 and a collector terminal connected to the base of the series transistor 44. A resistor 113, connected across the base~emitter junction of the transistor 111 provides a path for collector leakage cur-rents of the transistor 110. Similarly, a resistor 114 connected across the base-emitter junction of the series-pass transistor i 44 provides a path for collector leakage currents from the tran-sistor 111.
The resistors 105, 107, 109 normally bias the tran-sistor 110 into a nonconductive state which further causes the transistors 111 and 44 to also assume on conductive states.
However, shortly after the ballasting circuit in Figure 5 is turned on, a free-running multivlbrator will begin periodically switching the transistors 110, 113, 44 to an off conductive state.
An ampllfier 116 has a non-inverting input 117 connected to a reference voltage supplied by a pair of voltage divlding resistors 118j 119. The resistors 118, 119 are connected in series between the reference voltage supplied by the zener diode 100 at the terminal 102 and the second input lîne 33. A feedback resistor ~.......

120, connected between the non-inverting input 117 and an output 121 of the amplifier 116 is selected to fix the gain of the amplifier circuit. The output 121 of the amplifier 116 is con-nected to the lead 106 and is therefore capable of controlling bias voltage supplied by the resistors 105, 107, 109 to the base of the transistor 110 to control the conductive state thereof.
An inverting input 122 of the amplifier 116 is referenced to the second input line 33 through a capacitor 123. The inverting input 122 is also connected to the output 121 through the series combination diode 124 and a resistor 125, and throu~h another series combination o~ another diode 126 and another resistor 127.
The diode 124 and 126 are poled in opposite directions with the anode terminal of the diode 124 connected to the input 122 and the cathode of the diode 126 connected to the input 122.
Operation of the free-running multivibrator is as follows. When the ballasting circuit in Figure 5 is first ener-gi2ed across the input leads 32, 33, the reference voltage applied to the non-inverting input 117 by the voltage dividing resistors 118, 119 causes the output of the amplifier to assume a high output condition which enables the biasing resis~ors 105, 107, 109 to bias the transistors 110, 111, 44 into on conductive states. At the same time, the capacitor 123 connected to the inverting input 122 of the amplifier 116 begins to charge through the resistor 127 and the diode 126. The diode 124, bei.ng re-verse biased, is nonconductive. At some later time, a voltage across the capacitor 123 will begin to exceed the reference voltage at: the non-inverting input 117, thereby causing the amplifier 116 to assume a low voltage output condition at the output~121. When the output 121 of the amplifier 116 assumes a low output condition,~the transistor 110 loses its bias and assumes an off conductive state, thereby also causing transistors 3,.f~
111, 44 to assume off conductive states. The low output of the ampliier 116 also causes the voltage across-the capacitor 123 to exceed voltage at the output 121. Thus, the capacitor 123 begins discharging into the output 121 of the amplifier 116 through the diode 124 and the resistor l25. Diode 126 is now reverse-biased and therefore nonconductive. The capacitor 123 will continue to discharge until the voltage at the inverting input 122 is less than that of the non-inverting input 117, at which time, the output 121 of the amplifier 116 will again assume a high voltage condition.
It will be readily appreciated that the timing of the high and low voltage conditions at the outpu~ 121 can be con-trolled by selection of the resistors 125 and 127. For instance, if the resistive value of the resistor 127 exceeds that of the resistor 125, the capacitor 123 will take a longer time period in which to charge than the time period in which the capacitor 123 will discharge through the resistor 125. ~hus, the duty cycle, defined as the ratio of the time in which the output 121 will remain at a high voltage condition to the sum of the time periods of the high and low voltage conditions at the output 121, may be controlled depending upon the selection of the resistors 125, 127. Since the output 121 directly controls the conductive state of the series-pass switching transistor 44, the duty cycle of the multivibrator circuit directly correlates to the duty cycle of the series-pass transistor 44. To ensure r~hat sufficient energy is delivered ~o the load by the series~
pass transistor 44, a duty cycle for the multivibrator circuit of approximately 70 per cent lS preferred.
However, with a duty cycle in the vicinity of 70 per cent, the series-pass transistor 44 may deliver more energy than is desirable and therefore some means of controlling or overriding .

the output of the multivibrator circuit is desired. To this end, the resistor 129, a low ohmic value, is placed in series with the commutating diode 51 to sense the current level there-through. As previously presented, the commutating diode 51 main-tains continuity of current through the inductor 48 when the series transistor 44 assumes a nonconductive state. During steady state of operation o~ the ballasting circuit, the current of level through the inductor 48 will remain constant from cycle to cycle during any instant thereof. Thus, current through the commutating diode, while the transistor 44 is off~ is directly related to current through the transistor 44 when the transistor 44 is in a conductive sta~e. There~ore, current sensing means may be placed in series with the commutating diode 51 instead of in series with the series-pass swltching transistor 44 as is done in Figures 3 and 4.
Current threshold aetecting means is used to sense cur-rent levels in the commutating diode 51 above a predetermined threshold. A light emitting diode 130 is connected in series with an adjustable resistor 131 across the current resistor 129.
The light emitting diode 130 is placed in close proximity to a photosensitive transistor 132 such that light from the diode 130 is detected by the photosensitive transistor 132.
The electrical isolation o~ the diode 130 from the transistor 132 provides a means of voltage shifting. The collector o~ the transistor 132 is connected directly to the terminal 104 and the emltter terminal is referenced through a resistor 133 through the second input line 33. The emitter of the transistor - 132 is also connected to an inverting input 134 of a voltaye comparator 135. A non-inverting input 136 of the comparator 135 3Q lS reerenced to the second input 11ne 33 to a resistor 137 and is also referenced to the terminal 104 through another resistor ~ ' 138. The resistors 137, 138 provide a voltage reference for the non-inverting input 136. An output 139 of the comparator 135 is connected through diode 140 to the lead 108. The diode 140 is poled such that the cathode terminal thereof is connected to the output 139.
The operation o the current threshold detecting means will now be considered. Current through the current sensing means, resistor 129, and through the commutating diode 51 cause the light emitting diode 130 to emit illumination. The adjust able resistor 131 provides adjustment of the intensity of the illumination from the diode 130. Illumination on the photo-sensitive transistor 132 causes conduction of current through the transistor 132, with higher le~els of illumination causing larger currents to be conducted in the transistor 132. Because the inputs 134, 136 of the comparator 135 are both of quite high impedance, current through the transistor 132 flows through the resistor 133 thereby establishing an analog voltage at the in-verting input 134. The analog voltage is related in magnitude to the level of current through the commutating diode 51. When the current through the commutating diode 51 becomes sufficiently high, the analog voltage at the inverting input 134 will exceed the reference voltage at the non-inverting input 136 thereby causing the output 139 of voltage comparator 135 to assume a low voltage condition. ~he diode 140 which was previousl~ re-verse biassed now becomes forward biassed and diverts bias current away from the transistor 110. The current threshold detecting means will thus be able to override the output of the free-running multivibrator circuit/ thereby providing a pulse-width modulated control means for controlling the on and of conductive states of the series-pass transistor 44.

-24~

5~
It will be appreciated that the ballasting circuit in Figure 5 will use a starting circuit between the junction 50 and the second input line 33 for striking an arc in a gaseous dis-charge lamp such that the ballasting circuit of Figure 5 will then operate in its current-regulated output mode.
Turning to Figure 7, there is disclosed a schematic - --diagram wherein a single starting pulse circuit is capable of sensing a nonfunctioning lamp on any one of a plurality of bal-lasting circuits and independently starting the nonfunctioning lamp. The secondary winding 144 of a pulse transformer 145 is provided in series with the output of each of a plurality of ballasting circuits. In each of the circuits, a lead 146 con-nects one terminal of the secondary winding 144 to a junction 150 as in Figures 3, 4 and 5, while another lead 147 connects the other terminal of the secondary winding 144 to the output termi-nal 34 as in Figures 3, 4 and 5. A diode 148 is provided for each of the plurality of the circuits lI with an anode of each of the diodes 148 connected to the leads 146. The cathode termi-nals of all diodes 148 are connected together at a junction 149, such that the diodes are wired in a logic l'or" confi~uration.
An energy storage capacitor 150 is referenced through a resistor 151 to the second input line 33. From the junction 149, each primary winding~152 of the plurality of pulse transformers 145 is wired in series to a junction 153. Polarity markings must be observed as indicated in Figure 7 when wiring the primary windings 152 in series. ~Connected in parallel ~etween the junction 153 and a lead 154 are a capacitor 155, a spark gap 156, and a resistor 157. The lead I54 is also connected to a junction 158 between the capacitor 150 and the resistor 151.
When a nonenergized lamp is connected to any one of the ballasting and starting circuits, the potential on one of the leads 146 will rise toward the potential present across the , inputs 32, 33 of the ballasting circuit. The energy storage capacitor 150 will begin charging toward the potentials supplied by one of the diodes 148, as determined by the voltage dividing resistors 151, 157. The potential at the leads 146 of other ballasting circuits with energized lamps will remain unaffected because the diodes 138 associated with those circuits will be reverse biassed. When the voltage across the energy storage capacitor reaches the voltage threshold point of the spark gap 156, an arc is established in the spark gap 156 and capacitor 155 provides sufficient current when the arc is first established in the spark gap 156 to ensure that the spark ~ap 156 assumes a low impedance mode. The energy storage capacitor 150 is there-upon discharged through each of the primary windings 152 o~ the pulse transformers 1~5 through the spark gap 156. Because the ballasting circuit with the nonionized lamp will exhibit a much greater impedance than those circuits with ionized lamps, most of the energy in the capacitor 150 will be presented to the pulse transformer 145 which is associated with the nonionized lamp.
Thus, the starting circuit is capable of sensing which one of a plurality of ballasting circuits has a nonionized lamp and is further capable o~ delivering a starting pulse of sufficient magnitude an~ duration to ionize said lamp.
Another embodiment of the starting circuit 53 of Figures 2 and 3 is illustrated in Figure 6. Similar to Figures

2 and 3, a secondary winding 64 of a pulse transformer 65 is connected between the junction 50 and the outpùt terminal 34 by leads 160 and 161, respectively. A voltage dividing resistor 163 is connected between the lead 160 and a junction 164 which ; is in turn referenced to the second input lead 33 by another voItage dividing resistor 165. A capacitor 166 connected between the junction 164 and the second input lead 33 filters electrical ~26-noise at the junction 16~. A voltage level detecting means, such as a diac 167, monitors the voltage level at the junction 164, The diac 167 is connected in series with a resistor ].68 between the junction 164 and the gate terminal of an SCR 169.
The resistor 168 limits the amount of current which may be received by the gate terminal of the SCR 169 when the voltage threshold level of the diac 167 is exceeded. Another resistor 170 is connected between the gate terminal of the SCR 169 and the second input line 33 for passing any leakage currents from the diac 167.
The energy s~orage capacitor 162 is connected in series with a resistor 171 to the lead 160. One end o~ the primary winding 69 ~f the pulse transformer 65 is connected to a junction 172 between the capacitor 162 and resistor 171. The other end of the primary winding 69 is connec~ed to the anode terminal of the SCR 169.. The cathode terminal of -the SCR 169 is referenced to the second input line 33.
The line 160 will rise to a potential nearly equal to the D C. supply voltage to the inputs 32, 33 o~ the ballasti.TIg cixcuit, as previously discussed wit.h the starting circuits i.n Figures 2 and 3. When this occurs, the voltage at the junction 164 will rise slowly causing the threshcld voltage of the diac 167 to be exceeded. The diac 167 will the~eupon become conductive and discharge`the potential across the capacitor 166 into the gate terminal of the SCR 169, thereby firing the SCR 169. The SCI~ 169 . will in turn discharge the energy storage capacitor 162 -through the primary winding 69 of the pulse transformer 65, thereby generating a pulse on the secondary winding 64 of suficient magni-tude and duration to s~rike an arc in .~he lamp~ The resistor 171 is of a high resistive value such that when the SCR 169 has discharged the capacitor 162 to a low level, the resistor 171 ~27 . : ;. .. .
. . .

will not supply sufficient holding current to keep the SCR
169 in a conductive state. The SCR 169 will then cease conduc-tion and the capacitor 162 will again begin to charge toward the potential on the line 160 through the resistor 171. The starting circuit 53 thus resets itself and continues to monitor the potential at the lead 1600 Should some malfunction or other problem arise, the starting circuit 53 is prepared to restart the lamp.
For proper operation of the circuity disclosed above and in the Various drawings, selection and design of the Various magnetic components is important. For example, i:f the switching transistor 44 is to operate near the 2û kilo~ertz range, powdered ferrite cores are preferred. The magnetic components may utilize pairs of "E" cores with the legs of the pairs of "E" cores butted together. Air gaps of various widths are also employed.
For examplel the inductor 48 typically has 400 turns of #28 copper wire wound on a pair of "E" cores with a 0.020 inch air gap between the legs of the "E" cores. Suitable cores are com-mercially available from Ferroxcube Corp., Saugerties, New York 12477 as part number 782E272-3E2A.
Similarly, the transformer 75 has a primary winding 77 o:E 134 turns of #39 wire and a secondary winding of 8 turns of #29 wire with a 0. 005 inch air gap between a pair of "E"
cores of part number 206F440-3E2A (Ferroxcube). The pulse trans-formers 65, 145 have a primary winding 69~ 152 of 9 turns of #23 wire and a secondary winding 64, 144 of 200 turns o~ #26 wire with a 0.010 inch air gap between pairs of "E" cores oi~
`~ part number 782E272~3E2A (Ferroxcube).
It will be understood that various changes and modifi~

cations can be made without departing from the spirit of the invention as defined in the folIowing claims, and equivalents thereof.

Claims (17)

WE CLAIM:

1. Direct current ballasting and starting circuitry for operating a gaseous discharge lamp on direct current there-from; said circuitry comprising:
a pair of input lines for attachment to a source of direct current voltage;
a pair of output terminals connectable to said lamp;
switching means in series in one of said pair of input lines, said switching means alternatively switchable bet-ween on and off conductive states for periodically supplying pulses of energy from said source of direct current voltage;
current sensing means for limiting the maximum current conducted by said switching means curing the conductive state thereof;
filter means in series with said switching means for smoothing the pulses of energy delivered by said switching means into direct current with a comparatively small alternating current component;
starting means in series in connection between said filter means and one of said pair of output terminals, said start ing means responsive to a non-energized state of said lamp to provide a starting voltage pulse at an output terminal of a said circuitry of sufficient magnitude and duration to initiate ioniza-tion in said lamp, said starting means permitting uninterferred passage of current from said filter means to said output terminal after initiation of ionization in said lamp; and control means momentarily excited by said starting voltage pulse and thereafter interacting with said switching means to control the conductive states of said switching means.

2. The direct current ballasting and starting cir-cuitry as in Claim 1 wherein said switching means comprises a transistor with a collector terminal and an emitter terminal, said terminals connected in series in one of a pair of said input lines, and a base terminal connected to said control means.

3. The direct current ballasting and starting cir-cuitry as in Claim 2 wherein said current sensing means comprises a resistor in series with the emitter terminal of said transistor and a zener diode connected between the base terminal of the transistor and another terminal of said resistor opposite to the terminal connected to the transistor, said zener diode poled to bypass base current drive from said transistor when the sum of the potentials generated across said resistor and across a base-emitter junction of said transistor exceed the zener break-down voltage of said zener diode, thereby limiting the maximum current conductible by said transistor during the conductive states thereof.

4. The direct current ballasting and starting cir-cuitry as in Claim 3 wherein another diode is connected in series with said zener diode to temperature compensate for variations in the potential across the base-emitter junction of said tran-sistor.

5. The direct current ballasting and starting cir-cuitry as in Claim 1 wherein said current sensing means is in series connection with said switching means, and said filtering means comprises inductive means in series connection between said current sensing means and said starting means; a capacitor, with one terminal connected to said inductive means on the starting means side thereof and with another terminal connected to the second of said pair of input lines, for filtering current through said inductive means; and a commutating diode, with one terminal connected to said inductive means on the current limiting said thereof and with the other terminal connected to the second of said pair of input lines, for maintaining continuity of current of said inductive means during nonconductive states of said switching means.

6. The direct current ballasting and starting cir-cuitry as claimed in Claim 1 wherein said control means comprises an oscillator momentarily excited by the starting voltage pulse and thereafter interacting with said switching means to remain in an oscillatory condition, and an output signal of said oscillator adapted to control one of the conductive states of said switching means, said switching means operating at the natural resonant frequency of said oscillator.

7. The direct current ballasting and starting cir-cuitry as claimed in Claim 6 wherein said oscillator includes phase-shifting means for shifting phase of said output signal of said oscillator to cause said switching means to have nominal duty cycle of approximately 70 per cent.

8. The direct current ballasting and starting cir-cuitry as claimed in Claim 2 wherein said control means comprises an oscillator with a resonant tank circuit including a primary winding of a transformer and a capacitor connected in series, one end of said tank circuit connected to the emitter terminal of said transistor, another terminal of said tank circuit connected to a second of said pair of input lines; amplification means connected between the capacitor and the primary winding of said tank circuit, and between said output terminal for momentarily exciting said tank circuit with said starting voltage pulse; and a secondary winding of said transformer in series connection with a phase-shifting network across the base-emitter junction of said series-pass transistor for controlling base drive to said transistor and for resonant interaction between said transistor and said tank circuit.

9. The direct current ballasting and starting cir-cuitry as claimed in Claim 1 wherein said starting means comprises a transformer with a secondary winding thereof connected in series between said filter means and one of said pair of output terminals, a primary winding with an energy storage means at one terminal thereof, threshold sensitive means connected to another end of said primary winding and to said energy storage means to form a loop with said primary winding, said threshold sensitive means being rendered conductive when the voltage between said output terminal and the second of said input lines is indicative of a nonionized state of said lamp, said threshold sensitive means thereupon discharging said energy storage means through the pri-mary winding of said transformer to create a starting voltage pulse in said secondary winding of sufficient magnitude and duration to initiate ionization in said lamp.

10. The direct current ballasting and starting cir-cuitry as claimed in Claim 9 wherein said energy storage means comprises a first capacitor and said threshold sensitive means comprises a gaseous spark gap, with a second capacitor in parallel therewith to provide sufficient initial current through said spark gap to insure that said spark gap operates in a low impedance mode for rapidly discharging said first capacitor through said primary winding.

11. The direct current ballasting and starting cir-cuitry for operating a gaseous discharge lamp on direct current therefrom; said circuitry comprising:
a pair of input lines for attachment to a source of direct current voltage;
at least one output terminal connectable to said lamp;
switching means in series with one of said pair of input lines, said switching means alternatively switchable bet-ween on and of conductive states for periodically supplying pulses of energy from the source of direct current voltage;
current sensing means in series with said switch-ing means for limiting the current conducted by said switching means;
filter means in series with said switching means for smoothing the pulses of energy delivered by switching means to direct current with a comparatively small alternating current component;
starting means in series connection between said filter means and said output terminal, said starting means res-ponsive to a nonionized state of said lamp to provide a voltage pulse at said output terminal of said circuitry of sufficient magnitude and duration to initiate ionization in said lamp, said starting means permitting passage of current from said filter means to said output terminal after initiation of ionization in said lamp;
control means comprising a resonant circuit in-cluding a pair of primary windings of a resonant transformer in series connection between said current sensing means and said filter means, and a capacitor connected between a second of said pair of input lines and said pair of primary windings, and a secondary winding of said resonant transformer adapted to control the conductive state of said switching means.

12. The direct current ballasting and starting circuitry as claimed in Claim 11 wherein said switching means comprises of a series-pass transistor with a collector terminal and an emitter terminal connected to said control means, and said secondary winding of said resonant transformer connected in series with a resistor across the base-emitter junction of said series-pass transistor.

13. Direct current ballasting and starting circuitry for operating a gaseous discharge lamp on direct current therefrom, said circuitry comprising:
a pair of input lines for attachment to a source of direct current voltage; at least one output terminal connectable to said lamp;
switching means in series with one of said pair of input lines, said switching means alternatively switchable between on and off conductive states for periodically supplying pulses of energy from said source of direct current voltage;
inductive filter means in series with said switching means for smoothing the pulses of energy delivered by said switching means into a direct current with a comparatively small alternating current component;
starting means in series connection between said inductive filter means and said output ter-minal, said starting means responsive to a nonionized state of said lamp to provide a voltage pulse at said output terminal of sufficient magnitude and duration to initiate ionization of said lamp, said starting means permitting passage of current from said filter means to an output terminal after initiation of ionization of said lamp;
a commutating diode with one end connected between said switching means and said inductive filter means and another end connected to the second of said pair of input lines, said commutating diode being con-ductive during nonconductive states of said switching means to maintain continuity of current through said inductive filter means;
drive means for rendering said switching means periodically conductive;
current threshold detecting means for detecting whether the current through said commutating diode exceeds a predetermined threshold level, said current threshold detecting means further adapted to in-hibit said drive means as long as the current through said commutating diode exceeds said threshold level, whereby said current threshold detecting means pulse-width modulates said drive means and said switch means to regulate the current delivered by said ballasting cir-cuitry to said lamp.

14. The direct current ballasting and starting circuitry as claimed in Claim 13 wherein said drive means comprises a free-running multivibrator.

15. The direct current ballasting and starting circuitry as claimed in Claim 14 wherein said free-running multivibrator has a duty cycle of approximately 70 per cent.

16. The direct current ballasting and starting circuitry as claimed in Claim 14 wherein said current threshold detecting means comprises resistive means in series with said commutating diode to sense the current level therethrough; a reference voltage; voltage shifting means for providing a voltage analog related magnitude to the level of current through said commutating diode;
and comparator means for comparing the reference voltage to the analog voltage, said comparator means having an output adapted to inhibit said drive means whenever said analog voltage exceeds said reference voltage.

17. The direct current ballasting and starting circuitry as claimed in Claim 16 wherein said voltage shifting means comprises a light emitting diode with a portion of the potential generated across said resistive means applied thereto; and a photosensitive transistor in proximity to said light emitting diode whereby illumination from said light emitting diode regulates current conduction through said photosensitive transistor;
and a resistor in series with said photosensitive tran-sistor whereby said analog voltage is generated at a junction between said photosensitive transistor and said resistor and said analog voltage is related in magnitude to the current level through said commutating diode.