CA1077579A - Frequency discriminator - Google Patents
Frequency discriminatorInfo
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- CA1077579A CA1077579A CA236,914A CA236914A CA1077579A CA 1077579 A CA1077579 A CA 1077579A CA 236914 A CA236914 A CA 236914A CA 1077579 A CA1077579 A CA 1077579A
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- frequency
- discriminator
- output
- sweeping
- resonator
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Abstract
ABSTRACT OF THE DISCLOSURE
A frequency discriminator having a bandwidth variable from a relatively narrow range to a very wide range in which the center frequency is simultaneously tunable over an extremely wide frequency range, particularly useful for microwave appli-cations, including devices for the frequency control of signal sources, frequency measurement of signal sources, and closed-loop frequency tracking for other devices. The basic discrimin-ator element may be a ferrimagnetic resonator.
A frequency discriminator having a bandwidth variable from a relatively narrow range to a very wide range in which the center frequency is simultaneously tunable over an extremely wide frequency range, particularly useful for microwave appli-cations, including devices for the frequency control of signal sources, frequency measurement of signal sources, and closed-loop frequency tracking for other devices. The basic discrimin-ator element may be a ferrimagnetic resonator.
Description
11 ~17757~ `
This invention relates to improvements in frequency discriminators.
Center frequency tuning oiE prior art resc)nant circuit discriminatQrs consists of basically two di$ferent types: mechaniçally and electronically tuned center fre-quency types. Both types use elther the amplitude or phase ver~us requenay characteristia~ oiE a single ar dual re~onant circulk to indicate ~he Erequency oiE khe i-nput siynal relative to the discriminator center freguency. In the amplitude comparison mode, a dual mode resonant circuit is preferred with dual detectors. In the phase comparison mode, a single resonant circuit is often suiEficient with dual detectors being used to convert phase to amplitude.
The output of su~h prior art discriminators is very sensi-lS tive to khe amplltude oiE the input signal, and they re~uirecritical and expensive mean~ to lessen such amplitude dependence~
The bandwidth of the prior art discriminators is generally fixed. Therefore, in many applications r compro-mises must be made between the desired discriminator band-width linearity and resolution. A~other characteristic oiE
the prior art discriminators is their difficulty in achiev-ing a wide tuning range of the center fre~uen~y~ Th~ limi-tations of mechanically tuned circuits are due t:~ the fact that physical dimensions have to be changed and oiE~en ;; ~2 ~775'7~
multiple resonant circuits must be trackecl to maintain constant discriminator bandwidth. In the microwave fre-quency region, this limitation has been overcome to some extent by substituting ferrimagnetic cavities that can be electronically tuned over about an oc~:ave bandwidth. Thus, yittrium-iron-garnet ~YIG) discriminators have been built in the manner of Nathanson ~United States Patent No.
3,274,519) using amplitude comparison or Goodman et al, ~United States Patent No. 3,364,430), Hoover et al, (United States Patent No. 3,562,6Sl) and Pircher ~United States Pat~nt No. 3,622,896) u5ing phase characteri~tics of YIG
reson~tor.
In the speciEic prior ar~ approaches appl~ed to automatic Erequency control, a single, fixed oenter fre-quency mechanically tuned cavity has been used to stabilizethe fre~uency of high frequency generators. In one method of approach for this application, a cyclical mechanical modulation of the center frequency of the cavity has been used to sense the position of the generator frequency rela-kive to the aavity cen-tor frequency and provide a correc-tion signal to control the gene~ator. The purpo~e oE this approach was to use the superior mechanical stability o the cavity to stabilize the frequency of the generator. The cavity was tuned sinusoidally by mechanical means and the rate of tuning was limited to slow variations inherent in mechanical variations of the cavity, The application of the preC~ent invention in automatic frequency control is to vary the center frequency of the generators by tracking them to the variable center frequency of the discriminator. Bandwidth adjustments on the discriminator can be made to Eacilitate the 1~7757~
initial capture of the generator and then to maximize the frequency resolution.
According to the inventionr there is provided a frequency discriminator comprising ferrimagnetic resonator 5 means adapted to receive an input signal, magnetizing means for tuning said resonator means to a quiescent center fre-quency, sweeping means for electronically providin~ a rela--tiva frequency SWeQping between said resonator means and said input signal over a predetermined frequency range 10 about the quiescent center fre~uency of said resonator means, detector means receiving an output from said resonator means, and mean~ for aomparing ths phase of ~aid sweeping means to the output of said detector to provide an output signal .
There is thus provided a frequency discriminator whose bandwidth and center frequency can be tuned elec-tronically. The discriminator slope will be insensitive to input signal amplitude or changes in center frequency, The linearity of the discriminator will be maintained for both wide and narrow bandwidths and the center Erequency of the discriminator can be tuned over multi-octave frequency ranges, This discriminator will also be able to demodulate low .level signals by u~ing high gain amplification after crystal detection, In various applications of the frequency dis-criminator of this inven~ion, it can be used to demodulate frequency modulated voice or data communications, to control the frequency of a multiple number of o~cillators or the out-put of a harmonic generator, to overcome the sensitivity and inability to operate properly in the presence of multiple -~77579 signals of heterQdyne converters used for frequency measure-ment, and to provide a means which can be incorporated with other ferrimagnetic components to close-loop track them to a par~icular input signal.
More particularly, the frequency discriminator in-cludes an electronically tunable resonant circuit, such as a YIG rqsonator and a detector connec1ted to the resonator output. The band or spectrum of frequencies containing the input signal or signals of interest are linearly swept by the resonator circuit. The relative position of the detector output is then compared with the sweep waveform to provide th~ disariminator output. Since the relative position o~
the detectad output does not change as a ~unction of the in-put signal level, the discriminator slope is independent of signal amplitude. The sweep rate of the resonator must pro-vide at least two samples within one period of the frequency of the highest modulation component in order to accurately demodulate the information, The resonator may be used in either a band-pass or band-reject mode and several advantages accrue for each in particular applications, In the case where a ferrimagnetic resonator is used as the resonant element, the center fre-quency of the discriminator can be readily changed by con-trolling the current through an electromagnet~ The sweeping signal can then be superimposed on the center frequency tuning current to generate a sweep of the resonator, Prefer-ably, however, an auxiliary air core inductor is used to superimpose a variable magnetic field across the resonator~
By using an air core inductor, the resonator can be swept : 30 . without magnetic hysteresis or saturation and at much faster ~L~7757~
rates than would be possible through the electromagnet.
Since the ferrimagnetic resonator tunes linearly with maynetic field, a very high degree of discriminator linear-ity is achievahle by using a linear current driving source, The bandwidth of the discriminator can be controlled very accurately from a few MHz to several hundred MHz by chang-ing the magnitude of the current drive. In most of the pre-ferred embodiments disclosed herein, -the sweeping filter alternative is used; however, it is to be understood that with suitable modifications within the ordinary ski.ll in the art the ~ilter can be fixed and the requency of the ~ignal aan be linearlv swept ln like ashion creati.ng out~
puts e~uall~ suitable in several embodiments~
In order that the lnvention may be more Eully understood, it will now be described with reference to the accompanying dra~ings, in which:
Figure 1 is a cross-sectional view of an exemplary YIG resonator assembly including a magnetic housing, YIG
sphere and electromagnetic and air tuning coils for use in a fre~uency discriminator embodying the present invention;
Figure 2 is a block diayram oE a frequency dis-criminator embodying the present invention;
Figure 3 is a graphic presentation of a cyclic current waveform suitable for tuning the air coil oE Figure l;
Figures 4A - 4G relate to Figure 3 and are graphic presentation of a series of waveforms illustrating typical outputs f~om the YIG resonator of Figures 1 and 2 when the resonator is operated in a ~and-reject mode as the air coil tuning current tunes the resonator frequency f:rom FL to FH;
Figures 5A - 5D are a series o~ wave:Eorms illustrat-~7757~
ing that tho position of the peaks as in Figures 4A - 4G are independent of input signal level;
Figures 6A and 6B are block diagrams of phase com-parison methods useful in this invention when the RF detector output is used to generate a trigger at its peak response;
Figures 7A and 7B are block diagrams of phase com-parison methods useful in this invention when the analog out~
put of the RF detector is processed;
Figure 8 is a bl~ck diagram of a frequency control embodiment of the frequency discriminator according to the present invention;
Figure 9 i~ a graphical presentation of waveforms useEul in understanding the embodiment o~ Figure 8~
Figure 10 is a block dlagram of a heterodyne frequency counter embodiment of the frequency discriminator according to the present invention7 Figure 11 is a partially schematic, block diagram of a frequency modulation receiver embodiment of the frequency discriminator according to the present invention;
Figure 12 is a partially schematic block diagram of a Erequency trackin~ device embodiment of the frequency discriminator according to the present invention; and Figure 13 is a block diagram of broadband oscil-lator using the frequency discriminator according to the present invention to control the low frequency oscillator driver of a broadband harmonic generator.
Referring first to one basic preferred embodiment of the frequency discriminator according to the present in-vention as shown in Figures 1 and 2 r wherein Flgure 1 shows in cross-section a YIG filter 10 which is usable in the block ~7~
~LÇ17~75~
diagram of Figure 2. The YIG filter 10 includes a single resona~or YIG sphere 12 whose center frequency is tuned by means of an electromagnet tuning coil 14 (shown in cross-section) as a function of the electromagnet coil 14 tuning current, and a small air core coil 16 which surrounds the YIG sphere 12 in the electromagnet gap 18. As explained further hereinafter, the air core coiL 16 is driven by a periodic waveform that displaces the resonant frequency of the YIG filter cavity by an amount on either side of the quiescent frequency proportional to the current in the air core coil set by bandwidth control 19. The advantages of an air core coil (although the same effect is possible by scanning in a periodic wave~orm into the main tuning part of the electromagnet) ar~: the offset is zero, i~e., i~ there is no current in the coil 16 there is no effect on the YIG
sphere 12 resonance, the time constant is considerably smaller because the inductance of air core coil 16 is less and an air core coil has no magnetic time cons~ant~ con-sequently, the resonance can be tuned faster; and an air coil core does not exhibit magnetic hysteresis or saturation, thereby ensuring a linearity and slope independent of center ~requency tuning.
The center frequency tuning source 20 applied to the electromagnet tuning coil 14 on lines 22 and 24 sets the center frequency operation for ~he YIG filter 10. The center frequency tuning source 20 provides a current for driving the ~- electromagnet tuning coil 14. The current could be made pro-portional to frequency so that a fixed level input $ignal sets the center frequency of the discriminator to a fixed frequency or, alternately, the control current could be con-~L~7757~ :
tinually swept so that the center freguency of the dis-criminator is continually tuned. At the same time that the center frequency tuning sourae 20 is operating, the air core coil 16 is being tuned by an air coil tuning source 25 with a cyclic current as shown in Figure 3 and applied to lines 15 and 17.
The shape and generation of the driving waveforms can be of a variety of forms, They, in fact, can be shaped to provide a particularly deRirable ~ran~fer unction for the discriminator. I~, as is commonly required, the desired discriminator curve i~ linear, then the resonator tuning ~hould be linear.
~he amplitude of the tuning current ~o khe air core coil 16 whlch ~ set by the bandwidth control 19, determines th~ extent that the YIG re~onance is tuned of its quie9cent (or Fo) value. Therefore, by varying the amplitude of the drive current in 19, the width of ~he resonance sweep can be readily adjusted. As explained further below, this i~ equiv-alent to varying the bandwidth of the discriminator. Typical variations of bandwidth could be -~400 ~lHZ to as low as ~ a ~ew M~z, To this point the drive currents ~or the YIG sphere resonator 12 have been de~cribed. The drive is made up of two part~: (1) the bias or center frequency tuning to the electromagnet tuning coil 14 which controls ~hP quiescent or Fo value, (2) the cyclical delta F (~F) tuning ~rom FL to FH
the amplitude of which can be varied to var~ the bandwidth of the YIG sphere resonator 12 and there~ore the bandwidth o~ the discriminator~
Alternately, the bias tuning could be provided by a ~9~
~77579 permanent magnet, a mechanical variation of the magnetic gap or any combination of electrical and permanent magnet bias-ing. The cyclical magnetic field can be introduced at rates from DC up to several MHz.
As the YIG sphere resonator 12 is tuned across the ~F range with a controlled sweep waveform, the position in the sweep where the resonator is tunecl to a particular fre-quency must be sensed. In order to accomplish this~ the YIG
sphere resonator 12 is coupled to a transmi~sion line 21 in such a way tha~ ~he amplitude of a signal on this line within the ~F range is amplitude modulated by the resonant YIG
~phere resonator 12. The YIG sphere resonator can be aoupled in~o the tran~mi3slon line in either a band-pass or a band-re~ea~ mode. In th~ band-pass mode the RF detector 28 ls ~ coupled to the output of the transmission line 27 and as the -~, tuning of the resonator 12 approaches the ~? signal frequency the amplitude of the RF detector increases. In the band-reject mode, shown in Figure l, the RF detector 28 is coupled to the transmission line 27 and as the tuning of the resonator 12 approaches the RF signal frequency the amplitude of the RF detector 28 decreases. The choice o coupling will be dotermined by the partiaular applicationJ the prinaiple o operation is the same~
. .
Figure 4 shows waveforms illustrating typical out-puts when the YIG sphere resonator 12 is arranged in a band-reject mode. The outputs indicated occur as the input fre-quency is tuned from FL to FH.
In Figure 4A the signal frequency is just ~arely below the frequency FL~ Therefore only a small portion of the reqonance amplitude change is detected. In ]?igure 4B
- 1~7757~
the signal frequency is at the FL frequency and the ampli-tude response reaches a peak value. In Figure 4C the signal frequency falls within the ~F range and therefore it is detec-ted twice; once on the positive slope and once on the nega-S tive slope. In Figure 4D the signal frequency is at the center of the sweep, ~0. The resonance peak is e~ually spaced on both the negative and positi.ve slopes from the sweep extremes.
Figures 4E, 4F and 4G repeat the same sequence except that they recur inverse to Figures 4A, 4B and 4C.
Therefore, if only one peak occurs during the triangular sweep, the signal is at either FL or FH. Exactly where is readil~ indicated ~y determininy the time in the ~weep they occur. It i~ important to note that all of the necessary phase information is located on either the posi-tive or negative slope. The second slope contains redundant information, Since the YIG resonance is directly proportional to the changing magnetic field of the air core coil caused by the tuning current, if the current is linear the resonance point moves linearly over the entire ~F range. Therefore, the exaat position of the resonance gives an exact measure of the o~f~et of the signal frequency rom the quiescent frequency tFo) when the air core coil tuning coil current is zero.
Referring again to the block diagram of Figure 2, the RF detector output is amplified in video amplifier 29 and coupled to an amplitude detector circuit 30 which processes ths peaks or dips in t~e ~F detector 28 output amplitude.
The amplitude detector circuit 30 output is applied to a phase comparator 32 which also receives an input from the air coil ~077S7~
tuning current source 25. The output of phase comparator 32 is the frequency discriminator output. Preferred embodi-ments for processing the RF detector amplitude and comparing its phase to the air coil tuning sweep to generate the con-ventional discriminator characteristic are described herein-after.
Figure 5A indicates that the position of the peak i5 independent of the signal level~ As the amplitude of FS
increases, the output signal amplitude from RF detector 28 1~ also changes as shown in Figures 5B, 5C and 5D; however, the phase or the relative timiny of the peak of the resonance to the ~weep ~aveEorm is unchanged.
The bandwidth of the frequency discrirninator em-bodying the present invention can be reaclily varied from a few MHz to several hundred MHz~ This is accomplished by varying the amplitude of current driving the air core coil 16 of Figure 1. The lower limit is set by the loaded Q of the resonator 12: as the scan is decreased the peak of resonance is more difficult to sense. The upper limit is set by the maximum amount o current that can be forced ~hrough air core coil 16 until there is damage to the coil or ~IG sphere resonator 12 due to thermal e~fects.
The frequency discriminator embodying this inven-tion can, therefore, provide the widest possible bandwidth 2S for capturing or measuring a signal frequency and yet also provide the narrowest bandwidth for maximum resolution and highest tuning slope. Both wide and narrow band operation retain the excellent linearity charac~eristics and insensi-tivity to signal amplitude variations~
Under one principle embodimen~, operat:ion depends ~177579 on a tunable resonant cavity which is not limi~ed to a YIG
cavity, and whether a band-reject or band-pass cavity is used depends on the application -- whether a single pole or multiple pole filter is used also depends on ~he application.
The electronic control of the sweep of the air coil tuning source 25 and the capabili~y to adjust the band- ;
width of the discriminator in bandwidth control 19 of Figure 2 make possible several preferred embodiments for generating the characteristic output voltage versus input frequency curve of the discriminator, Figure 6 illustrates two methods of comparing the RF detector output to the phase of the sweeping mean~, Both depend on generati.ng a trigger pulse at the peak ~or fixed khreshold level) of the detected output signal illus~rated in Fiyures 4B through 4F, These methods are preferred for those applications in which the discriminator bandwidth is very wide with respect to the bandwidth of the resonator and/or accurate frequency informa-tion is required in a single sweep, In Figure 6A the output of the detector is coupled through a logarithmic video amplifier 34 whose output is proportional to the input power, When a band-reject resonator, as illustrated in Fiyure 1, is used to yenerate the detector output, the rejected power is independent of the absolute in-put power and the resultiny voltage waveform from the log ~5 video amplifier 34 is normalized, This simplifies the design of peak detector 36 which generates a standard output triyger at the peak (or fixed threshold level) of the detected waveform. At the same time, a sawtooth or triangular wave-form, similar to that shown in Fiy~ 3, is used to drive the resonator cyclically ~etween FL and F~ in a linear fashion.
The position of the trigger with respect to the input voltage applied to the sweeping means can then be calibrated to read out the frequency directly relative to the center frequency of the discriminator~
In Fig~ 6A the calibrated sweeping means consists of a digital clock 38 driving a reversible digital counter 40 which counts sequentially in either increasing or decreas-ing counts as controlled by a counter control 42. The : counter then programs a standard digital to analog converter 44 to generate an output voltage linearly proportioned to the nurnber of clock pulse~ counted~ The output of the D/A
converter 4~ is then Eed to khe alr coil drlver 46 sweeping the resonator llnear~y across the predetermined bandwidth~
Bandwidth control 48 is a scaling control which adjusts the current drive into the air coil to correspond to the desired discriminator bandwidth~ The input frequency to the dis-criminator is then determined by gating out the digital count in counter 40 by readout gate 50 at the time of the inter-cept trigger generated in trigger generator 52~ This pro-vides an immediate digital indication of frequency relatlve to discriminator center frequency~ Alternatively, the cali-brated analog output of the bandwidth control 48 could be gated through readout gate 54 to generate an analog measure of relative frequency. Obviously, analog outputs could also be provided with simpler voltage generated sweeping means than illustrated; the prime requirement, however, is that a linear relationship exists between the sweeping means and the position o~ the resonator in order to obtain a linear discriminator curve.
The digital or analog measures of input frequency ~14-~77579 relative to the discriminator center fre~uency represent a sampled demodulation of the input signal a~ rates equal to sweep rate of the particular sweeping means~ Standard pro-cedures for processing sampled input signals can be used to recover the input modulation.
,, Figure 6B illustrates an alternate embodiment for comparing the phase of the RF detector output to the sweep-ing means to generate the standard discriminator curve.
Again the trigger generating means of Fig. 6A is used to detect the peak output of the resonator. The sweeping wave form is generated by dividing b~ 2 the frequency of clock generator S5 in divider 56. The output pulse Erom divider 56 is integrated in ramp generator 58 to generate a linear triangular waveform similar to that shown in Fig. 3, The voltage output of this waveform is scaled in bandwidth con-trol 60 to set the desired discriminator bandwidth~ The output of control 60 is used to drive air coil driver 62 to generate a linear resonator sweep, The trigger pulse from trigger generator 52 in Figure 6A is used to control a bistable multivibrator 66 that is referenced to divider 56. This phase comparison method is similar to that described in the prior art b~
C.E. Arnold et al. (United States Patent No~ 2,764,682)~ The output of multivibrator 66 is integra~,ed in integrator 68 to provide an analog voltage proportional to the position of the trigger pulse (the detected output)relative to the driving waveform of the resonator, The output therefore measures the relative phase between the detected input signal and the sweep-ing waveform. If the sweeping waveform is linear over the predetermined bandwidth, the analog output will be a linear -15~
~L~77~79 function of frequency across the entire bandwidth of the discriminator.
In those applications where ~he bandwidth o the ,; discriminator is required to be relatively narrow with ,' S respect to the bandwidth of the resonator, or where spurious ' magnetostatic mo~es on the resonator could cause false triggeLs in the digital processing techniques, analog pro-ceSsing to determine relative position of the output with respect,to the sweeping waveform is possible. Two such approaches are shown in Fig. 7. Figure 7A is similar to the prior art approaches associated with mechanical scanning of the resonators except that the sweepiny waveorm is a linear trian~ular waveform and an attempt is made to normal-ize the logarithmic amplifier 70 at its output. 'rhe analog phase detector comprises an analog mul~iplier 72, which multiplies the output from the logarithmic amplifier 70 by the output of triangular waveform generator 71 (which also drives the bandwidth control 73 and air coil driver 75), and an integrator 74, The relative position or phase of the dete~ted output and t:he sweeping waveform is measured b~ the voltage output of integrator 74. The limitation5 of this approach are the slope dependence of the overall dis-criminator curve on the shape and output voltage level of the logarithmic amplifier.
,'; ~ In the present invention, one unique embodiment of the means for comparing the phase of the resonator sweep-ing waveform to the output of the RF detector uses a saw-tooth waveform shown in Figure 7B to drive the resonator, In this case the analog outputs from the detec:tor are all in proper phase so that a narrow band filter can be used . .
~L1377579 to extract the fundamental frequency from the waveform~ It can readily be shown that the phase of the fundamental com-ponent of the waveform is a linear function of the position of the detector outpu~ relative to the sweeping wave~orm, Figure 7B is a block diagram illustrating operation of this uni~ue phase comparison embodiment. The output of clock generator 76 is divided by 2 in digital divider 78.
The divider output triggers sawtooth g~enerator 80 putting out the voltage waveform 82 shown. The sawtooth retrace is about 10 per cent of the total period. The sawtooth voltage is scaled in bandwidth control 84 and then applied to air coil driver 86 to sweep the resonator in one direction only~
The RF detector 28 output is ampli~ied in logarithmic ampli~ie~ 88 and khe fundamental component of the detecked output is ~iltered in narrow bandwidth filter 90 tuned to the reciprocal I/T of the sweep time T. The output o~ this filter is applied to limiter 92 to eliminate amplitude varia-tions in the input, and the limited signal is applied to phase detector 94 and compared with the reference phase ~rom divider 78. Phase detector 94 provides an output voltage that i~ linearly proportional to the relative position of the detector output with the sweeping sawtooth waveorm, thereby generating the desired discriminator curve, Some significant applications for the present
This invention relates to improvements in frequency discriminators.
Center frequency tuning oiE prior art resc)nant circuit discriminatQrs consists of basically two di$ferent types: mechaniçally and electronically tuned center fre-quency types. Both types use elther the amplitude or phase ver~us requenay characteristia~ oiE a single ar dual re~onant circulk to indicate ~he Erequency oiE khe i-nput siynal relative to the discriminator center freguency. In the amplitude comparison mode, a dual mode resonant circuit is preferred with dual detectors. In the phase comparison mode, a single resonant circuit is often suiEficient with dual detectors being used to convert phase to amplitude.
The output of su~h prior art discriminators is very sensi-lS tive to khe amplltude oiE the input signal, and they re~uirecritical and expensive mean~ to lessen such amplitude dependence~
The bandwidth of the prior art discriminators is generally fixed. Therefore, in many applications r compro-mises must be made between the desired discriminator band-width linearity and resolution. A~other characteristic oiE
the prior art discriminators is their difficulty in achiev-ing a wide tuning range of the center fre~uen~y~ Th~ limi-tations of mechanically tuned circuits are due t:~ the fact that physical dimensions have to be changed and oiE~en ;; ~2 ~775'7~
multiple resonant circuits must be trackecl to maintain constant discriminator bandwidth. In the microwave fre-quency region, this limitation has been overcome to some extent by substituting ferrimagnetic cavities that can be electronically tuned over about an oc~:ave bandwidth. Thus, yittrium-iron-garnet ~YIG) discriminators have been built in the manner of Nathanson ~United States Patent No.
3,274,519) using amplitude comparison or Goodman et al, ~United States Patent No. 3,364,430), Hoover et al, (United States Patent No. 3,562,6Sl) and Pircher ~United States Pat~nt No. 3,622,896) u5ing phase characteri~tics of YIG
reson~tor.
In the speciEic prior ar~ approaches appl~ed to automatic Erequency control, a single, fixed oenter fre-quency mechanically tuned cavity has been used to stabilizethe fre~uency of high frequency generators. In one method of approach for this application, a cyclical mechanical modulation of the center frequency of the cavity has been used to sense the position of the generator frequency rela-kive to the aavity cen-tor frequency and provide a correc-tion signal to control the gene~ator. The purpo~e oE this approach was to use the superior mechanical stability o the cavity to stabilize the frequency of the generator. The cavity was tuned sinusoidally by mechanical means and the rate of tuning was limited to slow variations inherent in mechanical variations of the cavity, The application of the preC~ent invention in automatic frequency control is to vary the center frequency of the generators by tracking them to the variable center frequency of the discriminator. Bandwidth adjustments on the discriminator can be made to Eacilitate the 1~7757~
initial capture of the generator and then to maximize the frequency resolution.
According to the inventionr there is provided a frequency discriminator comprising ferrimagnetic resonator 5 means adapted to receive an input signal, magnetizing means for tuning said resonator means to a quiescent center fre-quency, sweeping means for electronically providin~ a rela--tiva frequency SWeQping between said resonator means and said input signal over a predetermined frequency range 10 about the quiescent center fre~uency of said resonator means, detector means receiving an output from said resonator means, and mean~ for aomparing ths phase of ~aid sweeping means to the output of said detector to provide an output signal .
There is thus provided a frequency discriminator whose bandwidth and center frequency can be tuned elec-tronically. The discriminator slope will be insensitive to input signal amplitude or changes in center frequency, The linearity of the discriminator will be maintained for both wide and narrow bandwidths and the center Erequency of the discriminator can be tuned over multi-octave frequency ranges, This discriminator will also be able to demodulate low .level signals by u~ing high gain amplification after crystal detection, In various applications of the frequency dis-criminator of this inven~ion, it can be used to demodulate frequency modulated voice or data communications, to control the frequency of a multiple number of o~cillators or the out-put of a harmonic generator, to overcome the sensitivity and inability to operate properly in the presence of multiple -~77579 signals of heterQdyne converters used for frequency measure-ment, and to provide a means which can be incorporated with other ferrimagnetic components to close-loop track them to a par~icular input signal.
More particularly, the frequency discriminator in-cludes an electronically tunable resonant circuit, such as a YIG rqsonator and a detector connec1ted to the resonator output. The band or spectrum of frequencies containing the input signal or signals of interest are linearly swept by the resonator circuit. The relative position of the detector output is then compared with the sweep waveform to provide th~ disariminator output. Since the relative position o~
the detectad output does not change as a ~unction of the in-put signal level, the discriminator slope is independent of signal amplitude. The sweep rate of the resonator must pro-vide at least two samples within one period of the frequency of the highest modulation component in order to accurately demodulate the information, The resonator may be used in either a band-pass or band-reject mode and several advantages accrue for each in particular applications, In the case where a ferrimagnetic resonator is used as the resonant element, the center fre-quency of the discriminator can be readily changed by con-trolling the current through an electromagnet~ The sweeping signal can then be superimposed on the center frequency tuning current to generate a sweep of the resonator, Prefer-ably, however, an auxiliary air core inductor is used to superimpose a variable magnetic field across the resonator~
By using an air core inductor, the resonator can be swept : 30 . without magnetic hysteresis or saturation and at much faster ~L~7757~
rates than would be possible through the electromagnet.
Since the ferrimagnetic resonator tunes linearly with maynetic field, a very high degree of discriminator linear-ity is achievahle by using a linear current driving source, The bandwidth of the discriminator can be controlled very accurately from a few MHz to several hundred MHz by chang-ing the magnitude of the current drive. In most of the pre-ferred embodiments disclosed herein, -the sweeping filter alternative is used; however, it is to be understood that with suitable modifications within the ordinary ski.ll in the art the ~ilter can be fixed and the requency of the ~ignal aan be linearlv swept ln like ashion creati.ng out~
puts e~uall~ suitable in several embodiments~
In order that the lnvention may be more Eully understood, it will now be described with reference to the accompanying dra~ings, in which:
Figure 1 is a cross-sectional view of an exemplary YIG resonator assembly including a magnetic housing, YIG
sphere and electromagnetic and air tuning coils for use in a fre~uency discriminator embodying the present invention;
Figure 2 is a block diayram oE a frequency dis-criminator embodying the present invention;
Figure 3 is a graphic presentation of a cyclic current waveform suitable for tuning the air coil oE Figure l;
Figures 4A - 4G relate to Figure 3 and are graphic presentation of a series of waveforms illustrating typical outputs f~om the YIG resonator of Figures 1 and 2 when the resonator is operated in a ~and-reject mode as the air coil tuning current tunes the resonator frequency f:rom FL to FH;
Figures 5A - 5D are a series o~ wave:Eorms illustrat-~7757~
ing that tho position of the peaks as in Figures 4A - 4G are independent of input signal level;
Figures 6A and 6B are block diagrams of phase com-parison methods useful in this invention when the RF detector output is used to generate a trigger at its peak response;
Figures 7A and 7B are block diagrams of phase com-parison methods useful in this invention when the analog out~
put of the RF detector is processed;
Figure 8 is a bl~ck diagram of a frequency control embodiment of the frequency discriminator according to the present invention;
Figure 9 i~ a graphical presentation of waveforms useEul in understanding the embodiment o~ Figure 8~
Figure 10 is a block dlagram of a heterodyne frequency counter embodiment of the frequency discriminator according to the present invention7 Figure 11 is a partially schematic, block diagram of a frequency modulation receiver embodiment of the frequency discriminator according to the present invention;
Figure 12 is a partially schematic block diagram of a Erequency trackin~ device embodiment of the frequency discriminator according to the present invention; and Figure 13 is a block diagram of broadband oscil-lator using the frequency discriminator according to the present invention to control the low frequency oscillator driver of a broadband harmonic generator.
Referring first to one basic preferred embodiment of the frequency discriminator according to the present in-vention as shown in Figures 1 and 2 r wherein Flgure 1 shows in cross-section a YIG filter 10 which is usable in the block ~7~
~LÇ17~75~
diagram of Figure 2. The YIG filter 10 includes a single resona~or YIG sphere 12 whose center frequency is tuned by means of an electromagnet tuning coil 14 (shown in cross-section) as a function of the electromagnet coil 14 tuning current, and a small air core coil 16 which surrounds the YIG sphere 12 in the electromagnet gap 18. As explained further hereinafter, the air core coiL 16 is driven by a periodic waveform that displaces the resonant frequency of the YIG filter cavity by an amount on either side of the quiescent frequency proportional to the current in the air core coil set by bandwidth control 19. The advantages of an air core coil (although the same effect is possible by scanning in a periodic wave~orm into the main tuning part of the electromagnet) ar~: the offset is zero, i~e., i~ there is no current in the coil 16 there is no effect on the YIG
sphere 12 resonance, the time constant is considerably smaller because the inductance of air core coil 16 is less and an air core coil has no magnetic time cons~ant~ con-sequently, the resonance can be tuned faster; and an air coil core does not exhibit magnetic hysteresis or saturation, thereby ensuring a linearity and slope independent of center ~requency tuning.
The center frequency tuning source 20 applied to the electromagnet tuning coil 14 on lines 22 and 24 sets the center frequency operation for ~he YIG filter 10. The center frequency tuning source 20 provides a current for driving the ~- electromagnet tuning coil 14. The current could be made pro-portional to frequency so that a fixed level input $ignal sets the center frequency of the discriminator to a fixed frequency or, alternately, the control current could be con-~L~7757~ :
tinually swept so that the center freguency of the dis-criminator is continually tuned. At the same time that the center frequency tuning sourae 20 is operating, the air core coil 16 is being tuned by an air coil tuning source 25 with a cyclic current as shown in Figure 3 and applied to lines 15 and 17.
The shape and generation of the driving waveforms can be of a variety of forms, They, in fact, can be shaped to provide a particularly deRirable ~ran~fer unction for the discriminator. I~, as is commonly required, the desired discriminator curve i~ linear, then the resonator tuning ~hould be linear.
~he amplitude of the tuning current ~o khe air core coil 16 whlch ~ set by the bandwidth control 19, determines th~ extent that the YIG re~onance is tuned of its quie9cent (or Fo) value. Therefore, by varying the amplitude of the drive current in 19, the width of ~he resonance sweep can be readily adjusted. As explained further below, this i~ equiv-alent to varying the bandwidth of the discriminator. Typical variations of bandwidth could be -~400 ~lHZ to as low as ~ a ~ew M~z, To this point the drive currents ~or the YIG sphere resonator 12 have been de~cribed. The drive is made up of two part~: (1) the bias or center frequency tuning to the electromagnet tuning coil 14 which controls ~hP quiescent or Fo value, (2) the cyclical delta F (~F) tuning ~rom FL to FH
the amplitude of which can be varied to var~ the bandwidth of the YIG sphere resonator 12 and there~ore the bandwidth o~ the discriminator~
Alternately, the bias tuning could be provided by a ~9~
~77579 permanent magnet, a mechanical variation of the magnetic gap or any combination of electrical and permanent magnet bias-ing. The cyclical magnetic field can be introduced at rates from DC up to several MHz.
As the YIG sphere resonator 12 is tuned across the ~F range with a controlled sweep waveform, the position in the sweep where the resonator is tunecl to a particular fre-quency must be sensed. In order to accomplish this~ the YIG
sphere resonator 12 is coupled to a transmi~sion line 21 in such a way tha~ ~he amplitude of a signal on this line within the ~F range is amplitude modulated by the resonant YIG
~phere resonator 12. The YIG sphere resonator can be aoupled in~o the tran~mi3slon line in either a band-pass or a band-re~ea~ mode. In th~ band-pass mode the RF detector 28 ls ~ coupled to the output of the transmission line 27 and as the -~, tuning of the resonator 12 approaches the ~? signal frequency the amplitude of the RF detector increases. In the band-reject mode, shown in Figure l, the RF detector 28 is coupled to the transmission line 27 and as the tuning of the resonator 12 approaches the RF signal frequency the amplitude of the RF detector 28 decreases. The choice o coupling will be dotermined by the partiaular applicationJ the prinaiple o operation is the same~
. .
Figure 4 shows waveforms illustrating typical out-puts when the YIG sphere resonator 12 is arranged in a band-reject mode. The outputs indicated occur as the input fre-quency is tuned from FL to FH.
In Figure 4A the signal frequency is just ~arely below the frequency FL~ Therefore only a small portion of the reqonance amplitude change is detected. In ]?igure 4B
- 1~7757~
the signal frequency is at the FL frequency and the ampli-tude response reaches a peak value. In Figure 4C the signal frequency falls within the ~F range and therefore it is detec-ted twice; once on the positive slope and once on the nega-S tive slope. In Figure 4D the signal frequency is at the center of the sweep, ~0. The resonance peak is e~ually spaced on both the negative and positi.ve slopes from the sweep extremes.
Figures 4E, 4F and 4G repeat the same sequence except that they recur inverse to Figures 4A, 4B and 4C.
Therefore, if only one peak occurs during the triangular sweep, the signal is at either FL or FH. Exactly where is readil~ indicated ~y determininy the time in the ~weep they occur. It i~ important to note that all of the necessary phase information is located on either the posi-tive or negative slope. The second slope contains redundant information, Since the YIG resonance is directly proportional to the changing magnetic field of the air core coil caused by the tuning current, if the current is linear the resonance point moves linearly over the entire ~F range. Therefore, the exaat position of the resonance gives an exact measure of the o~f~et of the signal frequency rom the quiescent frequency tFo) when the air core coil tuning coil current is zero.
Referring again to the block diagram of Figure 2, the RF detector output is amplified in video amplifier 29 and coupled to an amplitude detector circuit 30 which processes ths peaks or dips in t~e ~F detector 28 output amplitude.
The amplitude detector circuit 30 output is applied to a phase comparator 32 which also receives an input from the air coil ~077S7~
tuning current source 25. The output of phase comparator 32 is the frequency discriminator output. Preferred embodi-ments for processing the RF detector amplitude and comparing its phase to the air coil tuning sweep to generate the con-ventional discriminator characteristic are described herein-after.
Figure 5A indicates that the position of the peak i5 independent of the signal level~ As the amplitude of FS
increases, the output signal amplitude from RF detector 28 1~ also changes as shown in Figures 5B, 5C and 5D; however, the phase or the relative timiny of the peak of the resonance to the ~weep ~aveEorm is unchanged.
The bandwidth of the frequency discrirninator em-bodying the present invention can be reaclily varied from a few MHz to several hundred MHz~ This is accomplished by varying the amplitude of current driving the air core coil 16 of Figure 1. The lower limit is set by the loaded Q of the resonator 12: as the scan is decreased the peak of resonance is more difficult to sense. The upper limit is set by the maximum amount o current that can be forced ~hrough air core coil 16 until there is damage to the coil or ~IG sphere resonator 12 due to thermal e~fects.
The frequency discriminator embodying this inven-tion can, therefore, provide the widest possible bandwidth 2S for capturing or measuring a signal frequency and yet also provide the narrowest bandwidth for maximum resolution and highest tuning slope. Both wide and narrow band operation retain the excellent linearity charac~eristics and insensi-tivity to signal amplitude variations~
Under one principle embodimen~, operat:ion depends ~177579 on a tunable resonant cavity which is not limi~ed to a YIG
cavity, and whether a band-reject or band-pass cavity is used depends on the application -- whether a single pole or multiple pole filter is used also depends on ~he application.
The electronic control of the sweep of the air coil tuning source 25 and the capabili~y to adjust the band- ;
width of the discriminator in bandwidth control 19 of Figure 2 make possible several preferred embodiments for generating the characteristic output voltage versus input frequency curve of the discriminator, Figure 6 illustrates two methods of comparing the RF detector output to the phase of the sweeping mean~, Both depend on generati.ng a trigger pulse at the peak ~or fixed khreshold level) of the detected output signal illus~rated in Fiyures 4B through 4F, These methods are preferred for those applications in which the discriminator bandwidth is very wide with respect to the bandwidth of the resonator and/or accurate frequency informa-tion is required in a single sweep, In Figure 6A the output of the detector is coupled through a logarithmic video amplifier 34 whose output is proportional to the input power, When a band-reject resonator, as illustrated in Fiyure 1, is used to yenerate the detector output, the rejected power is independent of the absolute in-put power and the resultiny voltage waveform from the log ~5 video amplifier 34 is normalized, This simplifies the design of peak detector 36 which generates a standard output triyger at the peak (or fixed threshold level) of the detected waveform. At the same time, a sawtooth or triangular wave-form, similar to that shown in Fiy~ 3, is used to drive the resonator cyclically ~etween FL and F~ in a linear fashion.
The position of the trigger with respect to the input voltage applied to the sweeping means can then be calibrated to read out the frequency directly relative to the center frequency of the discriminator~
In Fig~ 6A the calibrated sweeping means consists of a digital clock 38 driving a reversible digital counter 40 which counts sequentially in either increasing or decreas-ing counts as controlled by a counter control 42. The : counter then programs a standard digital to analog converter 44 to generate an output voltage linearly proportioned to the nurnber of clock pulse~ counted~ The output of the D/A
converter 4~ is then Eed to khe alr coil drlver 46 sweeping the resonator llnear~y across the predetermined bandwidth~
Bandwidth control 48 is a scaling control which adjusts the current drive into the air coil to correspond to the desired discriminator bandwidth~ The input frequency to the dis-criminator is then determined by gating out the digital count in counter 40 by readout gate 50 at the time of the inter-cept trigger generated in trigger generator 52~ This pro-vides an immediate digital indication of frequency relatlve to discriminator center frequency~ Alternatively, the cali-brated analog output of the bandwidth control 48 could be gated through readout gate 54 to generate an analog measure of relative frequency. Obviously, analog outputs could also be provided with simpler voltage generated sweeping means than illustrated; the prime requirement, however, is that a linear relationship exists between the sweeping means and the position o~ the resonator in order to obtain a linear discriminator curve.
The digital or analog measures of input frequency ~14-~77579 relative to the discriminator center fre~uency represent a sampled demodulation of the input signal a~ rates equal to sweep rate of the particular sweeping means~ Standard pro-cedures for processing sampled input signals can be used to recover the input modulation.
,, Figure 6B illustrates an alternate embodiment for comparing the phase of the RF detector output to the sweep-ing means to generate the standard discriminator curve.
Again the trigger generating means of Fig. 6A is used to detect the peak output of the resonator. The sweeping wave form is generated by dividing b~ 2 the frequency of clock generator S5 in divider 56. The output pulse Erom divider 56 is integrated in ramp generator 58 to generate a linear triangular waveform similar to that shown in Fig. 3, The voltage output of this waveform is scaled in bandwidth con-trol 60 to set the desired discriminator bandwidth~ The output of control 60 is used to drive air coil driver 62 to generate a linear resonator sweep, The trigger pulse from trigger generator 52 in Figure 6A is used to control a bistable multivibrator 66 that is referenced to divider 56. This phase comparison method is similar to that described in the prior art b~
C.E. Arnold et al. (United States Patent No~ 2,764,682)~ The output of multivibrator 66 is integra~,ed in integrator 68 to provide an analog voltage proportional to the position of the trigger pulse (the detected output)relative to the driving waveform of the resonator, The output therefore measures the relative phase between the detected input signal and the sweep-ing waveform. If the sweeping waveform is linear over the predetermined bandwidth, the analog output will be a linear -15~
~L~77~79 function of frequency across the entire bandwidth of the discriminator.
In those applications where ~he bandwidth o the ,; discriminator is required to be relatively narrow with ,' S respect to the bandwidth of the resonator, or where spurious ' magnetostatic mo~es on the resonator could cause false triggeLs in the digital processing techniques, analog pro-ceSsing to determine relative position of the output with respect,to the sweeping waveform is possible. Two such approaches are shown in Fig. 7. Figure 7A is similar to the prior art approaches associated with mechanical scanning of the resonators except that the sweepiny waveorm is a linear trian~ular waveform and an attempt is made to normal-ize the logarithmic amplifier 70 at its output. 'rhe analog phase detector comprises an analog mul~iplier 72, which multiplies the output from the logarithmic amplifier 70 by the output of triangular waveform generator 71 (which also drives the bandwidth control 73 and air coil driver 75), and an integrator 74, The relative position or phase of the dete~ted output and t:he sweeping waveform is measured b~ the voltage output of integrator 74. The limitation5 of this approach are the slope dependence of the overall dis-criminator curve on the shape and output voltage level of the logarithmic amplifier.
,'; ~ In the present invention, one unique embodiment of the means for comparing the phase of the resonator sweep-ing waveform to the output of the RF detector uses a saw-tooth waveform shown in Figure 7B to drive the resonator, In this case the analog outputs from the detec:tor are all in proper phase so that a narrow band filter can be used . .
~L1377579 to extract the fundamental frequency from the waveform~ It can readily be shown that the phase of the fundamental com-ponent of the waveform is a linear function of the position of the detector outpu~ relative to the sweeping wave~orm, Figure 7B is a block diagram illustrating operation of this uni~ue phase comparison embodiment. The output of clock generator 76 is divided by 2 in digital divider 78.
The divider output triggers sawtooth g~enerator 80 putting out the voltage waveform 82 shown. The sawtooth retrace is about 10 per cent of the total period. The sawtooth voltage is scaled in bandwidth control 84 and then applied to air coil driver 86 to sweep the resonator in one direction only~
The RF detector 28 output is ampli~ied in logarithmic ampli~ie~ 88 and khe fundamental component of the detecked output is ~iltered in narrow bandwidth filter 90 tuned to the reciprocal I/T of the sweep time T. The output o~ this filter is applied to limiter 92 to eliminate amplitude varia-tions in the input, and the limited signal is applied to phase detector 94 and compared with the reference phase ~rom divider 78. Phase detector 94 provides an output voltage that i~ linearly proportional to the relative position of the detector output with the sweeping sawtooth waveorm, thereby generating the desired discriminator curve, Some significant applications for the present
2~ discriminator include:
1. Frequency control of signal sources;
2. Frequency measurement of signal sources;
1. Frequency control of signal sources;
2. Frequency measurement of signal sources;
3. Demodulation of FM signals~ and
4. Tracking control for other YIG devices.
Each of these will be described along with exemplary ~77579 preerred embodiments of the invention.
Freque~y Control of S~nal Sources Solid state signal sources are available commercial-ly to cover the frequency range from 500 MHz through 18,000 MHz in a single unit. These sources generally use fundamental oscillators, which are electronically tuned over octave ranges by either varactor or YIG devices. Each octave range oscil-lator then requires individual calibration and control to provide the necessary frequency stability and accuracy. ~he net result is a system that is extremely expensive,.
The application of the wide bandwidth tunable dis-criminator in source control is to provide a single device which fixes the accuraay, stability, linearity and reset-ability of frequency, reduces frequency pushincJ and pulling;
eliminates turn-on drift; and provides continuous tuning at band cross-over frequencies.
Referring now to Figures 8 and 9, a frequency con-trol embodiment including the wide bandwidth frequency dis-criminator of Fig~ 2 is shown. Assume that the desired opera-tion consists of contiguous scan of RF oscillators 98, 100 and 102, and the RF coupler 103 sample~ the oscillator output to provide input ~or discriminator 108.
Initially, the center frequency control 104 is tuned to the lowest frequency of oscillator 98 and a coarse frequency tune is applied to oscillator 98 vla summing ampliier 106. The wide bandwidth discriminator 108 is centered at the correct Fo by tuning unit 104 (see Figure 9), Next the wide bandwidth position (~F max~ of ~igure 9~ is switched into operation ~y bandwidth adjust con-trol 110 and the bias voltage B~ is applied to oscillator 98 ~L077579 by bias control 99, In the ~F max~ position, the oscillator is quickly tuned near Fo by the closed loop feedback control network 101. If necessary, the gain of the loop can then be increased by switching in the ~F min. bandwidth on the dis-criminator. Now as Fo is tuned by cen~er frequency control 104, the oscillator will be forced to track Fo by the feed-back control action of the discriminator 108.
When the Fo control reaches the highest frequency in c)scillator 98, then oscillator 98 is turned off and oscillator 100 is turned on. The discriminator bandwidth i5 again switched to aF max. to capture oscillator 100, whlch is also forced to track the Fo of the discriminator.
Note that the Fo tuning is continuous and there is only a transient as one oscillator i9 turned of while khe other is turned on. The discriminator frequency reerence can be stopped momentarily at this time to allow any transients to die down before continuing the scan~
It is not necessar~ to turn the oscillator off and on i there is sufficient isolation in the RF switch 112 to meet the spuriouq signal requirements. However, high isola-tion fast RF qwitches are expensive and this offers an economical approach with significantly improved isolation over the techniques that are commercially available, Control of oscillator 102 is passed on from oscillator 100 in much the same way as from oscillator 98 to 100. At the end of the tuning range of oscillator 102 the sequence can be repeated. The sweep rate is limited by the scanning capability of the discriminator 108.
Frequency Measurement of Signal Sources Another useful capability of the disc:riminator is : . . .
~1~7757~
to augment the operation of an automatic heterodyne counter by determining the exact frequency offset of ~he unknown in-put frequency relative to a harmonic comb line. In this manner, the IF bandwidth of the amplifier following the RF
converqion process can be reduced, This signiicantly avoids two of the principle limitations of prior art heterodyne counters: the susceptibility to interference in the presence of multiplo signals and the difficulty in measuring low level signals due to the wide noise bandwidths of the IF amplifiers currently in use. It also allows for the u~e of harmonic comb line spacing much wider than is currently possible, e.g., 500 MHz versus 100 to 200 MHz, A wider comb line spaaing inareAses ~he ~F comb power available rom the harmonic generator at requencies above 12 GHz and provide~ better i~olation between adjacent com~ line. This further improves sensitivity and extends frequency measurement to much higher requencies (e.g., to 40 GHz). At the same time, the presence o an input signal and the selection of the proper harmonic for converting it to IF can be provided by the dis-criminator directly rom the RF input signal rather than a~ter~F proce~sing as i9 currently re~uired. A block diagram o an embodiment of the invention for improving automatic heterodyne counting is ~hown in Fig. 10.
The RF input signal at terminal 112 is applied through a 3db at~enuator 113 to a band-reject type YIG filter 114 which is included in the same magnetic housing as YIG
tuned harmonic generator 116. Harmonic generator 116 is driven from a ixed IF oscillator 118 which is phase locked to the ba~ic digital counter 120 as is common in the prior art. Power control }22 provides a means to turn oEf or reduce ~O--1C97~579 the input drive to the harmonic generator. Digital-to-analog converter 128 in response to drlve control 124 provides the tuning control for electromagnet 126.
Initially, the IF oscillator 118 input drive to the harmonic generator 116 is turned off and D/A converter 128 continually step tunes YIG filter 114 over the desired operating range ~e.g., 018 to 26 GHz) in steps coarsely equal to the comb line spacing (equal to the frequency of IF
oscillator 118). In this mode (Mode A), YIG ilter 114 operates in conjunction with RF detector 127, video amplifier 130, phase comparison means 132 and sweeping means 134 to form the discriminator of Figure 2. Sweeping mean~ 134 cyclically scans YIG filter 11~ over a ranye equal to or greAter than the comb line spacincJ. When the dlsarlminator intercepts an inpu~ signal, an output similar to Figure 3 is generated. The novel ability of this discriminator to select the largest signal will eliminate false-locking pro-blems.
When a signal is detected in the discriminator, D/A drive control 124 stops the scanning of electromagnet 126. A frequency measurement with the discriminator of the offset of the signal with respect to the nominal comb line frequency is used to select the best comb line for hetero-dyne conversion. The comb line number is preset in digital counter 120.
At this point, power control 122 turns on the IF
; drive to YIG tuned harmonic generator 115. Note, however, that electromagnet 126 has not necessarily tuned the YIG
harmonic generator exactly on to the desired comb line.
ThiS is accomplished ~y using a combination of the .
~77S~
YIG tuned harmonic generator 116 (now in a band-pass embodi-menk), RF detector 127, video amplifier 130, phase comparison means 132 and sweeping means 134 to form the discriminator of Figure 2. In this mode ~Mode B), RF switch 136 is opened to isolate the input signal from RF detector 127 and RF
coupler 138 applies the RF output from ~he harmonic generator 116 to RF detector 127. The output from pha~e compari~on means 132 is coupled back to D/A drive control 124 through feedback network 137. This fine control on the electro-magnet current centers the quiescent tuning of the housingat the exact comb line ~requency.
Therefore, the harmonic comb line that: will hetero~
dyne the lnput signal frequency lnto digital counter 120 has been selected and its output power optimized, At this time, operation reverts back to Mode A where the input signal off-set is now measured by the discriminator with respect to the updated tuning of the housing (now centered on the comb line frequency). Filter/ampliier natwork 140 is then tuned tto the offset frequency measured in network 141 by the discrim-- !
inator. The bandwidth of network 140 can be considerably less than the IF comb spacing thereby signi~icantly improving the capability of the automatic heterodyne counter to measure low level input signals and to operate in the presence o multiple inputs. The IF oscillator 118 providing an input drive to harmonic generator 116 is then turned on, and RF
detector 127 heterodynes the input signal and the selected comb line in the conventional prior art manner. Periodically, prefera~ly during the reset period of digital counter 120, the discriminator reverts into Mode A assuring that: network 0 140 is tuned to the proper IF offset and that the YIG tuned 16)775'79 harmonic generator 116 is tuned to the optimum comb line~
Demodulation of FM~ nals This technique provides a discriminator that makes possible practical low cost FM communications systems at microwave frequencies.
Referring to Figure 11 whereln an exemplary embodi-ment of an FM receiver having a frequency discriminator according to the invention is shown, the FM mo~ulated input signal is inaident upon a microwave antenna system 142. The signal is routed directly to RF detector 144 through tran~-mission line 146. The air coil drive 148 is set for maximum ~F to ensure capturing the signal, The offset frequency of Fo relative to the permanent magnet 149 bia~ field is measured in phase detector 150 and a DC current to air coil drlve 1~8 lS aenters the di~criminator (provides automatic frequency con-trol). Next, the ~F range is optimized to match the input signal deviation. Video amplifier 152 amplifies the signal generated as YIG sphere resonator 154 sweeps through the RF
input signal (either band-pass or band-reject can be used) by means of air core coil 155. Phase detector 150 basically compares the RF detector 144 output position to the sweep waveform creating a conventional FM discriminator curve using methods such as depicted in Figures 6 and 7. This output is filtered in the RC circuit 1$6 to remove the sample rate from the discriminator and to pass the modulation rate. For ex-ample, for voice communication, t~e sample rate should be twice the maximum modulation rate to recreate the original information. Audio amplifier 158 drives speaker 160.
Trackin~ Control for Other Devices Because the wide bandwidth discriminator is so simple, 75'7~
' :
so small and requires so little signal power, it is possible to include it under the same magnetic pole piece as other YIG
tuned components to provide an output tha~ tracks a filter to a given input signal, For example, suppose it is necessary to track a multipole YIG filter to a fundamental fre~uency of a signal source while rejecting harmonics or spurious signals~ Cur-rentiy, such tracking can only be done in an open l.oop which makes it guite impractical over wide frequency ranges, parti-cularly when narrow band filters are required, or when opera-tion is desired over environmental extremes. Generally, in these ca~es, it i~ not possible to monitor the output of the filter to peak the desired response because this would ampli-tude modulate the output. Also, because -the filter is scanning at a ast rate, such ~uning could not be accomplished through the main tuning coil quickly enough to establish an effective tracking signal.
The embodiment shown in Fig.12 can be used to . achieve such results. A small portion (>-30dbm) o~ the in-put signal at Fo is coupled from a standard YIG filter 171 and through the band-pass/band-reject YIG filter 172 of the wide bandwidth discriminator 174. (See Figure 2.) Using techniques described previously, an output voltage corre-sponding to the difference between Fo and the quiescent operating frequency of the YIG housing is fed back into center frequency summing amplifier 176 through feedback net-work 178. Thiq feedback voltage is phased so that the total tuning voltage corresponds to a quiescent tuning current through electromagnet 180 that will center standard YIG
filter 171 at Fo~ Since the air coil tuning 182 is sampling -2~-~77579 at a much higher rate than the sweep rate tuning of Fo the feedback can be very effective in tracking the electromagnet tuning to the input signal Fo~ The only requirement is tha~
the ~IG filter 172 sphere in the discriminator 174 be under the same coarse tuning field as the YIG filter 171 sphere and that both spheres be aligned in frequency (basically the anisotropic fields are on the zero temperature drift axis).
It is also possible to couple directly off a filter 171 sphere to provide the resonator for the discriminator, Since the discriminator YIG filter 172 can be as ~mall or smaller than O.Q10 inches, such traaking i9 readily achieved. It is also possible that the filter aan be center-ed at a ~recluen¢y offse~ ~rom Fo by independently of~settiny the discrlminator center fre~uency. This can be accomplished in several ways including: a DC bias on the air core coil, varying the reference phase of the feedback loop, varying the spherc position or offsetting the resonance by using the anisotropic field of the YIG~ A particularly novel way would be to couple to one of the well known magnetostatic tracking resonances of ferrimagnetic materials to form the r~onator. These 0purious modes occur at a ~ixed offset from the main mode, are in~ependent of Erequency, and the ofset can be controlled over a considerable range by varying the saturation magnetization of the material. Such offsets are particularly useful in eliminating unwanted mixing products and feedthrough in high level heterodyne mixers and tracking local oscillators and preselectors.
In the tracking applica~ion of the wide ~andwidth discriminator, an alternative method in certain c:ases could 0 consist of cyclical sweeping of the input signal rather than ~LCP7757~
the resonant structure. The detected output and phase relationships with the sweeping means are identical to those generated by sweeping the resonant structure. A feedback signal generated in the phase comparator could be applied to either the signal source or the resonant device causing the two elements to track one another.
Another tracking applicatio:n of the discriminator is in a broadband harmonic generator where the output from the harmonic generator is used to control the frequency of the input source. Prior art harmonic generators use open-loop tracking between input source and output filter, are difficult to align and track, and have discontinuou~ output tuning o~ ~requency~ By u~ing the wide center frequency tuning range o~ the discriminator, the desired output ~re-quency can be tuned continuously while succes~ively filter-ing out the harmonics of the driving source In this manner, an inexpensive 1 to 12.4 GHz or 2 to 18 GHz signal source can be built with excellent linearity and frequency accuracy using only a single fundamental oscillator. This invention is the key element in providing closed-loop control of fre-quency as shown in Fig 13.
A center frequ~ncy tuning control unit 184 provides s the means for tuning the discriminator 186 over the entire , output range. At the same time, a high power input RF source 1~8 is coarsely tuned by harmonic tuning network 190 to a frequency equal to the output frequency divided by the harmonic number. A sample of the output is obtained in . - coupler 191 and an error correction signal generated in dis-criminator 186 is fed back through network 192 and summed with the coarse control in amplifier 194 to provide fine ~' ., :.
775~9 frequency control of the input source 188. Thus, the out-put frequency scan appears to be con~inueus even while the input source is being switched to corr,espond to the proper drive frequency for each harmonic. If the input source is a varactor oscillator, the discontinuity in the output can be as little as a fraction of a millisecond~
The output of step-recovery diode 196 consists of all of the harmonics of the input source 188~ The desired output frequency (NFo~ is filtered by YIG filter 198 to eliminate unwanted harmonics. Since the output frequency reerence (discriminator 186) is also a YIG device operating at the same frequency it is easy to track them versus fre-quen¢y. Al~ernatively, the YIG re~onator incoxporated in discriminator 186 can be tuned within the same magnetic tuning structure as YIG filter 198 thereby closiny the loop around the filter as well as the input source 188.
If the YIG resonator of discriminator 186 is placed before YIG filter 198, then a band-pass approach is needed to select the desired harmonic in the presence of other higher level harmonics. If the YIG resonator used for the discriminator is placed a~ter the output filter, as shown in Figure 13, then a band-reject approach can be used.
Each of these will be described along with exemplary ~77579 preerred embodiments of the invention.
Freque~y Control of S~nal Sources Solid state signal sources are available commercial-ly to cover the frequency range from 500 MHz through 18,000 MHz in a single unit. These sources generally use fundamental oscillators, which are electronically tuned over octave ranges by either varactor or YIG devices. Each octave range oscil-lator then requires individual calibration and control to provide the necessary frequency stability and accuracy. ~he net result is a system that is extremely expensive,.
The application of the wide bandwidth tunable dis-criminator in source control is to provide a single device which fixes the accuraay, stability, linearity and reset-ability of frequency, reduces frequency pushincJ and pulling;
eliminates turn-on drift; and provides continuous tuning at band cross-over frequencies.
Referring now to Figures 8 and 9, a frequency con-trol embodiment including the wide bandwidth frequency dis-criminator of Fig~ 2 is shown. Assume that the desired opera-tion consists of contiguous scan of RF oscillators 98, 100 and 102, and the RF coupler 103 sample~ the oscillator output to provide input ~or discriminator 108.
Initially, the center frequency control 104 is tuned to the lowest frequency of oscillator 98 and a coarse frequency tune is applied to oscillator 98 vla summing ampliier 106. The wide bandwidth discriminator 108 is centered at the correct Fo by tuning unit 104 (see Figure 9), Next the wide bandwidth position (~F max~ of ~igure 9~ is switched into operation ~y bandwidth adjust con-trol 110 and the bias voltage B~ is applied to oscillator 98 ~L077579 by bias control 99, In the ~F max~ position, the oscillator is quickly tuned near Fo by the closed loop feedback control network 101. If necessary, the gain of the loop can then be increased by switching in the ~F min. bandwidth on the dis-criminator. Now as Fo is tuned by cen~er frequency control 104, the oscillator will be forced to track Fo by the feed-back control action of the discriminator 108.
When the Fo control reaches the highest frequency in c)scillator 98, then oscillator 98 is turned off and oscillator 100 is turned on. The discriminator bandwidth i5 again switched to aF max. to capture oscillator 100, whlch is also forced to track the Fo of the discriminator.
Note that the Fo tuning is continuous and there is only a transient as one oscillator i9 turned of while khe other is turned on. The discriminator frequency reerence can be stopped momentarily at this time to allow any transients to die down before continuing the scan~
It is not necessar~ to turn the oscillator off and on i there is sufficient isolation in the RF switch 112 to meet the spuriouq signal requirements. However, high isola-tion fast RF qwitches are expensive and this offers an economical approach with significantly improved isolation over the techniques that are commercially available, Control of oscillator 102 is passed on from oscillator 100 in much the same way as from oscillator 98 to 100. At the end of the tuning range of oscillator 102 the sequence can be repeated. The sweep rate is limited by the scanning capability of the discriminator 108.
Frequency Measurement of Signal Sources Another useful capability of the disc:riminator is : . . .
~1~7757~
to augment the operation of an automatic heterodyne counter by determining the exact frequency offset of ~he unknown in-put frequency relative to a harmonic comb line. In this manner, the IF bandwidth of the amplifier following the RF
converqion process can be reduced, This signiicantly avoids two of the principle limitations of prior art heterodyne counters: the susceptibility to interference in the presence of multiplo signals and the difficulty in measuring low level signals due to the wide noise bandwidths of the IF amplifiers currently in use. It also allows for the u~e of harmonic comb line spacing much wider than is currently possible, e.g., 500 MHz versus 100 to 200 MHz, A wider comb line spaaing inareAses ~he ~F comb power available rom the harmonic generator at requencies above 12 GHz and provide~ better i~olation between adjacent com~ line. This further improves sensitivity and extends frequency measurement to much higher requencies (e.g., to 40 GHz). At the same time, the presence o an input signal and the selection of the proper harmonic for converting it to IF can be provided by the dis-criminator directly rom the RF input signal rather than a~ter~F proce~sing as i9 currently re~uired. A block diagram o an embodiment of the invention for improving automatic heterodyne counting is ~hown in Fig. 10.
The RF input signal at terminal 112 is applied through a 3db at~enuator 113 to a band-reject type YIG filter 114 which is included in the same magnetic housing as YIG
tuned harmonic generator 116. Harmonic generator 116 is driven from a ixed IF oscillator 118 which is phase locked to the ba~ic digital counter 120 as is common in the prior art. Power control }22 provides a means to turn oEf or reduce ~O--1C97~579 the input drive to the harmonic generator. Digital-to-analog converter 128 in response to drlve control 124 provides the tuning control for electromagnet 126.
Initially, the IF oscillator 118 input drive to the harmonic generator 116 is turned off and D/A converter 128 continually step tunes YIG filter 114 over the desired operating range ~e.g., 018 to 26 GHz) in steps coarsely equal to the comb line spacing (equal to the frequency of IF
oscillator 118). In this mode (Mode A), YIG ilter 114 operates in conjunction with RF detector 127, video amplifier 130, phase comparison means 132 and sweeping means 134 to form the discriminator of Figure 2. Sweeping mean~ 134 cyclically scans YIG filter 11~ over a ranye equal to or greAter than the comb line spacincJ. When the dlsarlminator intercepts an inpu~ signal, an output similar to Figure 3 is generated. The novel ability of this discriminator to select the largest signal will eliminate false-locking pro-blems.
When a signal is detected in the discriminator, D/A drive control 124 stops the scanning of electromagnet 126. A frequency measurement with the discriminator of the offset of the signal with respect to the nominal comb line frequency is used to select the best comb line for hetero-dyne conversion. The comb line number is preset in digital counter 120.
At this point, power control 122 turns on the IF
; drive to YIG tuned harmonic generator 115. Note, however, that electromagnet 126 has not necessarily tuned the YIG
harmonic generator exactly on to the desired comb line.
ThiS is accomplished ~y using a combination of the .
~77S~
YIG tuned harmonic generator 116 (now in a band-pass embodi-menk), RF detector 127, video amplifier 130, phase comparison means 132 and sweeping means 134 to form the discriminator of Figure 2. In this mode ~Mode B), RF switch 136 is opened to isolate the input signal from RF detector 127 and RF
coupler 138 applies the RF output from ~he harmonic generator 116 to RF detector 127. The output from pha~e compari~on means 132 is coupled back to D/A drive control 124 through feedback network 137. This fine control on the electro-magnet current centers the quiescent tuning of the housingat the exact comb line ~requency.
Therefore, the harmonic comb line that: will hetero~
dyne the lnput signal frequency lnto digital counter 120 has been selected and its output power optimized, At this time, operation reverts back to Mode A where the input signal off-set is now measured by the discriminator with respect to the updated tuning of the housing (now centered on the comb line frequency). Filter/ampliier natwork 140 is then tuned tto the offset frequency measured in network 141 by the discrim-- !
inator. The bandwidth of network 140 can be considerably less than the IF comb spacing thereby signi~icantly improving the capability of the automatic heterodyne counter to measure low level input signals and to operate in the presence o multiple inputs. The IF oscillator 118 providing an input drive to harmonic generator 116 is then turned on, and RF
detector 127 heterodynes the input signal and the selected comb line in the conventional prior art manner. Periodically, prefera~ly during the reset period of digital counter 120, the discriminator reverts into Mode A assuring that: network 0 140 is tuned to the proper IF offset and that the YIG tuned 16)775'79 harmonic generator 116 is tuned to the optimum comb line~
Demodulation of FM~ nals This technique provides a discriminator that makes possible practical low cost FM communications systems at microwave frequencies.
Referring to Figure 11 whereln an exemplary embodi-ment of an FM receiver having a frequency discriminator according to the invention is shown, the FM mo~ulated input signal is inaident upon a microwave antenna system 142. The signal is routed directly to RF detector 144 through tran~-mission line 146. The air coil drive 148 is set for maximum ~F to ensure capturing the signal, The offset frequency of Fo relative to the permanent magnet 149 bia~ field is measured in phase detector 150 and a DC current to air coil drlve 1~8 lS aenters the di~criminator (provides automatic frequency con-trol). Next, the ~F range is optimized to match the input signal deviation. Video amplifier 152 amplifies the signal generated as YIG sphere resonator 154 sweeps through the RF
input signal (either band-pass or band-reject can be used) by means of air core coil 155. Phase detector 150 basically compares the RF detector 144 output position to the sweep waveform creating a conventional FM discriminator curve using methods such as depicted in Figures 6 and 7. This output is filtered in the RC circuit 1$6 to remove the sample rate from the discriminator and to pass the modulation rate. For ex-ample, for voice communication, t~e sample rate should be twice the maximum modulation rate to recreate the original information. Audio amplifier 158 drives speaker 160.
Trackin~ Control for Other Devices Because the wide bandwidth discriminator is so simple, 75'7~
' :
so small and requires so little signal power, it is possible to include it under the same magnetic pole piece as other YIG
tuned components to provide an output tha~ tracks a filter to a given input signal, For example, suppose it is necessary to track a multipole YIG filter to a fundamental fre~uency of a signal source while rejecting harmonics or spurious signals~ Cur-rentiy, such tracking can only be done in an open l.oop which makes it guite impractical over wide frequency ranges, parti-cularly when narrow band filters are required, or when opera-tion is desired over environmental extremes. Generally, in these ca~es, it i~ not possible to monitor the output of the filter to peak the desired response because this would ampli-tude modulate the output. Also, because -the filter is scanning at a ast rate, such ~uning could not be accomplished through the main tuning coil quickly enough to establish an effective tracking signal.
The embodiment shown in Fig.12 can be used to . achieve such results. A small portion (>-30dbm) o~ the in-put signal at Fo is coupled from a standard YIG filter 171 and through the band-pass/band-reject YIG filter 172 of the wide bandwidth discriminator 174. (See Figure 2.) Using techniques described previously, an output voltage corre-sponding to the difference between Fo and the quiescent operating frequency of the YIG housing is fed back into center frequency summing amplifier 176 through feedback net-work 178. Thiq feedback voltage is phased so that the total tuning voltage corresponds to a quiescent tuning current through electromagnet 180 that will center standard YIG
filter 171 at Fo~ Since the air coil tuning 182 is sampling -2~-~77579 at a much higher rate than the sweep rate tuning of Fo the feedback can be very effective in tracking the electromagnet tuning to the input signal Fo~ The only requirement is tha~
the ~IG filter 172 sphere in the discriminator 174 be under the same coarse tuning field as the YIG filter 171 sphere and that both spheres be aligned in frequency (basically the anisotropic fields are on the zero temperature drift axis).
It is also possible to couple directly off a filter 171 sphere to provide the resonator for the discriminator, Since the discriminator YIG filter 172 can be as ~mall or smaller than O.Q10 inches, such traaking i9 readily achieved. It is also possible that the filter aan be center-ed at a ~recluen¢y offse~ ~rom Fo by independently of~settiny the discrlminator center fre~uency. This can be accomplished in several ways including: a DC bias on the air core coil, varying the reference phase of the feedback loop, varying the spherc position or offsetting the resonance by using the anisotropic field of the YIG~ A particularly novel way would be to couple to one of the well known magnetostatic tracking resonances of ferrimagnetic materials to form the r~onator. These 0purious modes occur at a ~ixed offset from the main mode, are in~ependent of Erequency, and the ofset can be controlled over a considerable range by varying the saturation magnetization of the material. Such offsets are particularly useful in eliminating unwanted mixing products and feedthrough in high level heterodyne mixers and tracking local oscillators and preselectors.
In the tracking applica~ion of the wide ~andwidth discriminator, an alternative method in certain c:ases could 0 consist of cyclical sweeping of the input signal rather than ~LCP7757~
the resonant structure. The detected output and phase relationships with the sweeping means are identical to those generated by sweeping the resonant structure. A feedback signal generated in the phase comparator could be applied to either the signal source or the resonant device causing the two elements to track one another.
Another tracking applicatio:n of the discriminator is in a broadband harmonic generator where the output from the harmonic generator is used to control the frequency of the input source. Prior art harmonic generators use open-loop tracking between input source and output filter, are difficult to align and track, and have discontinuou~ output tuning o~ ~requency~ By u~ing the wide center frequency tuning range o~ the discriminator, the desired output ~re-quency can be tuned continuously while succes~ively filter-ing out the harmonics of the driving source In this manner, an inexpensive 1 to 12.4 GHz or 2 to 18 GHz signal source can be built with excellent linearity and frequency accuracy using only a single fundamental oscillator. This invention is the key element in providing closed-loop control of fre-quency as shown in Fig 13.
A center frequ~ncy tuning control unit 184 provides s the means for tuning the discriminator 186 over the entire , output range. At the same time, a high power input RF source 1~8 is coarsely tuned by harmonic tuning network 190 to a frequency equal to the output frequency divided by the harmonic number. A sample of the output is obtained in . - coupler 191 and an error correction signal generated in dis-criminator 186 is fed back through network 192 and summed with the coarse control in amplifier 194 to provide fine ~' ., :.
775~9 frequency control of the input source 188. Thus, the out-put frequency scan appears to be con~inueus even while the input source is being switched to corr,espond to the proper drive frequency for each harmonic. If the input source is a varactor oscillator, the discontinuity in the output can be as little as a fraction of a millisecond~
The output of step-recovery diode 196 consists of all of the harmonics of the input source 188~ The desired output frequency (NFo~ is filtered by YIG filter 198 to eliminate unwanted harmonics. Since the output frequency reerence (discriminator 186) is also a YIG device operating at the same frequency it is easy to track them versus fre-quen¢y. Al~ernatively, the YIG re~onator incoxporated in discriminator 186 can be tuned within the same magnetic tuning structure as YIG filter 198 thereby closiny the loop around the filter as well as the input source 188.
If the YIG resonator of discriminator 186 is placed before YIG filter 198, then a band-pass approach is needed to select the desired harmonic in the presence of other higher level harmonics. If the YIG resonator used for the discriminator is placed a~ter the output filter, as shown in Figure 13, then a band-reject approach can be used.
Claims (14)
1. A frequency discriminator comprising ferrimagnetic reso-nator means adapted to receive an input signal, magnetizing means for tuning said resonator means to a quiescent center frequency, sweeping means for electronically providing a relative frequency sweeping between said resonator means and said input signal over a predetermined frequency range about the quiescent center fre-quency of said resonator means, detector means receiving an out-put from said resonator means, and means for comparing the phase of said sweeping means to the output of said detector to provide an output signal.
2. A frequency discriminator according to claim 1, wherein said sweeping means comprises means for electronically sweeping said ferrimagnetic resonator means about its quiescent center frequency.
3. A frequency discriminator according to claim 1, wherein said sweeping means comprises means for electronically sweeping said input signal frequency about its center frequency.
4. A frequency discriminator according to claim 1, wherein said sweeping means provides a linear frequency excursion.
5. A frequency discriminator according to claim 1, wherein said resonator means comprises a band-pass filter.
6. A frequency discriminator according to claim 1, wherein said resonator means comprises a band-reject filter.
7. A frequency discriminator according to claim 2, wherein the band width of said discriminator is electronically adjustable by electronically varying the amplitude of the relative sweeping of said resonator means.
8. A frequency discriminator according to claim 2, wherein said resonator is tuned by said sweeping means, said sweeping means comprising an auxiliary electromagnet and a current generating means driving said auxiliary electromagnet.
9. A frequency discriminator according to claim 1, in a frequency measuring system for measuring the offset frequency be-tween two signals, wherein one of said signals is generated by a low frequency oscillator and a harmonic generator and wherein said sweeping means provides a linear frequency excursion.
10. A frequency discriminator according to claim 1, in an automatic tuning control system, wherein said magnetizing means is a magnetizing structure, a device including at least one resonant element of ferrimagnetic material being enclosed in said magnetizing structure, and wherein means including said discrimin-ator are provided for maintaining a predetermined relationship between the tuning of said device and the frequency of a signal applied to said discriminator.
11. A frequency discriminator according to claim 10, in an automatic filter tracking system, wherein said device is a filter, said signal applied to the discriminator being the signal to be tracked and said predetermined relationship between the tuning of said device and the frequency of said signal is that they be at the same frequency, and wherein there are further provided means to tune the center frequency of the discriminator to the center frequency of said device and means responsive to the output of said discriminator to track the tuning of said device to said signal to be tracked.
12. A frequency discriminator according to claim 10, in a tracking tuner system, wherein said device is a band-pass filter and further including a local oscillator providing said output signal, said predetermined relationship being a frequency offset corresponding to the tuner intermediate frequency, and wherein there are provided means to tune said discriminator to the inter-mediate frequency offset from said band-pass filter, and means responsive to said discriminator output to track the tuning of the local oscillator to said band-pass filter with the inter-mediate frequency offset.
13. A frequency discriminator according to claim 10, in a ferrimagnetic harmonic generator system, wherein said device is a band-pass filter, said signal being the harmonic output of a oscillator means driving a harmonic generator means, whilst said predetermined relationship between said device and said signal is that said device be tuned to any selected harmonic output, and wherein there are provided means to tune the center frequency of the discriminator to the center frequency of the device, and means to coarsely tune the discriminator center frequency so that said relative sweeping over a common predetermined frequency range in-cludes the selected harmonic output.
14. A frequency discriminator according to claim 2, in an automatic frequency control system, wherein means are provided for coupling a portion of the power output of an electronically tunable oscillator means to said frequency discriminator as said input signal, said electronically tunable oscillator means being tracked to said frequency discriminator apparatus by means responsive to said output signal.
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Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA236,914A CA1077579A (en) | 1975-10-02 | 1975-10-02 | Frequency discriminator |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA236,914A CA1077579A (en) | 1975-10-02 | 1975-10-02 | Frequency discriminator |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1077579A true CA1077579A (en) | 1980-05-13 |
Family
ID=4104186
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA236,914A Expired CA1077579A (en) | 1975-10-02 | 1975-10-02 | Frequency discriminator |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA1077579A (en) |
-
1975
- 1975-10-02 CA CA236,914A patent/CA1077579A/en not_active Expired
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