CA1039660A - Method and means of predistorting digital - Google Patents

Abstract

METHOD AND MEANS OF
PREDISTORTING DIGITAL SIGNALS
ABSTRACT OF THE DISCLOSURE
A method and means for transmitting a bifrequency or biphase encoded binary data waveform over a transmission line in a way to minimize distortion is disclosed. Distortion is minimized by pre-distorting the transmitted waveform to substantially reduce the fre-quency content of the signal which results in less phase shift error per unit distance. One way in which this is accomplished is by reducing the amplitude of half waves of signals exceeding a predetermined interval of time to thereby reduce the low frequency content of the transmitted signal.

Description

BACKGROUND OF THE INVENTION
Field of the Invention This invention relates to data transmission systems, and more particularly to a bifrequency encoded binary data transmission system wherein the transmitted signal is predistorted to eliminate the need for equalization at the receiving end.
Descrjption of the Prior Art The distance over which an information carrying signal can be transmitted by a transmission line is limited primarily by two factors:
(1) the signal to noise ratio of the transmission, and (2) the phase distortion introduced by the transmission line. Increasing the signal to noise ratio of the transmission is relatively easily accomplished.
On the other hand, transmission line induced phase distortion is a more difficult problem since, even . . . . . . . . . .. .. .

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. . . ' . , ~. : ,, ~ : ' with a good signal to noise ratio, detection of the information may be possible only after complex transformation of the received signal. In a bifrequency encoded binary data transrnission sy~tem, an oscillograph of the transmitted signal reveals a characteristic pattern of two repeating small "eyes" each of which represents a half-bit period. The receiver gates on the center of these "eyes" for detection of the signal and on the level trans-itions for synchronization. Phase distortion which is a function of increas-ing line length has the effect of closing or causing the disappearance of these "eyes" with the result that detection of the signal is impossible.
In order to overcome the effects of phase distortion over long lines, it is known to provide an equalizer or filter at the end of the transmission line. The purpose of the equalizer is to introduce an inver~e distortion to compensate for the transmission line distortion. But, since the line distor-tion is a function of line length or distance, the equalizer must be tuned to the distance. This means that any change in line length effectively requires a redesign of the equalizer or an automatic equalizer. In addition to the inflexibility ~f equalizers, they have the additional disadvantage of being expensive to fabricate.
SUMMARY OF THE INVENTION -_ ~.... .
It is therefore an object of this invention to eliminate the need for equalization at the receiving end of a binary data transmission system.
It i8 another object of the invention to generate a bifrequency encoded binary data signal which can be readily detected at any distance up to a given maximum without any change in the sending or receiving circuitry.
It is a further object of the invention to extend the distance over which bifrequency encoded binary data signals can be transmitted without sacrificing short distance or direct-coupled transmission performance which Y09-72- 1 17 ;~

- 2 -~ 9660 is both very inexpensive and easily fabricated.
2 According to the present invention, the foregoing and other objects

3 are attained by reducing the low frequency content of the transmitted signal

4 in the line driver. This is done by lowering the amplitude of the second half of the long pulses to a value typically about one quarter or less of peak value.
6 Since each frequency propagates at a different speed and each has different 7 amounts of attentuation per unit distance of transmiæsion line, the resulting 8 narrowing of the band width has the effect of reducing distortion due to phase 9 shift error. This does not significantly affect signal detection on short lines ¦ and drastically reduces signal distortion on longer lines thereby simplifying 11 ¦ receiver design.
12 Implementation of the invention is quite simple and inexpensive using 13 ~ the techniques of modern technology. Logic circuitry is used to detect 14 I whether two consecutive half-bit intervals are equal. Two equal half-bit intervals represent a long pulse, and the logic causes the drive level to be 16 lowered in the Yecond half of the long pulse.
17 ¦ ERIEF DESCRIPTION OF THE DRAWINGS
18 The specific nature of the invention, as well as other objects, aspects 19 uses and advantages thereof, will clearly appear from the following descriptior ' and from the accompanying drawings, in which:
21 Figur~ 1 illustrates a bipolar waveform consisting of short and long 22 pulses;
23 Figure 2 illustrates a predistortion of the bipolar waveform wherein 2~L the second half of all long pulses is transmitted at a reduced level;
Figures 3 and 4 illustrate bipolar waveforms predistorted so that the 1~3C~660 1 trailing end of every pulse is lowered, the difference in the pre-d;stortion illustrated in the two Figures residing in the time at which the amplitude level of the pulses is lowered~
Figure 5 is a logic and schematic diagram illustrating one embodiment for implementing the predistortion of a bipolar waveform according to the invention;
Figure 6 shown on the sheet of drawings bearing Figure 1 is a timing diagram useful in understanding the operation of the logic and circuit diagram of Figure 55 Figures 7(a) through 7(f) are illustrations of oscillographs of a bipolar waveform without predistortion along varying lengths of a transmission line; and Figures 8(a) through 8(f) are illustrations of oscillographs of a bipolar waveform predistorted according to the teachings of the present invention at various lengths of a transmission line.
DESCRIPTION OF THE PREFERRED EMBODIMENT
In a bifrequency encoded binary data transmission system, the suitability of a received signal is judged based on two main aspects: ~.
1. At the receiving end the eye of the signal must open enough to allow reliable determination of the signal polarity, since the ~ .
signal is sampled near the center of the eye. .
2. Pattern dependent timing jitter in the received signal should .
be as small as possible; that is, the transitions through the zero level in the received signal should occur within a small time interval for both short and long pulses.
Figure 1 of the drawings illustrates a typical bipolar waveform ~

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1~3!~660 consisting of short and long pulses. For purposes of illustration, the long 2 pulses are considered to have a period 2T and the short pulses have a 3 period equal to T. Figure 1 illustrates two possible code as~ignments.
4 Thus, for example, two short pulses may represent a binary 1, and one long pulse may represent a binary 0. On the other hand for what is commonly 6 referred to as the biphase code, a binary 1 may be represented by the 7 transition frorn a positive to a negative pulse level in the middle of a bit 8 period, while a binary 0 may be represented by the negative to positive 9 transition of the pulse level in the middle of a bit period. The code assign-ment is, however, irrelevant with respect to the present invention since 11 the waveforms have identical features.
12 Given the bipolar waveform of Figure 1, it may be predistorted 13 before tran~mission over a transmission line according to any one of the 14 waveforms shown in Figures 2, 3 or 4. In the waveform shown in Figure 2, the second half of all long pulses is tran~mitted at a reduced level indicated 16 in the Figure by the drop in amplitude at the reference numerals 10. In the 17 waveforms illustrated in Figures 3 and 4, the trailing end of every pulse is 18 lowered. In the waveform of Figure 3, the transition from the high level 11 19 to the low level 12 occurs at every time T/2 after a polarity change in the ~-original waveform illustrated in Figure 1. In the waveform illustrated in 21 Figure 4, the ls~vel change occurs at time 3T/4.
22 In general, the waveforms of Figures 2, 3 and 4 are more costly 23 to implement in the order described. Selection of any one waveform depends 24 on the application. The bipolar waveform illustrated in Figure 1 is suitable only for relatively short distances. The waveform of Figure 2 roughly double 26 the range; however, both of the waveforms illustrated in Figures 1 and 2 yo9_72-117 _5_ generate con~iderable pattern dependent timing jitter at most distances.
2 Thus, in applications with chains of repeaters, the waveforms illustrated 3 in Figures 3 and 4 are preferred choice~, since they ~how very little timing 4 jitter at any distance up to the maximum. The relatively narrow peak pulse of the waveform illustrated in Figure 3 suffers slightly more attenuation 6 than wider pulses, but the waveform illustrated in Figure 4 is a little more 7 complex to implement.
8 Once a waveform has been selected, the optimum high and low 9 levels may be determined from the following procedure. The peak amplitude is chosen such that a sequence of all short pulses re~ults in a readily detect-11 able signal at the end of a maximu~n length of tran~mission line. Since 12 attenuation on normal transmission lines is proportional to the square root 13 of the frequency, the wide pulses ~uffer le~s attenuation due to their lower 14 frequency spectrum. These wider pulses are weakened at the sending end by lowering their trailing a~plitude enough, such that the amplitude of the 16 wide pulses at the end o$ the maximum length line exceeds the amplitude 17 of the short pulses by no more than about 20 to 50%O
18 In many cases it is desirable to adjust the amplitudes for minimum 19 timing jitter. For predistorted waveforms the trailing transition of a wide pulse leads other tran~itions at distances up to a certain cross-21 over distance, where the two tgpes of transitions coincide exactly. At 22 distances beyond the cross-over point, the trailing tran~ition of a wide 23 pulse lags the other transitions. A higher peak to low amplitude ratio 24 at the sending end will increase the timing lead for the short distances and decrease the lag for the maximum di~tances. The ratio could, 26 therefore, be considered optimum if the maximurn timing .. . . ~ : '' 1 lead equals the maximum timing lag. 1 ~ 3 9 6 6 0 An apparatus for generating the waveform illustrated by Figure 3, for example, is shown in Figure 5. The bipolar waveform of Figure 1, which may be denoted as code Q, is applied to the input terminal 21 which is connected to the enable or set input of flip-flop 22 and also to AND gate 23. A clock signal at a frequency of 2fo is applied to ter-minal 24 which is connected to the trigger input of flip-flop 22. To initiate operation, a clear and preset signal can be applied to ter-minal 25 which is connected to the reset input of flip-flop 22. The -true output of flip-flop 22 designated as Q is connected to the other input of AND gate 23. A second AND gate 26 is connected to receive the NOT true output of flip-flop 22 designated as ~. AND gate 26 also receives as an input the code ~ applied at terminal 27. The NOT or inverted outputs of AND gates 23 and 26 are applied as inputs to a third AND gate 28 which also receives the clock signal as an input.
Referring to the timing diagram of Figure 6, the clock signal at a frequency 2fo which is applied to the trigger input of flip-flop 22 and one input of AND gate 28 is illustrated at the top of the Figure. Code Q which is applied at terminal 21 to the set input of flip-flop 22 and AND gate 23 is illustrated as the second waveform. The opposite or in-version of code Q designated as code ~ is illustrated just below. The operation of flip-flop 22 is such that it is set in coincidence with the clock pulses when the bipolar code Q is at a high level or on the trailing edge of the bipolar code. The flip-flop 22 is reset in co-incidence with the clock pulses when the bipolar code is at a low level Or on the leading edge of the bipolar code. This operation is illus-trated by the waveforms designated as FFQ and FFQ representing the -. . .
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true and not true outputs of flip-flop 22. The outputs of flip-flop 22 are used 2 to gate the code and its inverse in AND gates 23 and 26. Specifically, the 3 code Q and the true output of flip-flop 22 produce a negative going gating 4 signal 29 at the output of AND gate 23. Note that this is the inversion of the true output of AND gate 23. In a similar manner, the inversion of the code 6 Q, that is, code Q, and the not true output of flip-flop 22 produce the output 7 gating signals 31 of the AND gate 26. Note again that these are negative 8 going or the not true outputs of AND gate 26. The negative going signals 29 9 and 31 from AND gates 23 and 26, respectively, may be considered as inhibiting signals which control the passage of the clock signal through AND
11 gate 28. The effect of this is illustrated in Figure 6 which shows the output 12 of AND gate 28.
13 The output of AND gate 28 is connected to one side of a collector 14 resistor R2. Transistors T10 through T16 represent a switchable current source. Such current sources are in common use in today's integrated 16 1 circuits. The collector and base of transistor T16 are connected in co~nmon 17 to the junction of AND gate 28 and the collector resistor R2. The emitter of 18 transistor T16 is connected to ground through re~istor R16, while the emitters 19 of transistors T10 through T15 are connected to ground through resistors ;~
R10 through R15, respectively. If a current I flows through the diode-21 connected transistor T16, nearly equal currents I flow through each of the 22 other six transistors, all of which have a common base with transistor T16.
23 Thus the common collector junction 32 of transistors T10 through T15 will 24 sink a current approximately six times the current of transistor T16. The amount of current flowing through T16 is determined by the value of the Y09-72-117 -~

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~)3~660 ¦ resistor R2, and this current is switched by the output of AND gate 28.
2 The operation of AND gate 28 is such that when the output i~ at a low level, 3 it will divert essentially the entire R2 current to ground by-passing transistor 4 Tl6.
The current source comprising tranYistors TlO through T16 drives 6 a push-pull line driver. The current drive provided by the transistors TlO
7 through Tl6 is the peak current as will be clear from the description of the 8 driver operation. In addition to the peak current source, there is a parallel 9 steady-state current source represented by the transistors TS, T6 and T7.
The base of transistor T5 is connected in common with its collector through 11 resistor Rl to a source of positive potential. The emitter of transistor T5 12 is connected through resistor R3 to ground, while the emitters of transistors 13 T6 and T7 are connected through resistor R4 to ground. The collectors of 14 transistors T6 and T7 are connected in common to the co~nmon junction 32.
As in the peak current source, the bases of transistors T6 and T7 are commo~
16 with the base of transistor T5. The common junction 32 is connected to the 17 emitters of transistors Tl and T2 which are current steering transistors.
18 The collectors of transistors Tl and T2 are connected to opposite ends of a 19 center tapped primary winding of transformer 33. The center tap of the primary winding of transformer 33 is connected to a source of positive voltage 21 and through a decoupling capacitor C2 to ground. If the base voltage on 22 transistor T2, for example, is sufficiently higher than on the base of trans-23 istor Tl, all the current, both peak and steady-state, will flow through 24 transistor T2 and the left section of the transformer primary, generating a more positive voltage on the output terminal 34 when referenced to output - . . ..

1 terminal 35 of the secondary winding of transFormer 33. With transis-tor Tl conducting and transistor T2 off, the output polarity at out-put terminals 34 and 35 is reversed.
The bivalue code wave Q and Q is connected to the circuitry which drives the bases of transistors Tl and T2. More specifically, the code Q applied at input terminal 21 is connected to the emitters of trans-istors T9 and T18, and the inverse of the code Q is applied to the ~;
emitters of transistors T8 and T17. Each of the transistors T8, T9, T17 and T18 is diode-connected, i.e., their base and collector junctions are connected in common. The collector of transistor T9 is connected ~
to the base of transistor T4 and through a collector resistor R6 to a ~-common junction 34. In a similar manner, the collector of transistor T8 is connected to the base of transistor T3 and through a collector re-sistor R7 to the common junction 34. The collector of transistor T18 is connected in common to the base of transistor T2 and the emitter of transistor T4, while the collector of transistor T17 is connected in com-mon to the base of transistor Tl and the emitter of transistor T3.
The collectors of transistors T3 and T4 are both connected to a source of positive potential. The common junction 34 is connected also to a source of positive potential through series connected resistor R5 and diode-connected transistors Tl9 and T20. Diode connected transistors Tl9 and T20 and resistor R5 are used to limit the maximum positive swing at the base of transistors Tl and T2 in order to avoid saturation.
Transistors T3 and T4 are emitter followers and provide the considerable base drive current required for the current steering transistors Tl and T2. Transistors T8 and T9 in diode connection disconnects the emitter followers in the output of the code source from the . .
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Il 1~;~9660 driver. This may be required where the code source up-level is not well Z controlled and may rise too high. Transistors T17 and T18 provide a 3 sufficient voltage drop to avoid shunt currents from transistors T3 and T4 4 into the code source outputs at the down level. Resistors R8 and R9 con-nected across the secondary winding and the primary winding, respecti~ely, 6 provide a match for the external transmission line. The capacitor Cl 7 connected across the primary winding is u~ed to ~mooth the waveform 8 and reduce possible radiation.
9 In operation, the code and its inverse Q and Q control steering transistors Tl and T2. The output of AND gate 28 controls the peak current 11 source comprising transistor~ T10 through T16 when the Code Q iB at a high 12 level, transistor T2 will conduct a peak current for the duration of one clock 13 pulse. The steady-state current source comprising transistor T5 through T7 14 p~4~ides the low le~rel pedestal for the duration of the output signal. When the inverse of the code Q is at a high level, transistor Tl will conduct a 16 peak current for the period of one clock pulse and thereafter the steady-state 17 current source provide the low pedestal for the duration of the output signal.
18 This operation is illustrated by the waveform at the bottom of Figure 6.
19 Figures 7(a) through 7(f) illustrate the line driver output without predistortion of a 1. 39 Mbit/sec signal over varying lengths of 24 gauge 21 twisted pair transmission line. Notice that at the end of 4, 000 feet of 22 transmission line as illustrated in Figure 7(e), one of the two small eyes 23 has disappeared for the regular unequalized signal. In comparison, 24 Figures 8(a) through 8(f) illustrate the signal on the transmission lines when subjected to predistortion according to the teaching~ of the invention.
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yO9-72-117 At all lengths of transmission line, the predistorted signal illustrated in 2 Figures 8(a) through 8(f) does not need any further equalization. Heavier 3 gauge wire like the common AWG22 in telephone exchange areas will, of 4 course, allow transmission over longer distances, It will be apparent that the embodiment æhown i9 only exenlplary 6 and that various modification~ can be made in construction and arrangement 7 within the scope of the invention as defined by the appended claims. For 8 example, while the circuit ~hown in Figure 5 is intended to generate the 9 waveform shown in Figure 3, those skilled in the art will recognize that simple logic circuits can be implemented which generate the waveform 11 shown in Figures 2 and 4 or, for that matter, any variation thereof de-12 pending upon the desired application.

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Claims (7)

The embodiments of the invention in which an exclusive property or privilege is claimed are defined as follows:

1. In a binary encoded data transmission system, an apparatus for predistorting the encoded wave form over a transmission line in a manner to minimize phase distortion due to the transmission line, said predistorting apparatus comprising:
input means for accepting a source of bipolar signals comprising a sequence of short and long pulses having periods T and 2T, respec-tively, and representing a binary code, push-pull line driver means connected to said input means and controlled by said bipolar signals for driving said transmission line, gated electrical source means connected to said line driver means for intermittently supplying a first driving signal thereto, steady-state electrical source means connected to said line driver means for supplying a second driving signal thereto in the range of 20 to 50% of said first driving signal, and gating means connected to said input means and responsive to said bipolar signal for gating said gated electrical source means on for a time period less than or equal to T after each polarity reversal in said bipolar signal,

2. An apparatus as recited in claim 1 wherein said gating means in-cludes:
timing means for establishing a timing period shorter than the period of a short pulse, and logic means responsive to said timing means for controlling said gated electrical source means at a predetermined time after the change in polarity of the original wave.

3. An apparatus as recited in claim 1 wherein said gating means com-prises:
a source of clock pulses having a frequency Nf0 wherein N is an integer greater than 1 and f0 is the code frequency, and bistable means having as inputs said bipolar signals and said clock pulses, said bistable means being set on the coincidence of a transition of said bipolar signals and a clock pulse and thereafter reset on the next succeeding clock pulse.

4. An apparatus as recited in claim 3 wherein said gating means further comprises:
coincidence means connected to the output of said bistable means and to said source of clock pulses for generating a control signal for controlling said gated electrical source means.

5. An apparatus as recited in claim 4 wherein said coincidence means comprises:
first AND gate connected to receive said bipolar signal and the true output of said bistable means, second AND gate connected to receive the inversion of said bipolar signal and the NOT true output of said bistable means, and third AND gate connected to receive said clock pulses and the inverted output of said first and second AND gates.

6. An apparatus as recited in claim 1 wherein said push-pull line driver means comprises:
a transformer having primary and secondary windings, said secondary winding being connected to said transmission line, said primary winding having a center tap connected to a source of reference potential, and first and second coupling means connected to opposite ends of said primary winding and in common to both said gated electrical source means and said steady-state electrical source means.

7. An apparatus as recited in claim 6 wherein both said gated electri-cal source means and said steady-state electrical source means comprise:
a diode-connected transistor having collector, base and emitter, said collector and base being connected in common, a collector resistor connected between the collector of said diode-connected transistor and a source of potential, the output of said gating means being connected to the junction of the collector of said diode-connected transistor and said collector resistor in said gated electrical source means, 14 an emitter resistor connected between the emitter of said diode-connected transistor and a potential reference, and a plurality of current-sinking transistors each having collector, base and emitter, the bases of all of said current-sinking transistors being common to the base of said diode-connected transistor, the col-lectors of said current-sinking transistors being connected in common to said first and second coupling means.