AU602351B2 - Adaptive gain control amplifier - Google Patents

Description

ignature(s) of declarant(s).

To: The Commissioner of Patents, Australia ~r

LA

,;4 AU-AI-78075/ 8 7 WORLD INTELLECTUAL PROPERTY ORGANIZATION International Bureau 01

PCT

INTERNATIONAL APPLICATION PUBLISHED UNDER THE PATENT COOPERATION TREATY (PCT) (51) International Patent Classification 4: (11) International Publication Number: WO 88/ 01453 03G 3/20 Al (43) International Publication Date: 25 February 1988 (25.02.88) (21) International Application Number: PCT/AU87/00259 (81) Designated States: AT (European patent), AU, BE (European patent), CH (European patent), DE (Euro- (22) International Filing Date: 12 August 1987 (12.08.87) pean patent), FR (European patent), GB (European patent), IT (European patent), JP, LU (European patent), NL (European patent), SE (European patent), (31) Priority Application Number: PH 7468 US.

(32) Priority Date: 13 August 1986 (13.08.86) Published (33) Priority Country: AU With international search report.

With ompn d ,nirn (71)1 SECTION 34(4)(a) DIRECTION SEE FOLIOIL NAME DIRECTED A4odLA Auclto Appicth'6-s Py. L c, sAl so+ EV iczn oeRj V (71)7ZTjAppIrm 'mnventor uor AU u oKy -Try -r E- L Anatol, Zygmunt [AU/AU]; 21 Walstab Street, East 3 1 Brighton, VIC 3187 I AUSTRALIAN (74) Agent: SANDERCOCK, SMITH BEADLE; P.O. MAR 1988 Box 410, Hawthorn, VIC 3122 FI PATENT OFFICE i '4 s.'ctror j It (54) Title: ADAPTIVE GAIN CONTROL AMPLIFIER As I 12- I A

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(57) Abstract U255 I 29 2

I

30 27 25 I 26 I An adaptive gain control amplifier (12) is described in relation to an audio system for adjusling output volume of a loudspeaker (16) automatically in response to an interference signal (21) such as background noise. The amplifier (12) includes a divider circuit (30) which computes the ratio of a time average of a corrupted signal (20, 21) which is a combination of a controlled output signal (20) and the interference signal to produce a control voltage which controls the gain of the amplifier. To reduce the dynamic range of the quotient of the ratio the divider circuit (30) incorporates a logarithmic ratio circuit. The circuitry of the amplifier includes a "sensing channel" and a "reference Thannel" for extraction of the above averages of the corrupted signal and controlled output signal, respectively. A method is also claimed.

WO 88/01453 PCT/A U87/00259 1 ADAPTIVE GAIN CONTROL AMPLIFIER 2 This invention relates to adaptive gain control 3 amplifiers for automatic enhancement of desired signals 4 delivered to a receiver or receivers which are susceptible to external interference of a kind which may corrupt the 6 received signal. More specifically the invention relates to 7 an adaptive gain control amplifier and to a method of 8 adaptive gain control which has particular utility in audio 9 systems such as public address systems, telephones, automotive and domestic hi-fi and the like where there is a 11 need to adjust the output volume of a loudspeaker 12 automatically in response to variations in background noise.

13 It will however, become readily apparent to those skilled in 14 the art that there are many other applications of the invention such as for example in adaptive radio transmitter 16 control and automatic contrast control in illumination.

17 Most known electronic circuits for signal enhancement 18 are designed to minimize the signal degradation at the 19 source, during electronic processing or during storage or playback (for example, Dolby (Registered Trade Mark) or DNR 21 (Registered Trade Mark)). However, these techniques are not 22 capable of signal enhancement directly at the receiver as is 23 the case with this invention. Thus, the action of the 24 adaptive gain control amplifier is complementary to that offered by the abovementioned circuits.

26 Most of the circuits designed specifically for E.

27 acoustical systems such as public address, intercoms, 28 telephones, automotive and domestic audio, where the desired 29 signal enhancement is required to occur at the ear of the listener are based on two techniques. The first technique 31 requires the corrupted .,-gnal to be decomposed into its 32 desired and interference componets. Simpler systems use 33 apriori separation based on assumed or empirically 34 determined spectral (or temporal) characteristics of the signal or interference. Examples of this technique are 36 found in automotive audio modules such as Hitachi ASLC 37 (Automatic Sound Level Control) and Alpine 3015 38 (Preamplifier, 7 Band Computer Graphic Equalizer). These I i t9--.

L WO 88/01453 2- PCT/A-U87/00259 1 are specifically designed for automotive audio use and rely 2 on measured road noise spectral characteristics, which 3 renders them restrictive, arbitrary and generally low 4 performance. Ideally, dynamic or adaptive decomposition of the corrupted signal into its desired and interference 6 components would improve the performance of these systems.

7 However, in most cases this decomposition or analytical 8 separation is very complex and costly to implement.

9 This is because in acoustical applications sufficiently precise separation of the corrupted signal into its desired 11 and interference components is rendered complicated or 12 impossible because of diffraction and mult-i-path effects 13 which preclude simple spatial separation as well as temporal 14 and spatial distortions induced by transducer and medium non-linearities, time delays, reverberations, and other 16 reflection, refraction and absorption phenomena resulting 17 from material properties and discontinuities thereof.

18 Imperfect separation renders some known enhancement system 19 susceptible to instabilities (such as howling in public address systems) or reduces their effectiveness. Some known 21 systems (for example Alpine) avert the first consequence by 22 imposing severe restrictions on gain control thereby 23 reducing it, for example, to a single discrete step 24 adjustment. Another technique (noise cancellation), uses a controlled filter to substract out the said interference, 26 employing a least mean square algorithm to derive the 27 control signal. This technique is normally used for signal 28 enhancement at the so'irce, but can be adapted for use on 29 receivers. The amount of signal enhancement obtainable by this method is the highest known at this time. However, 31 this perceived enhancement varies drastically depending on 32 the listener's location, and can even result in actual 33 degradation at some locations. In addition, the 34 implementation of the algorithm and filter is expensive and the system is rather slow and requires a "learning period".

36 Accordingly, it is an object of this invention to 37 provide an improved automatic signal enhancement processor 38 which avoids or at least reduces one or more of the 3aforementioned disadvantages of known apparatus.

The invention therefore provides an adaptive gain control amplifier for converting an input signal to an. analog controlled output signal in response to varying interference signals which mix with said output signal and result in a corrupted signal, said amplifier comprising: j an amplifier stage for receiving said input signal and A providing an output signal having a contvolled output signal level; a prescaler circuit for receiving said output signal of said amplifier stage and comprising a secondary gain control amplifier for prescaling said output signal of said amplifier stage to provide a prescaled output signal; a time averaging circuit means, connected to said prescaler circuit, for providing a time averaged output signal of said prescaled output signal of said amplifier stage; a sensor means for detecting said corrupted signal and for providing a time averaged output signal of said S 0d. corrupted signal; a divider circuit having a first input receiving said time averaged output signal of said prescaled output signal from said amplifier stage, a second input receiving said time averaged output signal of said corrupted signal, and an i output provided to a control input of said amplifier stage, said divider circuit comprising means for providing a control signal at said output which is a ratio of said time averaged output signal of said corrupted signal and said time averaged output signal of said prescaled output signal f 30." from said amplifier stage to said control input of said amplifier stage; and wherein said prescaler circuit has a control input which receives a control signal which is proportional to said control signal at said output of said divider circuit.

In a further form the invention provides a method of processing an input signal to produce a controlled output Q signal when said output signal has an interference signal 1 ixed therewith to produce a corrupted signal at a receiver, SV psspe.016/aranda 90 6 3A_ said method comprising: sensing and processing said corrupted signal to produce a first control signal for controlling processing of said input signal to compensate for said interference signal, said processing of said corrupted signal comprising producing a time-averaged version of said controlled output signal, producing a time-averaged version of said corrupted signal, computing a ratio of the modulus of said time averaged version of said corrupted signal and the modulus of said time averaged version of said controlled output signal to produce said first control signal, employing said control signal to control an o amplification stage of an adaptive gain control amplifier which produces said output signal, providing a second control signal proportional to Soso said first control signal, and prescaling said controlled output signal in response to said second control signal prior to computing said ratio.

In order that the invention may be more readily understood, a particular embodiment will now be described with reference to the accompanying drawings wherein: Figure 1 is a basic schematic block diagram depicting an automatic signal enhancement processor according to the r° invention; Figure 2 is a more specific schematic block diagram of the invention disclosed in Figure 1 and directed specifically to an acoustic application thereof.

Figure 3(b) and 3(c) show alternatives, in detailed circuit format, of a gain control amplifier of Figures 1 and 2..

Figure 4 is a detailed circuit diagram of a logarithmic divider of Figures 1 and 2.

Figure 5 is a detailed circuit diagram of a precision s 96 sspC O 016/aranda 90 6 20 f PiV Ifr WO 88/01453 4 PCT/AU87/00259 1 rectifier of Figure 2, 2 Figure 6 is a graph showing the response of a precision 3 rectifier of Figure 5, and 4 Figure 7 is a more detailed circuit diagram of a gated integrator shown in Figure 2.

6 In Figure 1 a transmitter 10 transmits a signal 11 7 which, in this embodiment, is an audio signal but in another 8 embodiment could be a radio frequency or light signal for 9 example. The signal processor of the embodiment is shown within the broken lines 12 and is located between the 11 transmitted signal 11 and an intended receiver 13. In this 12 case the receiver 13 is the ear of a listener. The 13 embodiment could be that of a Public Address System, for 14 example. Some components of the signal processo 7 12 may be present in a conventional audio system to provide 16 amplification. An input transducer 14, gain control 17 amplifier 15 and output transducer 16 would normally provide 18 conversion and amplification of the audio signal which, upon 19 leaving the output transducer 16 travels the airwaves to the receiver 13. In other words, the input transducer 14 21 intercepts the desired signal 11 and converts it into 22 electrical form on connection 17. The resultant electrical 23 signal on 17 is multiplied, in amplifier 15, by a control 24 voltage on 18 whereby the enhancement is affected. The enhanced output of amplifier 15 on connection 19 is 26 converted into its final, usable form, by the output 27 transducer 16 to produce the audio signal 20 which is heard 28 by the listener.

29 The derivation of the control voltage 18 is based on /r 30 the computation of the quotient between relevant statistical 31 averages relating to the corrupted and desired signals 32 respectively. Henceforth, the circuits involved in the 33 extraction of the above averages are referred to as the 34 "sensing channel" which relates to the sensing of the corrupted signal, and the "reference channel", which relates 36 to the desired signal.

37 The corrupted signal, which is a combination of the 38 audio signal 20 and interference signal 21 is sensed and WO 88/01453 PCT/A U87/00259 1 transformed into its electrical equivalent by sensor 2 transducer 22. The interference signal 21 emanates from a 3 source or sources referenced by 23. The electrical 4 equivalent of the corrupted signal appears on connection 24 to functional converter and time averager 25. The objective 6 here is to ensure that the corrupted signal intercepted by 7 sensor transducer 22 is representative of the signal 8 incident upon the receiver 13. The signal on 24 is 9 converted by functional converter and time averager 25 into a functional form appropriate to conform with the 11 sensitivity function of the receiver 13. This is then time 12 averaged in accordance with the requisite signal statistics 13 over the gating interval determined by a clock signal on 14 connection 26. Thus the output on connection 27 from the device 25 represents the required measure of the corrupted 16 signal over the duration of the averaging or gating period 17 t as controlled by the clock on connection 26.

18 A similar process is performed by another functional 19 converter and time averager 28 within the reference channel, wherein the enhanced (desired) electrical signal is 21 subjected to a similar transformation and averaging. The 22 output of 28 is provided on connection 29 which together 23 with connection 27 is applied to the divider 30 which 24 computes the ratio between the two transformed and averaged signals. This resultant quotient is conditioned by 26 interface circuit 31 in order to render it suitable for 27 application as the control voltage on connection 18.

28 Signals appearing on 19 and 24 may be prescaled (by slaved 29 gain controlled amplifiers identical to 33) prior to their application to 28 and 25 respectively. This allows for 31 different types of enhancement to be affected resulting in 32 different governing equations and asymptotic and transient 33 behaviour.

34 The control loop is recursive, with updates in the control voltage on 18 occurring one computation cycle 36 (approximately At) after the sensing. In order to ensure 37 loop stability, the correction should be a monotonic 38 function of signal degradation. It should be noted that the ,i WO 88/01453 -6 PCT/A IU87/00250 1 signal enhancement processor 12 offers quantized or 2 continuously variable adjustments of the control voltage on 3 18. More detailed explanations of the internal operati~on of 4 the processor will become apparent by way of description relating to the more detailed circuitry of Figures 2 and 3.

6 Referring now to Figure 2 which shows an acoustical 7 application in more detail, it will be noted that features 8 of the diagram common with or similar to those of Figure 1 9 are referenced by corresponding reference numerals. In most audio applications, the desired signal is already in its 11 electrical form and hence available for enhancement by the 12 gain control amplifier 15. The gain contro. amplifier 13 may take various forms depending on cost, distortion, 14 isolation between signals on input connection 17 and output connection 19 and control voltage on 18, and dynamic 16 enhancement range. Figures and illustrate 17 examples of gain control amplifier 15 circuits designed 18 specifically for 19 low cost, high fidelity and 21 simplicity.

22 The enhanced signal on 19 is usually amplified bDy a power 23 amplifier (not shown), and transformed into its acoustical 24 form 20 by loud speaker 16 (output transducer of Figure 1) The dynamic range of most audio systems is in excess of 26 dB and since the control loop relies on ratiometric 27 computation, the dynamic range of the quotient can exceed 28 100 dB. Therefore, some compression is advantageous within through the the use of a logarithmic ratio circuit 30, as 31 will be described in detail hereinbelow. In order to ensure 32 stability, and acceptable transient and asymptotic 33 behaviour, an additional gain control amplifier in the form 34 of a prescaler 33 of gain Gm is provided in the reference channel (control voltage is shared with the control voltage 36 of the main gain control amplifier 15, and individual 37 amp'lifier characteristics are assumed to be well matched).

38 The governing equation for voltage adjustments 47, in the WO 88/01453 PCT/AU87/00259 *7 1 control voltage (18) immediately following the n th gating 2 interval is: 3 4 oG loGS N AV log n+ log n n n Gz/ Gm+1 6 n n 7 where n refers to n th gating time interval 8 n 1 refers to N 1 th gating time interval 9 Sn is input signal (17)

N

n is sensed interference component (21) 11 6 is gain factor of logarithmic ratio circuit 12 -c and subsequent amplifier (31).

13 m is the number of prescalers in reference channel (negative 14 if prescalers are in sensing channel).

(proportionality coefficients of all transducers are 16 assumed to be 1 for simplicity).

17 If the noise is not correlated with the signal over 18 the gating period 19 2 2 2 G S +N E n n n 2 2 og 2 m+2 2 21 G Sn 22 23 Static Asymptotic Solution is one for which &V 0, 24 Hence G Go 2m+2 2 2 2 2 G S N. =0 26 27 It can be shown that for m 1 28 2 29 G 1 +2 1 2 31 which has desirable properties (prescaler gain Gn), for 32 example, for high S/N ratios 33 G 34 For high N/S ratios 36 N

I'.

L i. WO 88/01453 PCT/AU87/00259 1 This functional implementation has been studied for its 2 transient behaviour, and it has been established that for 3 e< 0.65 there is no overshoot. (e has similar properties to 4 a damping coefficient).

As a compromise between rapid convergence and 6 undesirable ringing, e= 0.6 was chosen as a practical value.

7 If required, upper and lower limits can be applied to AV by 8 a limiter (not shown) and a deadband circuit (not shown) 9 respectively. Practical values chosen are 15 dB and 1.5 dB respectively.

11 In addition to the prescaler 33 the reference channel 12 processing consists of a gain stage and filter 34, used to 13 scale the magnitude of the desired output signal on 14 connection 35 and limit its bandwidth to a relevant frequency range, for example, to match the response 16 characteristics of the receiver 13 in Figure 1. This is 17 followed by a precision rectifier, or absolute value circuit 18 36 which removes the polarity dependence of the signal, a 19 requirement imposed by the characteristics of the sensing microphone 22 (sensor transducer in Figure 1) and the human 21 ear. The output on 37 from circuit 36 is time averaged by 22 the integrator 28 over the gating interval At as controlled 23 by the clock signal on 26, resulting in a mean average 24 deviation (MAD). Alternative functional implementations can involve the RMS or simply the mean square MS.

26 The averaging period At is chosen to be approximately 27 200 mS. This reflects a compromise between the accuracy of 28 averaging and the desired speed of response. The former is 29 influenced by reverberation and other environmentally induced effects. Appropriate choice of the latter is 31 required. This is because excessively rapid response 32 updates are likely to result in objectionable incidental 33 modulation distortion, whilst unduly slow response can 34 result in unacceptably long correction delays and consequent lack of transient suppression. Gradual application of the 36 correction is a potential solution, but this requires proper 37 weighting of the statistical average to compensate for the 38 time dependence of the gain during averaging. r i ii WO 88/01453 PCT/AU87/00259 1 The sensing channel in Figure 2 consists of a sensing 2 microphone 22 (sensing transducer in Figure followed by 3 a gain stage and filter 38, precision rectifier/filter 39 4 and gated integrator 40, whose functions are similar to their equivalent counterparts in the reference channel 6 comprising components 34, 36 and 28, respectively, with 7 slight parameter variations to compensate for the microphone 8 characteristics and the required frequency response of the 9 channel. The outputs on connections 27 and 29 from 40 and 28 respectively, thus represent the MAD values for the 11 preceding interval At relating to the reference channel and 12 the sensing channel respectively.

13 These outputs are subsequently applied to the precision 14 logarithmic ratio circuit 30 corresponding to the divider 30 of Figure 1. The logarithmic characteristic described 16 above, is implemented herein to increase the operational 17 dynamic range and to optimize the matching to the amplitude 18 response function of the typical human ear. The output of 19 30 is processed by conditioning circuit 31, the purpose of which is to sample the output of 30 and maintain it for the 21 duration of the gain adjustment period determined by a clock 22 on connection C. If required, conditioning of this held 23 output should be performed in order to match the response 24 caaracteristics of the control input on 18 to 15 and 33.

This may include scaling and offset adjustments, limit 26 setting (for minimum and maximum corrections, functional 27 transformation, for example, log-linear, impedance matching, 28 deglitching, attack time control, etc. Some of this 29 matching controls the dynamic approach to equilibrium or tracking by acting as damping.

31 Figures 3 and show various forms of the 32 gain control amplifier 15 in more detail. In each diagram 33 the capacitors 41 are 1' 1 F and the capactitor 42 in Figure 3 34 is a 0.01IF capacitor. The circuitry of Figure 3 (a) utilizes a transistor 43 which is a 2N3904 and a diode 44 36 which is an IN914. Resistor 45 is a 470KS resistor and 37 resistor 46 is a 22Kn resistor. The circuitry of Figure 3 38 utilizes an operational amplifier 47 which is an LM318 WO 88/01453 PCT/AU87/00259 10 1 and a FET 48 which is a PN4391. The resistor 49 is 2 resistor 50 is 20KO and the two resistors 51 are 1Mn 3 The circuitry of Figure 3 is a much simplified 4 circuit which incorporates intergrated circuit package 52 which is an MC3340, which uses a Gilbert type trans- 6 conductance multiplier.

7 Reference should now be made to Figure 4 which shows a 8 more detailed circuit diagram of the logarithmic ratio 9 circuit 30. Inputs to the circuit 30 of Figure 4 appear on connections 27 and 29 which correspond to the connections 11 shown in Figure 2. The circuit is divided into two parts 12 and 61 respectively, which are separated by the broken line 13 shown in the drawing. The first part 60 is a voltage to 14 current conversion part and the second part 61 is the logarithmic circuit. The output appears on connection 62.

16 In the circuit 30 the resistors 53 are 10K2 resistors, 17 resistors 54 are 1KO resistors and resistors 55 are 12KO 18 resistors (all resistors have a 1% tolerance). The 19 transistors 56 are matched transistors of type LM394. The integrated circuits 59 are all operational amplifiers of 21 type TL074. The capacitors 57 are 100 pF and the capacitors 22 58 are 0.001iF.

23 Input voltages on 27 and 28 are converted into currents 24 11 and 12 by the voltage to current converters. These currents are applied to the matched pair transistors 56 26 whose base/emitter voltage is dependent on the log of the 27 current. These base emitter voltages are subtracted by the 28 final stage to yield the ratio. This method offers wide 29 dynamic range, approximately 80 dB per channel or 160 dB in 30 the quotient. Resistor tolerance and operational amplifier 31 offsets are important in ensuring proper behaviour ,f the 32 overall circuit.

33 Reference should now be made to Figure 5 which shows 34 the precision rectifier/filter circuit 39 in more detail.

The circuit of the absolute value circuit 36 (Figure 2) is 36 very similar to the circuit for the precision 37 rectifier/filter circuit 39 and only differs in component 38 values due to the different gain requirement. The ^YLYltl~l~~:I WO 88/01453 PCT/AU87/00259 1 integrated circuits 63 in Figure 5 are operational 2 amplifiers of type 0P07 for low offset or type LF412 for 3 economy. The resistors 64 are 10KQ as are resistors 4 The capacitor 66 is 0.001 F and the capacitors 67 are 1nF.

Diodes 68 are of type IN914. The characteristics of the 6 circuit are represented by the following formula: 7 v2 7 VOUT IN 8 9 The filter response of the circuit of Figure 5 is represented by the graph shown in Figure 6 where the slope 11 70 is a slope of 6 dB per octave and the slope 69 is 12 dB 12 per octave.

13 Referring now to Figure 7 which shows the integrator 14 circuit 40 in more detail, it should be pointed out that this circuit is the same as the circuit used for the 16 integrator 28 of the reference channel. The resistors 71 17 are 27KQ resistors and the capacitor 73 is a IiF capacitor.

18 The integrated circuit 72 is an amplifier type LF412 and the 19 switch 74 is type CD4066. Connection 26 to the switch 74 is the clock connection shown in Figure 2. The characteristics 21 of the circuit are represented by the. following formula: 22 T 23 VOUT INdt 24 The requirements of the integrator circuits 28 and 26 are low leakage of the switch and low operational-amplifier 27 offset.

28 Apart from electronic circuit imperfections, the 29 overall effectiveness of the signal enhancement processor of this invention is dependent on the deviations from perfect 31 representation of the corrupted signal by the sensing 32 microphone 22. For perfect representation, this microphone 33 must be identical and coincident with the actual receiver 34 (the listener's ear). Since this is impossible, any system reliant on the sampling of the corrupted signal such as is 36 the case here and with adaptive cancellation systems, 37 suffers some degree of misrepresentation. This is of 38 limited consequence in cases where this effect is static, p.-i WO 88/01453 1PCT/AU87/00259 12 1 since an initial calibration is sufficient to establish and Simplement sufficient corrective measures. Dynamic 3 misrepresentation is more difficult to correct. It can 4 arise from variations in the microphone coupling to the interference or to the desired signal. The former effect 6 can be reduced by careful design and the use of multiple 7 sensors and averaging or selecting the appropriate outputs.

8 Fluctuations due to variable signal coupling can be 9 compensated for by a dynamic control loop. A convenient calibration signal can be injected by addition to the 11 enhanced desired signal. This signal can be rendered 12 unobtrusive by making it sub-audible (by spread spectrum 13 techniques for example). Since this signal is deter- 14 ministic, it is not difficult to extract it by a matched filter from the filter picked up by sensing microphone 22, 16 an amplified by 38, which should be converted to a voltage 17 controlled amplifier just like 15 or 33. The gain control 18 is derived from the magnitude of the output of the matched 19 filter. The resultant loop functions just like a standard AGC maintaining the calibration signal output from 38 at a 21 constant level, thus compensating for microphone signal 22 coupling fluctuations.

23 Specific results of the enhancement of the signal to 24 interference ratio at the listener's ear as a result of the invention include improved speech intelligibility and noise 26 masking. Under normal conditions, the minimum ratio 27 required for acceptable recognition is plus 7 dB, ranging up 28 to plus 15 dB for hearing-impaired listeners. Thus an 29 electromechanical source of human speech is intelligible within a zone for which the above criterion is satisfied.

31 Without active source control, zone bo',idaries move in 32 response to variable interference and speech content.

33 Through its active source control, the signal enhancemeint 34 processor acts to suppress these fluctuations, maintaining the zone boundary extended as far as possible within the 36 limits of source power or distortion characteristics and 37 tolerances thereof. Secondary sources such as those used in 38 active noise cancellation can be controlled in a similar WO 88/01453 13 PCT/AU87/00259 1 manner, to improve performance of the cancellation 2 algorithm. The signal processor's effects diminish 3 progressively with decreasing interference levels. In noise 4 masking, the processor may help to attenuate the undesirable phsycho-acoustical effects of objectionable interference.

6 Its application may be indicated by aesthetic or therapeutic 7 reasons.

8 In summary, the automatic signal enhancement processor 9 of this invention is an extremely simple cost effective high performance signal enhancement system. The circuitry used 11 to construct all the building blocks may be analog or 12 digital, off the shelf or custom designed, precision or 13 economy. The last statement is a result of the tolerance of 14 closed loop operation to individual component errors because of its inherent compensation and suppression abilities. Of 16 course it will be readily apparent to persons skilled in the 17 art that modifications can be effected to the above 18 embodiment without departing from the spirit and scope of 19 the invention. For example, improvements in the integrated circuit packages may enable simplification and improved 21 performance. Thus, it is to be understood that the 22 invention is not limited to the particular embodiment 23 described, by way of example, hereinabove.

24 To assist in interpreting the present specification a glossary of some of the terms used hereinabove is set forth 26 below.

27 GLOSSARY 28 29 ENHANCEMENT In this context, enhancement is deemed to imply an 31 improvement in such qualities which results in imprcved 32 interpretation of the information conveyed by a signal.

33 SIGNAL 34 The term signal is deemed to include any means of conveying information, for example, acoustical, optical, 36 electrical, radio-frequency etc., or a stimulus applied to 37 evoke a measurable response, provided that it is capable of 38 unambiguous transformation into an electrical equivalent.

18 e 19: th neto.Freape mpoeet nteitgae WO 88/01453 14 PCT/ALU87/00259 -14- 1 DESIRED 2 The term desired implies that the signal conveys the 3 required message, or represents the required variable with 4 perfect precision, free from any pre-existing distortions.

In cases where a signal of suitable integrity is not 6 available, standard techniques of recovering or extracting 7 it from its perturbed or embedded state can be employed to 8 render it suitable. It is assumed that the signal 9 intercepted by the processor of the invention is the desirable signal.

11 RECEIVER 12 In this context, a receiver may be a person or device 13 with no control over the intefering variable. The receiver 14 may or may not have manual control over the source of the desired signal. Since the objective of the processor of the 16 invention is to render this control automatic, the result is 17 the liberation of the receiver from this burden, although 18 the application of manual override is not precluded.

19 INTERFERENCE The term interference is construed to include any 21 undesirable or competing signal incident on a receiver, thus 22 corrupting the quality of the received signal. It can be 23 natural or artificial, incidental or deliberate, static or 24 dynamic, deterministic or stochastic. The interference may be in its unabated form, or the residual after available 26 screening or suppression techniques have been implemented.

27 The interference may be additive or multiplicative or 28 complex in nature. It is assumed that the (unavoidable) 29 degradation of the desired signal within the processor is less than that occurring externally.

31 REPRESENTATIVE 32 In this context, a representative sample of the 33 corrupted signal is one whose functional statistical average 34 is closely related to its counterpart as measured by the intended receiver. An example might be the output of a 36 sensing microphone, with similar characteristics to a human 37 ear, placed where a listener might be.

Vi

Claims (2)

1. An adaptive gain control amplifier for converting an input signal to an analog controlled output signal in response to varying interference signals which mix with said output signal and result in a corrupted signal, said amplifier comprising: an amplifier stage for receiving said input signal and providing an output signal having a controlled output signal level; a prescaler circuit for receiving said output signal of said amplifier stage and comprising a secondary gain control amplifier for prescaling said output signal of said amplifier stage to provide a prescaled output signal; a time averaging circuit means, connected to said prescaler circuit, for providing a time averaged output signal of said prescaled output signal of said amplifier stage; se* a sensor means for detecting said corrupted signal and

40.- for providing a time averaged output signal of said :0 corrupted signal; a divider circuit having a first input receiving said time averaged output signal of said prescaled output signal from said amplifier stage, a second input receiving said time averaged output signal of said corrupted signal, and an output provided to a control input of said amplifier stage, said divider circuit comprising means for providing a i control signal at said output which is a ratio of said time averaged output signal of said corrupted signal and said time averagedz-output signal of said prescaled output signal i from said amplifier stage to said control input of said I.I amplifier stage; and wherein said prescaler circuit has a control input which recives a control signal which is proportional to said cointrol signal at said output of said divider circuit. 2. An adaptive gain control amplifier as defined in claim 1, wherein said corrupted signal is an audio signal degraded by interference, said analog controlled output signal is the output signal of a loudspeaker and said sensor psspe.016/aranda 90 6 7 4 CT i__.lli- i I; ~;ill- J~ii_;^i b 16 includes a microphone. 3. An adaptive gain control amplifier as defined in claim 2, further including a reference channel means for modifying an electrical signal representing said analog controlled output signal prior to application of said time averaged version of said prescaled signal to said divider circuit, said time averaging circuit means and said prescaler circuit forming part of said reference channel means and said prescaler circuit multiplying said electrical signal by said control voltage for prescaling said electrical signal. 4. An adaptive gain control amplifier as defined in claim 3 wherein said reference channel further includes a gain stage and filter circuit to scale the magnitude of said electrical signal and limit the bandwidth thereof, and a :6 precision rectifier/filter circuit to remove polarity S dependence of said electrical signal, said gain stage and *6*6 filter circuit and said rectifier filter circuit being •connected between said prescaler circuit and said time *10 averaging circuit, and said time averaging circuit 666 comprising an integrator circuit to time average said electrical signal. An adaptive gain control amplifier as defined in S claim 4, further including a sensing channel incorporating said sensor, said sensing channel further comprising a gain 6@ 9 stage and filter circuit and an integrator circuit for *9 modifying a further electrical signal prior to application to said divider circuit, said gain stage and filter circuit and said integrator circuit being connected between said :3I microphone and said divider circuit and said further electrical signal being the output of said microphone which is modified in a similar manner to said electrical signal representing said controlled output signal. 6. A method of processing an input signal to produce a controlled output signal when said output signal has an interference signal mixed therewith to produce a corrupted signal at a receiver, said method comprising: sensing and processing said corrupted signal to produce a i, psspe.016/aranda 90 6 -"W.Is L 1 i I ii 1: 17 first control signal for controlling processing of said input signal to compensate for said interference signal, said processing of said corrupted signal comprising producing a time-averaged version of said controlled output signal, producing a time-averaged version of said corrupted signal, computing a ratio of the modulus of said time averaged version of said corrupted signal and the modulus of said time averaged version of said controlled output signal to produce said first control signal, employing said control signal to control an amplification stage of an adaptive gain control amplifier which produces said output signal, providing a second control signal proportional to e said first control signal, and prescaling said controlled output signal in response fees to said second control signal prior to computing said ratio. 7. An adaptive gain control amplifier substantially as hereinbefore described with reference to the accompanying drawings. 8. A method of processing an input signal to produce a controlled output signal when said output signal has an interference signal mixed therewith to produce a corrupted signal at a receiver suostantially as hereinbefore described with reference to the accompanying drawings. .DATED this 20 June 1990 SMITH SHELSTON BEADLE .*j0 Patent Attorneys for the Applicant: ARANDA AUDIO APPLICATIONS PTY. LTD. psspe.016/aranda 90 6