JP2000235423A - Reference voltage generating circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a reference voltage generating circuit for generating a reference voltage which is not affected by the change of a surrounding temperature by using a band gap voltage having a low temperature coefficient as a reference. SOLUTION: This circuit is provided with a VT generating circuit 10 in which the emitters of a pair of transistors whose current density is different are commonly connected for generating a voltage being a difference between the base and emitter in proportion to a temperature T, a non-linear ΔVbe generating circuit 20 for receiving the output of the VT generating circuit 10, and for generating a ΔVbe having a current density rate in proportion to the temperature, and for multiplying this by m and outputting it, and a Vref outputting circuit 30 for allowing constant currents Ic to flow in a transistor, and for adding a voltage Vbe between the base and emitter of the transistor to the output of the non-linear ΔVbe generating circuit 20 and outputting it. Then, an output voltage equal to a band gap voltage can be obtained from this Vref outputting circuit 30.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a reference voltage generating circuit for generating a DC reference voltage, and more particularly to a base-emitter voltage Vbe of a transistor having a negative temperature coefficient and a voltage having a positive temperature coefficient. The present invention relates to an improvement of a bias circuit based on a bandgap voltage having a low temperature coefficient by weighting and adding (difference ΔVbe between base-emitter voltages of two transistors).

[0002]

2. Description of the Related Art Conventionally, a voltage generating circuit of this type is well known. For example, US Pat. No. 3,887,863 shows a voltage generating circuit as shown in FIG. This circuit has a pair of transistors Q01 and Q02 connected to a positive power supply line 1 via resistors RL1 and RL2, and the emitter of one of the transistors Q02 has a negative voltage via resistors RE2 and RE1 connected in series. Connected to line 2 and the emitter of the other transistor Q1 is connected to a resistor R2
It is connected to the common connection point of R1.

The collector voltages of the transistors Q02 and Q01 are connected to the input terminal of a high gain operational amplifier 3, and the output of the operational amplifier 3 is connected to the output terminal 4 and to the bases of the transistors Q02 and Q01. I have.

The difference ΔVbe between the base-emitter voltages of the two transistors Q01 and Q02 has a positive temperature coefficient,
Although the base-emitter voltage Vbe of the transistor Q01 has a negative temperature coefficient, they act so as to cancel each other in the configuration of FIG. The output voltage Vout is set to approximately the energy bandgap voltage Vgo to bring the temperature coefficient (TC) closer to zero. Although Vgo is 1.205 V in the case of silicon, excellent results can be obtained by setting the output voltage Vout to a slightly higher voltage.

For example, the output voltage is set to Vout = Vgo + (m−1) kT ₀ / q in order to make TC zero. Where m is a constant, approximately 1.5,
k is the Boltzmann constant, T ₀ is the operating temperature, and q is the charge of the electrons.

The output voltage Vout can be adjusted to a desired value by appropriately selecting the resistor RE1. And
If the output Vout falls below a preset optimum level, the transistor Q02, the current ratio I ₂ / I ₁ of Q01 becomes greater than the resistance ratio RL1 / RL2, the output of the amplifier 3 is increased, the output voltage Vout Raise to optimal level. When the output Vout becomes higher than the optimum level, the output of the amplifier 3 decreases, and the output voltage Vout is lowered to the optimum level.

[0007] The circuit shown in FIG.
The output voltage can be continuously maintained at a desired level with a low temperature coefficient.

[0008]

However, such a conventional circuit has the following problems. High-precision trimming, such as laser trimming, is required to perform optimum weighting using resistors. Even if appropriate weighting is performed, the temperature dependence of ΔVbe is linear, whereas the temperature characteristic of Vbe is nonlinear, so that the temperature dependence of about ± 25 ppm / ° C remains.

The present invention has been made to solve the above problem, and provides a reference voltage generating circuit for generating a reference voltage which is not affected by a change in ambient temperature, based on a bandgap voltage having a low temperature coefficient. It is intended for.

[0010]

In order to achieve the above object, according to the first aspect of the present invention, the emitters of a pair of transistors having different current densities are commonly connected to each other, and the base-emitter is proportional to the temperature T. A V _T generation circuit for generating a voltage having a voltage difference, a non-linear ΔV be generation circuit for receiving the output of the V _T generation circuit, generating ΔV be having a current density ratio proportional to temperature, multiplying the output by m, and outputting the result; A Vref output circuit that supplies a constant current Ic to the transistor and adds the base-emitter voltage Vbe of the transistor to the output of the non-linear ΔVbe generation circuit to output the same; It is characterized in that it is configured to be obtained.

[0011] to give the m × .DELTA.Vbe by V _T generator and a nonlinear .DELTA.Vbe generating circuit, the m × delta at Vref output circuit
A voltage Vref obtained by adding Vbe to Vbe is output. The voltage Vref at this time is as follows. Voltage Vref = Vbe + m × ΔVbe = Vgo- (kT / q) ln {(KT r) / (IcN m)} However, Vgo the bandgap voltage of Si at an absolute temperature of 0 degree, k is Boltzmann's constant, T is the absolute temperature , q is the elementary charge, K is a constant independent of the temperature, r is constant objection to the process, Ic is a constant current, N is the current density ratio in the V _T generation circuit, m is the magnification of the non-linear ΔVbe generating circuit

[0012] In this ^{case, N m = KT r / Ic} = AT r (A Section without the temperature) can be a Vref = Vgo With, achieve a bandgap reference voltage generating circuit which does not depend on temperature can do.

In this case, the nonlinear ΔVbe generating circuit is
For example, as in claim 2, obtained by passing the resistor having a temperature coefficient equal to the resistance of constant voltage from the current proportional to the temperature output through a resistor from V _T generating circuit regulated voltage source And a current can flow into each of the pair of transistors to generate ΔVbe having a current density ratio proportional to temperature.

Further, as the non-linear ΔVbe generating circuit, for example as in claim 3, the first to obtain the logarithm of current corresponding to the voltage difference between the base and emitter voltage proportional to the temperature T from V _T generator A logarithmic amplifier, a second logarithmic amplifier that obtains a logarithmic value of a current corresponding to a constant voltage from the regulated voltage source, and a logarithmic conversion circuit that outputs a difference between the outputs of the two logarithmic amplifiers. Can be configured by an m-fold circuit that multiplies m by m.

Furthermore, as according to claim 4, the pair of transistors of V _T generation circuit and nonlinear ΔVbe generating circuit, each can be configured whereby a transistor in three stages in series. With such a configuration, when the circuit of the present invention is formed into an IC, the effect of the package stress is reduced.

[0016]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described below in detail. The present invention is a reference voltage generation circuit based on a band gap voltage.

The output voltage (bandgap reference voltage Vref) in the circuit of the present invention can be represented by the following model equation. Vref = Vbe + m × ΔVbe = {Vgo- (kT / q) ln (KT r / Ic)} + {m × (kT / q) ln (N)} = Vgo- (kT / q) ln (KT r / ( IcN ^m )｝ (1) where m is a coefficient representing a magnification, Vgo is a band gap voltage of silicon (Si) at zero degree of absolute temperature, T is absolute temperature, ln is a natural logarithmic symbol, and N is a current density. The ratio, r is a process-dependent constant, Ic is a current value, k is a Boltzmann constant, and K is a constant independent of temperature.

Here, the expansion equation of Vbe in equation (1) will be described. First, the general formula of Vbe is as follows. Vbe = (kT / q) ln (Ic / Is) (2) where Is is a reverse saturation current and has the following relationship.

Is = qDAn ² / Q where D is the diffusion coefficient, A is the junction area between the emitter and base, n is the intrinsic carrier density, and Q is the amount of impurities per unit area in the base. Of these, q, A, and Q are invariant with respect to the temperature, and when this is set as a constant B, the above equation becomes: Is = BDn ² (3)

The Einstein relation D = kT
When μ / q (μ is the carrier mobility) is substituted, Is = kTμBn ² / q (4) Further, since k and q are invariant to the temperature, if these are included in B and newly set as a constant C, then Is = CTμn ² (5).

[0021] The mobility μ of carriers, μ = ET ^x and expressed. Here, E is a temperature-invariant term, and a multiplier x of T x
Is a value that changes with the impurity concentration of the base region. Also,
As is well known, the intrinsic carrier density is n ² = (FT ³ ) exp (−Vgo / Vt) (6) Here, F is a value that does not change with temperature. Vt
= KT / q.

[0022] When substituted into the μ and n ² (2) wherein deployment, relational expression of (1), Vbe = Vgo- (kT / q ) ln (kT γ / Ic) is obtained.

Now, in the equation (1), N ^m = KT ^r / I
By setting c = AT (A is a coefficient independent of temperature), that is, by making the current density ratio N in the _VT generation circuit proportional to temperature and by setting m = r, the second term of the equation (1) is obtained. Can be reduced to zero, so that terms including variations in elements can be canceled, and a bandgap reference voltage independent of temperature can be obtained. The present invention realizes a reference voltage generating circuit which is not affected by a change in ambient temperature based on such a principle.

FIG. 1 is a block diagram showing one embodiment of a reference voltage generating circuit according to the present invention. This reference voltage generation circuit
And V _T generation circuit 10 for generating a difference in base-emitter voltage of the emitter proportional to the common connection and the temperature of the different pair of transistors the current density, the current density ratio in proportion to the temperature from the output of the V _T generation circuit 10 And a Vref output circuit 30 for generating a reference voltage Vref by generating a Vbe by applying a constant current Ic to a transistor, adding the output to the nonlinear ΔVbe generation circuit 20, and generating a reference voltage Vref. Is done.

Hereinafter, each part will be described in more detail. V
_{The T} generating circuit 10 includes a pair of transistors Q0 and Q3 whose emitters are connected in common, and a high-side power supply (voltage Vc
c) and the other end is connected to each of the transistors Q0 and Q3.
And a constant current source 17 having one end connected to the emitters of the transistors Q0 and Q3 and the other end connected to a low voltage source (voltage Vee). An operational amplifier OP10 having differentially received the collector voltages of transistors Q0 and Q3 and having its output connected to the base of transistor Q3;
A resistor R2 is connected between the bases of 0 and Q3.

According to such a connection, the transistor Q
0, between Q3 _{based, V T = (kT / q} ) lnN voltage V _T generated composed, hence the resistor R2 (the resistance value _{R 2), I 1 = (} kT / qR 2) lnN becomes current I ₁ flows.

The nonlinear ΔVbe generation circuit 20 includes a pair of logarithmic amplifiers Q6 and Q9, an operational amplifier OP11, and a resistor R3.
, An operational amplifier OP12 as a buffer amplifier, and a series connection circuit formed by connecting resistors R4 and R5 in series.

Here, transistors are used as the logarithmic amplifiers Q6 and Q9 (Q6 is called a first logarithmic amplifier and Q9 is called a second logarithmic amplifier). V _{T to} the collector of Q6
Current I ₁ flows generated in the generator 10, while the collector of Q9 current I ₂ flowing through the resistor R3 flows. This current I ₂ can be adjusted by the reference voltage V _AJ applied to the resistor R ₃ (the resistance value is R ₃ ). Note that the base of the transistor Q9 is connected to the common line gnd.

The operational amplifier OP11 has a non-inverting input terminal connected to the common line, an inverting input terminal connected to the collector of the transistor Q9, and an output terminal commonly connected to the emitters of the transistors Q6 and Q9. In this case, the inverting input terminal of the operational amplifier OP11 is equivalently at the common line potential, and one end of the resistor R3 is at the common line potential.

The operational amplifier OP12 has a non-inverting input terminal connected to the collector of the transistor Q6, and an output terminal connected to the inverting input terminal connected to resistors R4 and R5 connected in series. The common connection point of the resistors R4 and R5 is connected to the base of the transistor Q6.

When the resistance R4 and the resistance R5 are m = (R4 + R
5) If / R5, the output terminal of the operational amplifier OP12 has a voltage m × ΔV that is m times the voltage ΔVbe at the middle point of the series connection circuit.
Since be occurs, a portion including the operational amplifier OP12 and the series connection circuit is referred to as an m-fold circuit here. The voltage ΔVbe at the midpoint of the series connection circuit is the difference between the base-emitter voltages of the transistor Q6 and the transistor Q9.

The Vref output circuit 30 includes: a constant current source 18;
It comprises a buffer 16 and a transistor Q12.
The constant current source 18 has one end connected to the high voltage side power supply and the other end connected to the collector of the transistor Q12. The emitter of the transistor Q12 is connected to the nonlinear ΔVbe generation circuit 20.
Of the operational amplifier OP12.

The input terminal of the buffer 16 is connected to the transistor Q1
2 and the output terminal is connected to the transistor Q12.
And the output terminal VrefOUT.

The operation in such a configuration will now be described. Current I ₂ flowing through the resistor R2 of the V _T generation circuit 10 is _{I 2 = (kT / qR2)} lnN, flows the current to the transistor Q6. At this time, the base-emitter voltage Vbe1 of the transistor Q6
Is as follows. Vbe1 = (kT / q) ln (I 1 / I s) = (kT / q) ln {(kT / (q · Is · R2)) lnN}

On the other hand, the current I ₂ = V _AJ / R3 flowing through the resistor R3 is input to the transistor Q9,
The emitter-to-emitter voltage Vbe2 is as follows. Vbe2 == (kT / q) ln (I 2 / I s) = (kT / q) ln (V AJ / (Is · R3)

Therefore, the voltage ΔVbe at the middle point of the voltage dividing circuit
ΔVbe = Vbe1−Vbe2 = (kT / q) ln (CT /
V _AJ ) where C = {kR3 / (qR2)} lnN, and the voltage m × ΔVbe at the output terminal of the operational amplifier OP12 is m × ΔVbe = m × (kT / q) ln (CT / V _AJ ) = ( kT / q) ln (CT / V _AJ ) ^m where m = (R4 + R5) / R5.

Therefore, the output terminal Vr of the Vref output circuit 30
voltage Vref efOUT becomes Vref = Vbe + m × ΔVbe = Vgo- (kT / q) ln {(V AJ m KT r) / (I c C m T m)}. Here, if r = m, V _AJ ^m K = I _c C ^m , then Vref = Vgo, and a reference voltage which is not affected by changes in the ambient temperature can be obtained.

According to the present invention, the output can be adjusted to the known constant voltage Vgo in this way, the variation can be canceled, and the temperature coefficient ± 0 ppm / ° C can be realized. However, when this circuit is implemented by an IC, the bandgap voltage of each transistor changes due to the package stress, and a certain constant term is added to the above logical expression, so that it is difficult to achieve one-point adjustment.

FIG. 2 is an example of a circuit configuration capable of reducing the influence of such package stress. That, V _T generation circuit 10 and nonlinear ΔVbe
The pair of transistors of the generation circuit 20 are each stacked in several stages (three stages in the figure), thereby easily reducing the influence of stress.

In the circuit configuration of FIG. 2, the nonlinear ΔVbe
The magnification of the m-times circuit of the generation circuit 20 can be reduced (m
/ 3), and the error is not greatly amplified, which is extremely effective in practice.

FIG. 3 shows the temperature characteristics of the output voltage in comparison with the conventional example. Circle plots indicate the characteristics of the reference voltage generating circuit of the present invention, and square plots indicate the characteristics of the conventional circuit. It can be seen that the temperature characteristics of the present invention are much better than those of the prior art, and the fluctuations due to temperature are extremely small. The oblique straight lines shown for reference are +25 ppm / ppmC and -25 ppm / ゜ C.
Represents the temperature change.

[0042]

As described above, according to the present invention, the following effects can be obtained. According to the first aspect of the present invention, it is possible to easily generate a bandgap reference voltage independent of temperature.

According to the fourth aspect of the present invention, a circuit for reducing the influence of package stress can be easily realized.

[Brief description of the drawings]

FIG. 1 is a configuration diagram showing one embodiment of a reference voltage generation circuit according to the present invention.

FIG. 2 is a configuration diagram showing another embodiment of the present invention.

FIG. 3 is a temperature characteristic comparison diagram between a reference voltage generation circuit of the present invention and a conventional circuit.

FIG. 4 is a configuration diagram illustrating an example of a conventional reference voltage generation circuit.

[Explanation of symbols]

10 V _T generating circuit 16 Buffer 17, 18 Constant current source 20 Nonlinear ΔVbe generating circuit 30 Vref output circuit Q0, Q3, Q6, Q9, Q12 Transistors R0, R1, R2, R3, R4, R5 Resistance OP10, OP11, OP12 Operation amplifier

Claims (4)

[Claims] 1. A The emitters of the current density is different pair of transistors commonly connected, and V _T generating circuit for generating a voltage difference proportional to the base-emitter voltage of the temperature T, the output of the V _T generator And a non-linear .DELTA.Vbe generating circuit for generating a .DELTA.Vbe having a current density ratio proportional to the temperature and multiplying the multiplied by m and outputting a constant current Ic to the transistor, generating a base-emitter voltage Vbe of the transistor and generating the non-linear .DELTA.Vbe. A reference voltage generating circuit, comprising: a Vref output circuit for adding the output of the circuit and outputting the sum, and configured to obtain an output voltage equal to the band cap voltage from the Vref output circuit. Wherein said non-linear ΔVbe generating circuit, through a resistor having a temperature coefficient equal to the resistance of constant voltage from the current and the regulated voltage source proportional to the temperature to be output through the resistor from the V _T generating circuit 2. The reference voltage generation circuit according to claim 1, wherein the currents obtained as described above are respectively supplied to a pair of transistors to generate ΔVbe having a current density ratio proportional to temperature. Wherein said non-linear ΔVbe generating circuit includes a first logarithmic amplifier obtaining a logarithmic value of the current corresponding to the voltage difference between the base and emitter voltage proportional to the temperature T from the V _T generation circuit, adjusting A logarithmic conversion circuit for outputting a difference between an output of the second logarithmic amplifier and a logarithmic value of a current corresponding to a constant voltage from the voltage source; 2. The reference voltage generating circuit according to claim 1, further comprising a circuit. Wherein said V _T generation circuit and nonlinear ΔVbe generating circuit includes a reference voltage generating circuit according to claim 1, wherein the each of the pair of transistors is configured whereby a transistor in three stages in series.